K7ZGQ-ANTENNAS,P.F.O. Skunkworks

SB-200 AMPLIFIER CAPACITOR VALUES AND LOCATION BREAKDOWN

2026-08-31T17:38:00.000-07:00

The Heathkit SB-200 contains approximately 20 capacitors across its RF, power supply, and bias circuits. Below is a comprehensive list of every capacitor, its location, original value, and a valid modern replacement part number.

Note on Part Numbers: Modern capacitors are rarely sold with the exact legacy Heathkit part numbers. The replacements listed below are industry-standard equivalents available from suppliers like Digi-Key, Mouser, or RF Parts Company. Voltage ratings for replacements are equal to or higher than originals for safety.

1. High Voltage Power Supply (Main Filter & Bleeder)

Located on the main power supply board or chassis mount near the transformer. These are critical for safety and performance.

Component	Location	Original Value	Modern Replacement (Part Number Example)
C1 – C6	HV Filter Bank (6 caps in series)	125 µF, 450V Electrolytic	Panasonic ECG-UPA2V121 (120µF, 350V, use 6) or Cornell Dubilier 380LX121M450 (120µF, 450V). Upgrade: Use six 220 µF, 450V (e.g., Nichicon UHE2W221MHD) for better regulation, but add a Soft-Start kit to prevent inrush trips.
R-Bleeder	Across each C1-C6	(Resistor, not cap)	N/A
C7	HV Metering Filter	0.05 µF, 600V Paper/Plastic	Vishay MKP1848550064 (0.5µF is too high, use 0.047 µF, 630V Film like WIMA FKP2). Original was often 0.05uF; 0.047uF is standard.

2. Anode (Plate) Circuit & RF Output

Located near the tube sockets, band switch, and output tank coil. These handle high RF voltage and must be high-voltage ceramic or mica.

Component	Location	Original Value	Modern Replacement (Part Number Example)
C8	Anode Choke Bypass	1 nF (0.001 µF), 3kV Ceramic	Vishay 561R30TCC102 (1 nF, 3kV) or CDR31BX102BKSR (Mil-Spec). Mod: Replace with 5.6 nF (0.0056 µF), 3kV (e.g., Vishay 561R30TCC562) to improve 80m output.
C9	HV Feedthrough/Blocking	500 pF, 5kV Doorknob	CDR32BX501BKSR or generic 500 pF, 5kV Ceramic Doorknob (e.g., RF Parts C500-5).
C10	Output Tank (Fixed)	100 pF, 5kV Mica	CDR31BX101BKSR (100 pF, 1kV is too low, ensure 5kV rating). Use Cornell Dubilier DM19-5 series or equivalent HV Mica.
C11	Parasitic Suppressor	(None, part of choke)	N/A
C25	Plate Tuning Capacitor	150 pF Variable	Johnson 154-15 (150 pF, 5kV Variable). If replacing sections, ensure voltage rating >4kV.
C26	Plate Loading Capacitor	457 pF Variable (Dual Section)	Johnson 154-30 (approx 250pF per section in series/parallel config) or Hammarlund HF-150-X modified. Often requires finding a surplus Dual 250pF, 3kV variable.
C27	80m Padder (Load)	500 pF, 3kV Mica	CDR32BX501BKSR (Ensure 3kV rating).
C28A/B	See C26 above	(Sections of C26)	N/A

3. Input Tuned Circuit (Grid)

Located on the input band switch wafer and near the tube grid pins. Critical for SWR matching.

Component	Location	Original Value	Modern Replacement (Part Number Example)
C29	ALC Sample	5 pF, 1kV Mica	CDR30BX5R0BKS (5 pF, 1kV).
C30	80m Input	470 pF, 1kV Mica	CDR31BX471BKSR (470 pF, 1kV). Mod: Change to 910 pF for better 80m SWR (see NE7X mod).
C31	80m Input (Shunt)	(Part of L network)	N/A
C32	40m Input (Series)	310 pF, 1kV Mica	CDR31BX311BKSR (300 pF or 330 pF standard).
C33	40m Input (Shunt)	510 pF, 1kV Mica	CDR31BX511BKSR (510 pF, 1kV).
C34	20m Input (Series)	200 pF, 1kV Mica	CDR31BX201BKSR (200 pF, 1kV). Mod: Some reduce to 100 pF to lower Q/heat.
C35	20m Input (Shunt)	360 pF, 1kV Mica	CDR31BX361BKSR (360 pF, 1kV). Mod: Some increase to 470 pF.
C36	15m/17m Input	75 pF, 1kV Mica	CDR31BX750BKSR (75 pF, 1kV). Mod: Replace with Trimmer (e.g., Johnson 142-0101-801) for tuning.
C37	10m Input	66 pF, 1kV Mica	CDR31BX680BKSR (68 pF standard). Mod: Replace with Trimmer.
C38	Grid Bypass (Per Tube)	200 pF, 1kV Ceramic	CDR31BX201BKSR. Critical: Upgrade to 1 nF - 1.5 nF, 1kV (e.g., CDR31BX102BKSR) for better stability with modern 572B tubes.
C39	10m Input (Shunt)	(Varies by config)	Check schematic; often 100 pF.

4. Bias, Filament, and Relay Circuits

Low voltage electrolytics and ceramics located on the bias board, filament choke, and relay assembly.

Component	Location	Original Value	Modern Replacement (Part Number Example)
C12	Filament Choke Bypass	20 nF (0.02 µF), 50V+	Vishay K5050203Z or any 0.022 µF, 50V Ceramic Disc. Mod: Reposition to grid pin for better RF grounding.
C13	Bias Rectifier Filter	20 µF, 150V Electrolytic	Nichicon UHE2E220MHD (22 µF, 250V).
C14	Bias Filter	20 µF, 150V Electrolytic	Nichicon UHE2E220MHD (22 µF, 250V).
C15	Relay Delay/Filter	20 µF, 150V Electrolytic	Nichicon UHE2E220MHD (22 µF, 250V).
C16	Input Coupling	0.02 µF (20 nF), 600V	Vishay MKP1848420064 (0.22uF is wrong, use 0.022 µF, 630V Film).
C17	ALC Filter	0.1 µF, 50V Electrolytic	Nichicon UHE1H101MHD (100 µF is too high, use 10 µF, 50V or 0.1 µF Film). Original was likely electrolytic; 10 µF, 50V is safe.
C18	Meter Lamp Bypass	0.02 µF, 50V Ceramic	Vishay K5050203Z (0.022 µF, 50V).
C19	Relay Supply Filter	2.2 µF, 250V Electrolytic	Nichicon UHE2E2R2MHD (2.2 µF, 250V Axial).
C20	Relay Supply Filter	2.2 µF, 250V Electrolytic	Nichicon UHE2E2R2MHD (2.2 µF, 250V Axial).

Critical Modification Summary

Anode Choke (C8): Changing from 1 nF to 5.6 nF (or adding 4.7 nF in parallel) is the single most effective mod for 80m power.
Grid Bypass (C38): Increasing from 200 pF to 1.5 nF improves stability with modern tubes but requires retuning the input network.
Power Supply (C1-C6): Upgrading to 220 µF or 390 µF improves voltage regulation but requires a soft-start circuit (e.g., Harbach SK-200) to avoid blowing breakers.

Capacitor Materials in HF amplifiers

2026-08-28T03:58:41.310-07:00

For the Heathkit SB-200 (572B tubes) and SB-220 (3-500Z tubes), the critical capacitor positions are the RF input/output pi-network tuning and the grid bypass capacitors (200 pF in the SB-200, which are specifically chosen to resonate with the tubes' internal grid leads near ~70 MHz for VHF parasitic suppression).

This makes dielectric choice far more consequential than in a typical circuit.

Ceramic Dielectric Comparison for These Amps

Dielectric	Class	Stability	Loss / Q	Suitability in SB-200 / SB-220
C0G / NP0	1	±30 ppm/°C	Very low (tan δ ≈ 0.0015), high Q	Only recommended ceramic. Used at 1–2 kV rating for RF tuning and grid bypass. Preserves the grid-lead resonance and pi-network Q.
C0H	1	±60 ppm/°C	Very low	Technically fine but rarely stocked; no practical advantage over C0G in these amps.
P100, N33, N75	1	Compensating	Low	Designed for temperature-compensation networks, not general RF bypass. Not a substitute.
X7R	2	±15%	Moderate, piezoelectric (microphonic)	Not suitable for RF tuning or grid bypass. Capacitance drifts with temperature and DC bias, shifting the grid-lead resonance and detuning the pi-networks. Acceptable only for non-critical power-supply bypass.
X5R	2	±15%	Moderate	Same limitations as X7R, narrower temp range. Not suitable for RF-critical positions.
X8R	2	±15%	Moderate	Higher density, same drift problems. Not suitable.
Y5V	3	+22% / −82%	High loss	Unsuitable. Large capacitance swing would detune the RF networks and could destabilize the grid-lead resonance.
Z5U	3	+22% / −56%	High loss	Unsuitable for the same reasons.

Practical Guidance

RF tuning (pi-networks) and grid bypass: Use C0G/NP0 ceramic rated 1–2 kV (or the original silver mica at 500 V, which is the stock choice). The SB-220 literature explicitly recommends "1000 V ceramic, NP0 type" for these positions. Do not substitute X7R, Y5V, or any Class 2/3 dielectric here — the capacitance drift will shift the parasitic-suppression resonance and degrade SWR and output.
Power-supply bypass (filter caps, filament bypass): X7R or even Y5V would work electrically, but these positions in the SB-200/SB-220 are traditionally filled with electrolytic or paper caps, not ceramic, so it's rarely a question in practice.
Grid bypass value must stay at 200 pF (SB-200): The value is not arbitrary — it's tuned to the 572B's internal grid-lead inductance. Changing the dielectric changes the effective capacitance under operating conditions (especially for Class 2 types), which can break the ~70 MHz series resonance and allow VHF parasitics.

ADDING 160 METER COVERAGE TO HF AMPLIFIERS

2026-08-21T00:50:41.531-07:00

ADDING 160 METER COVERAGE TO HF AMPLIFIERS

(updated original manuscript, article published in January 1989 *QST*)

by Richard L. Measures, AG6K, April 1988

WHY?

If you have ever tried to use the 80 meter phone band for local communications on winter evenings, you probably experienced situations where under 200Km propagation was almost nonexistent.

The group of friends that I talk with on 80 meters was plagued by these "winter doldrum" conditions so a few of us put up 160 meter antennas to see if the situation could be improved. We discovered that, when the 80 meter band was essentially useless for short-haul propagation, the 160 meter band provided good communication.

There were tradeoffs to using the 160 meter band: Most of us had to put up compromise antennas that would fit on a city lot -- like the end-fed Marconi . These antennas were not as efficient as full-size dipoles. The only practical way to compensate for the lack of efficiency was to increase power. Another problem with 160 meters was a peculiar type of fading that would take a 100 watt signal down into the sky-noise for 10 to 20 seconds every couple of minutes. Again, the solution was to increase power.

Most HF amplifiers that were manufactured in the last twenty years did not include 160 meters because high power was permitted there only during the nearly useless daylight hours. The easiest but not the most cost effective or educational way of increasing power on 160 meters is: sell the old 80 through 10 meter amplifier and purchase a new model that has 160 meters included. I chose to convert my old SB-220 to include 160 meters and pocket the knowledge and experience plus the roughly 600 dollar difference.

After converting my own amplifier, I have been involved in modifying six other amplifiers. 160 meters can be added to almost any similar amplifier by using the same basic principles.

What follows is the theory, practice and pitfalls or "gotchas" of this modification.

HOW

To make an 80 meter circuit work on 160 meters, the capacitances and the inductances must be doubled. This includes the components in the output and input resonant circuits as well as the bypass capacitors and the anode RF-choke but usually not the filament-choke as will be explained later. The added bypass capacitors and the added anode RF-choke can be left in the circuit for operation on the other bands, but the 160 meter tuned-circuit components must be switched into the circuit for use on 160 meters only and switched out of the circuit for operation on the other bands. . .

If extra bandswitch contacts [75m/80m] had been provided by the manufacturer [like on the Henry Radio Company's 2K-4 and 3KA models] , the job of switching would be simpler. Therefore the switching must be done with relays controlled by a two position toggle-switch [S1a]. One switch position is used for 80 through 10 meters and the other switch position is for 160 meters only, with the main bandswitch set to the 80 meter position.

DIAGRAM

COMPONENT RATINGS:

The ratings of the components that are used in this project deserves special attention because we are dealing with surprisingly high RF-currents and RF-voltages. The RF circulating-current in a tank-circuit is much higher than the indicated anode-current [in this case,.8a] might imply.

CAPACITOR CURRENT-RATINGS:

The capacitors used must not only be rated for voltage, they must also be rated for the substantial RF currents encountered. For example, just because a 500pF "TV doorknob" capacitor is rated at 20,000VDC does not mean that it could handle even 530vRMS at 1.8MHz. At 1.8MHz, the reactance of 500pF is minus j176.7ohm. The {apparent} current through the capacitor is: 530v/176.7ohms = 3 amperes which would cause severe heating, a rapid change in capacitance and a meltdown in this type of capacitor.

Special capacitors that are made to handle RF current are required for this project. Some examples of RF current-rated types of capacitors are the transmitting mica, ceramic, transmitting "doorknob" and air or vacuum-dielectric capacitors.

The RF [circulating or flywheel] current in the tank-circuit of an amplifier is roughly equal to "Q" multiplied by the DC anode-current. In this case, the circulating current is [Q=12] x [.8A] c.10a. The tuning capacitor and the loading capacitor must be able to withstand this substantial RF-current. When the tuning capacitor's padder is switched into the circuit on 160 meters and the tuning capacitor is set to minimum capacity, the padder capacitor must be able to withstand more than c.90% of the RF circulating current.

The 300pF tuning-capacitor padder [C55a] is three 100pf, 5KV or two 150pF, 5KV ceramic transmitting capacitors in parallel. The 100pF capacitors are rated at 3.4 amperes each at 1MHz.

Note: Just because a transmitting ceramic capacitor is designed to handle RF-current does not guarantee that it will work in a tank-circuit. For example, a 500pF, 5KV "Series 58 Centralab™ {now Jennings™ "Series 58"} capacitor is RF-current rated but it is rated to handle only 1.1 amperes at 1MHz and 2.5 amperes at 10MHz. The rating at 1.8MHz would probably be around 1.5 amperes. This capacitor would be unsuitable for either the tuning capacitor padder [assuming that a 500pF size was needed] or the loading capacitor padder in this project. Even the 1000pF capacitor from the same series of capacitors would be unsuitable for the loading capacitor padder [C56a] since its current rating is only 1.4a at 1MHz.

RELAYS: The current and voltage ratings of the open-contacts in the output switching relays, RL1a and RL2a, deserves special consideration because RF voltage is much more capable of jumping across open switch contacts than voltage with a frequency of 60Hz. Breakdown voltage is inversely proportional to the frequency of the voltage. At 30MHz, the breakdown voltage is roughly half of the 60Hz value.

CONDUCTOR RATINGS: RF current travels on the surface of conductors and ignores the substantial path for current below the conductor's skin. This "skin-effect" means that the current is concentrated near the surface of an RF conductor so the current density there is much higher than it would be with DC or 60Hz current. The skin-effect becomes more pronounced as frequency increases. The increased current density with RF causes exponentially [current squared times resistance] more heating at the surface of the conductor and the current rating of wire and contacts must be proportionally derated as frequency increases. For example, #12 copper wire is normally rated for 20a at 60Hz but the 30MHz rating for continuous current in #12 copper wire is only c.5 amperes.

Relays that have RF current ratings are normally derated to about 20 to 30% of the DC or 60Hz ampere rating for use at 30MHz.

Unlike RF voltage breakdown ratings, for intermittent applications like CW and SSB, the RF current ratings of wire and relay contacts may be increased; the only tradeoff being a slight decrease in overall efficiency.

VOLTAGE RATINGS: The maximum RF peak voltage that is encountered by the tuning capacitor in a properly tuned, 3-500Z, grounded-grid amplifier is approximately equal to the anode supply voltage minus about 200v. This would be about 2700v minus 200v c. 2500V peak for the SB-220.

Note: The reason for this is that when the peak anode-current is maximum and the instantaneous anode-voltage is minimum, the anode must always be about 200 volts more positive than the [grounded] grid in order to attract most of the electrons that are emitted by the cathode. If the amplifier is adjusted for too-light loading, which causes the instantaneous, minimum, anode-voltage to dip below the c.+200v level, too many electrons will flow into the grid instead of into the anode. This causes excessive grid-current and, because those electrons never make it to the plate, results in amplifier non-linearity which is euphemism for rotten-splatter.

2500v is not difficult to switch at 60Hz. But at 29MHz it is not so easy. For example: If a relay with a 3000v peak/60Hz rating were used to switch the padder-capacitor across the tuning-capacitor [RL1a], its open contacts would arc- over during operation on the higher frequency bands. This means that a special relay that has an RF voltage rating must be used. The only practical relay for this difficult job is a vacuum-relay that is [open-contact] rated for at least 2500v peak at 30MHz. A suitable relay is the Jennings™ RD 5 or RB 1 or a Kilovac™ H-8. There are many other vacuum-relays to choose from that are rated at �2500v peak at 30Mhz. Those that I listed are more common at swap meets.

The SPDT relay, RL2a, at the low voltage, 50ohm end of the tank-circuit switches the added loading-capacitor padder and the added 9micro H, tank-inductor padder. The maximum RF voltage that is encountered at these points is less than 1200v peak. An ordinary 15a open-frame relay may be used if the open contact air-gap is above average and it will withstand >4500VDC on a breakdown voltage tester. Another Jennings™ RB1 or a Jennings™ RJ1A [like the one seen in the photograph] or a Kilovac™ H-8 would be even better.

POWERING THE RELAYS

The rated coil current for an RB1 is 100mA. If speed of relay closure is not a factor, as in this application, a lower actuating current may be used. The actual pull-in current for a RB1 is about 50mA. For reliable operation and maximum coil life, a coil current of c.70mA is appropriate.

The stock, half-wave rectified, +110V relay power supply in the SB-220 is adequate for the original current load of about 25mA. The extra current load that the 160 meter switching relays place on this supply can be accommodated by changing the rectifier circuit from the stock single-diode half-wave to a four-diode full-wave-bridge rectifier circuit. This involves removing the half-wave rectifier [D16] and replacing it with a c.1a,�200piv full-wave bridge rectifier unit. The previously grounded side of the transformer winding {red wire} must be lifted above ground and connected to the input of the new rectifier unit. See the schematic diagram.

One DPDT Jennings® RB 3 can do the jobs of RL1a and RL2a, but the RB3's rated coil current of 200mA {140mA} is probably too much for the SB-220's internal +110V power supply so an extra 18 to 24 volt power supply would be required.

FILAMENT CHOKE

Most articles that have been written about 160 meter amplifier modification mistakenly presume that it is necessary to throw away the stock 80 meter to 10 meter filament RF-choke and replace it with a more inductive choke.

The SB-220 has a 9micro H filament choke. The reactance of this choke at 1.9MHz is about plus j107ohm. If a capacitor whose reactance at 1.9MHz was minus j107ohm [800pF] was connected in parallel with the 9micro H choke, the circuit would resonate and look like the needed high impedance at 1.9MHz , the original choke would do the job.

The off-frequency impedance, at the 160 meter band edges, of this resonant filament-choke circuit deserves evaluation: The admittance, Y, of a 9micro H inductor in parallel with a 800pF capacitor calculates to be Yc.1/900S [siemens {previously called "mhos"}] at 1.8MHz and 2MHz. This admittance is equivalent to an impedance of Zc.900ohm which will make little difference to the RF-paralleled cathode-circuit of the 3-500Zs, whose [averaged over 360 degrees] impedance is about 65ohms.

The extra 800pF is added to the capacitance of the output capacitor in the 160 meter, pi-network tuned-input circuit so that it is automatically switched in only on 160 meters.

TUNED-INPUT CIRCUIT

Before the values of the components in the 160 meter band pi-network, tuned-input circuit can be calculated, a value of "Q" must be chosen. "Q" is defined as the input resistance of the pi-network divided by the reactance in ohms of the input capacitor, or: Q=Rin/Xcin. If the "Q" is too low, the bandwidth of the tuned-input will be good but the minimum attainable SWR will not be especially good anywhere on the band. If the "Q" of the input circuit is too high, the SWR at the middle of the band will be good but the SWR at the band edges will not be good. So a compromise "Q" is chosen to balance bandwidth against minimum attainable SWR. According to Eimac™, in their book Care And Feeding Of Power Grid Tubes, an input-circuit "Q" of c.2 should be used. The stock SB-220 uses a "Q" of only c.1 for the 80 meter through 10 meter tuned-input circuits. This results in an SWR that is acceptable for vacuum-tube output transceivers, but less than optimum for modern all-solid-state transceivers.

Using the formulas in Eimac's book and a "Q" of 1.6, the input capacitor, C1a c.2700pF, the inductor, L1a, is between 4micro H and 5micro H and the output capacitor, C2a c.2500pF plus the extra 800pF [total c. 3300pF] needed to cancel the reactance of the 9micro H filament choke. These capacitors must be rated at �500V with a mica dielectric. The inductor, L1a, should be wound from no smaller than 24 gauge insulated copper wire or the input SWR will be degraded. If the inductor is slug-tunable, it partially facilitates adjusting the input SWR.

ADJUSTING INPUT SWR: The impedance transformation of a pi-network can not be altered by adjusting only one of the three reactive components. The impedance transformation can be altered only by adjusting two or more components.

The 160 meter pi-network's input capacitor, C1a, is not adjusted because to do so would change the design "Q" of the circuit. Adjusting the inductance will change only the center-frequency and may not result in an acceptable minimum SWR. If this is the case, C2a should be adjusted as well as L1a until the proper impedance transformation ratio and minimum SWR is achieved.

ANODE-CHOKE

Designing an anode-choke to supply the HV to the anode of an amplifier-tube would be easy if the amplifier were to be used on only one frequency. If this were the case, the unavoidable, internal self-resonant frequencies of the choke could be "parked" elsewhere by adjusting the number of turns of wire and there would be no problem.

When an amplifier is to be operated over a wide frequency range, the design of the choke is complicated. The problem is this: Enough inductance must be provided so that, at the lowest operating frequency, there will be enough inductive reactance to limit the RF current that flows through the choke and the HV-to-ground RF-bypass capacitor [C7]. The solution sounds simple, just add more turns to make more inductive reactance to reduce the RF-current that flows through the choke. The complication is this: adding turns also increases the choke's distributed capacitance which increases the number of choke self-resonances below 30MHz. If one of these self-resonances falls on an operating frequency, an unpleasant choke-fire will result and the choke will be crispy-crittered.

One traditional non-solution to the problem of adding more turns was to allow an open space of c.2cm on the coil form between the added turns and the main winding. This was supposed to decouple the inductors and avoid the self-resonance problem. This solution looks good on paper but unfortunately, as can be easily confirmed by measuring the self-resonances with a dip-meter, same-axis coil sections that are placed end-to-end couple optimally to each other and the spacing must be increased far beyond any practical limit to achieve the desired results.

The best way to prevent inductive coupling between two or more inductors is to place the inductors at right-angles to each other.

The stock anode-choke, RFC1, in the SB-220 has 50micro H of inductance. This value was chosen wisely because the lowest self-resonant frequency with this amount of inductance is above 40MHz, which is safely above the operating frequency range of the amplifier. The reactance of this choke at 1.8MHz is +j565ohm. The AC anode-voltage in the SB-220 c. 2500v peak/�2 c. 1768vRMS, so the RF current through this choke would be 1768v/565ohm = 3.13a! This substantial current would cause severe heating of the #28 gauge wire in the choke and the HV bypass capacitor at the bottom of the choke so another choke is added in series with, and at a right-angle to, the original choke. The inductance of this choke [RFC1a] can be as little 65micro H or as much as 135micro H.

A 65micro H choke can be made by filling c.51mm of winding space on a 16mm diameter coil form with .36mm diameter [#27AWG] high-temperature enameled copper wire . This choke's lowest self-resonant frequency is c.35MHz so it will work well below 30MHz and also provide enough total inductance [115micro H] to avoid choke and bypass capacitor overheating at 1.8MHz. {NOTE: To avoid a meltdown, the coil form/insulating tube should not be made from PVC, nylon, or any other material that has a high RF-dissipation [D] factor. Delrin™ has an intermediate D factor. Polyethylene, TFE [Teflon], and ABS [acrylonitrile butadiene styrene] or other styrenes have low D factors. Thin-wall (c..5mm) G10 or G11 fiberglass tubing is excellent for this application because it is light weight, RF and heat resistant and very strong. It is so light that it can be supported by the #18 [1mm] solid wires that connect the choke into the circuit.

Coil form diameters up to c. 25mm can be used to make RFC1a. The inductance can be anything up to about 135uH as long as a self-resonance can not be found within about 5% of the edge of any operating frequency.

NOTE: The self-resonant frequencies of the choke are found by connecting the choke's two terminals together with a short cliplead, coupling a dipmeter to either end of the choke and sweeping the operating frequency range. The dips are quite sharp and are easy to find.

A suitable 135micro H choke can be made by filling c.46mm of winding space on a 25mm coil form with .36mm copper wire. This choke is self-resonant at 23MHz and 35MHz so it is safe for use on the 15 meter, 12 meter and 10 meter bands. A compact version of the 65micro H choke can be made by filling c.40mm of winding space on a 25mm diameter coil form with .36mm wire.

Other sizes of wire from 24 gauge to 28 gauge may also be used for RFC1a if the winding space on the coil form is adjusted accordingly. The self-resonant frequencies should be tested in each case and turns can be removed if needed to park the self-resonances in safe zones between the amateur-radio bands. The actual inductance is far less critical than the resonances. [1]

The finished choke should be given a thin coating of gloss, polyurethane varnish to prevent rattling during voice modulation.

INSTALLATION: The doorknob capacitor, C6, at the base of RFC1 is disconnected. RFC2 is removed and replaced by RFC1a. C6 may be removed or re-connected in parallel with C7 and C7a for improved bypassing of the HV lead.

If the compact version of the 65micro H choke is used, it is not necessary to remove the HV-interlock to make room for RFC1a. If a larger choke is used for RFC1a, the interlock must be removed to make room.

The low-grade ceramic coil-form that was used for the stock anode-choke [RFC1] can not withstand the extra RF voltage that is present at the [grounded] base of the choke when RFC1a is added to the anode-choke circuit. The phenolic insulator from the interlock or a #6-32 threaded ceramic spacer is used to replace the aluminum spacer under RFC1. Extra washers are needed to make the phenolic spacer the same length as the original aluminum spacer.

SAFETY NOTE: the presence of the HV-interlock does not make this amplifier safe because, if the amplifier is plugged in and the power switch is off, and the interlock is intact and functioning properly, a fatal electric shock can be sustained by contacting several different components under the chassis of the amplifier. This situation exists with or without the HV-interlock. The ONLY way to be safe is: Never put your hands or a conductive tool inside any amplifier unless it is unplugged from the electric mains. Don't let the presence of the interlock lull you into believing that the interlock makes this amplifier safe because it does not. This is why I had no qualms about removing the interlock to make room for RFC1a. The choice is yours.

L7a

The 160 meter tank-circuit inductor (L7a) is wound on an Amidon T-225A-2 powdered-iron toroidal core. The bare toroidal core should be taped with one overlapping layer of Plymouth Rubber Co. silicone-rubber tape. If silicone-rubber tape is not available, use fiberglass tape. Vinyl tape is not recommended. The wire that is wound {16 turns} on the core is stranded, #14 AWG, Teflon insulated hook-up wire. This wire is ideal for RF applications. The first and last turns of the winding are held in place with nylon cable ties. The finished. inductor is fastened on a square of 1/8 inch acrylic sheet using long nylon cable ties. Holes are drilled in the acrylic sheet for the cable ties to pass through. The photograph illustrates this mounting method.

RF WIRING: The leads that carry RF circulating-current in the tank-circuit should be at least 1.6mm [#14] or larger diameter copper wire. To prevent stress on the metal-to-glass seals of the vacuum relays, a c.1 inch long, copper-ribbon link is inserted between a stiff wire and the relay terminal.

METER LAMP BRIGHTENING -- On 160m, the meter lamps typically get brighter from RF. To keep 160m RF out of the meter lamps, connect a 0.1uF - 0.15uF disc ceramic capacitor across the terminal strip near the meters.

----------------

End notes

1. silicone-varnish insulated wire is available in electric motor re-winding shops. It has much better RF and HV qualities than any of the "formvar" or nylon-clad types of insulation that were used in ancient times.

2. Parts Suppliers: New vacuum-relays and RF current-rated transmitting capacitors: Surcom [Jennings] %[619] 722 6162 - ask for Lenk or the order-desk. // New vacuum-relays: Kilovac [805] 684 -4560 - ask for Gail or the order-desk. // New-surplus and used vacuum-relays and RF current-rated transmitting capacitors: Alan Emerald , K6GA [714] 962 5940; Fair Radio Sales [419] 223 2196.

Parts Costs: Swap-meet vacuum-relays average about $35 each. New relays cost about $100 each. Swap-meet relays should be tested before purchase with two seriesed 9 volt batteries to power the coil and an ohm-meter to make sure that the contacts open and close properly. A current-limited HV breakdown-tester should be used to test the open-contact voltage capability of the relay. This measures the integrity of the vacuum inside the relay. If it is not convenient to do this at the swap-meet, be sure to get the call-sign and vehicle license-plate number of the seller and a moneyback guarantee if the vacuum proves to be bad.

The total parts cost of this project for a skilled scrounger is about $100. The cost of 100% new parts is about $300.

Literature: If you would like to obtain a copy of Eimac's good book Care And Feeding Of Power Grid Tubes, they can be reached at [415] 592 1221.

If you would like to discuss any part of this article with me, my telephone number is [805] 386 3734.

END

A Balanced Balanced Antenna Tuner

2026-08-20T05:20:32.298-07:00

A Balanced Balanced Antenna Tuner

By Richard L. Measures, AG6K

The published version of this article appeared in the February 1990 issue of QST Magazine. This is the pre copy-edited version, updated in May, 2003.

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JPG of a remote controlled, balanced L-network tuner built by Mike, WA1SEO.

Antenna tuners are like shovels. It takes more than one kind of shovel to be able to efficiently perform a variety of jobs. For example, a snow shovel isn't suitable for digging holes in hard ground. A tiling spade could be used to shovel snow, but it wouldn't be very efficient. Similarly, no single antenna tuner circuit can do everything extremely well.

A balanced load tuner should be designed, from the ground up, for the job that it is intended to perform. This article describes a circuit that does a good job of feeding an open-wire transmission line [ladderline]. It is not intended to be used for unbalanced loads such as a coaxial transmission line or for end-fed, Marconi or Hertz antennas - although it seems to be capable of doing so.

Now that we have nine amateur radio bands below 30MHz, an open-wire transmission line, center-fed wire, all-band antenna system looks even more attractive than it did when it first came into popular use in in the 1930s. In those days, we had only five bands below 30MHz. Taking advantage of this versatile antenna system requires a box that will interface the 50ohm unbalanced output of today's transceivers to the highly variable impedance [Z] of the balanced feed points of the all-band antenna .

Contemporary antenna tuner circuits claim to be able to operate into an unbalanced load or a balanced load such as ladderline. In actual use, most of the contemporary "matches everything, balanced or unbalanced" antenna tuner circuits produce a semi-balanced output when used with a balanced load. Although the antenna will radiate in this situation, a semi-balanced output is like having a semi-balanced checking account. It is less than wonderful.

A look at the diagram for the contemporary "matches everything" antenna tuner circuits reveals that they are usually unbalanced, high-pass filter characteristic, T-Network circuits with an add-on balancing device hooked to the output of the unbalanced tuner circuit. This is a compromise design which, not surprisingly, also has compromise performance when used with a balanced load.

The imbalance in these "balanced" tuners can be easily confirmed with a RF voltmeter or RF amperemeter(s). When the actual current or voltage is measured at each output terminal, the observed imbalance gets progressively worse above about 7MHz. At 28MHz, it is not uncommon to have 50 more current or voltage in one of the legs than in the other leg.

Some may ask: "why not use the balanced tuner design that was in vogue in the 1930s?" As many of you old-timers know, the 1930s-era balanced tuner consisted of a resonant or near-resonant, center link-coupled, tank-circuit with movable-clip taps on the secondary. For each band change, the clip taps had to be moved and re-optimized, the total inductance changed and the tuning capacitor re-tuned. Changing bands was a labor intensive job. These ancient tuners were seldom built in enclosures because near constant access to the clip taps and the inductor(s) was a necessity for changing frequency. It was a common practice to build these tuners on a breadboard for maximum accessibility. Hence the name breadboarding.

In the 1950s, the E. F. Johnson Company came out with their Matchbox® series of balanced antenna tuners. These tuners used the same center link coupling arrangement as the earlier tuners, but they eliminated the movable clip arrangement by using a double-differential capacitive voltage divider across the tank inductor. A differential capacitor is the RF-equivalent of a potentiometer / DC voltage-divider. This allowed the operator to electrically increase and decrease the voltage fed to the antenna, without changing taps. The Johnson circuit worked, but the Z-matching range was severely limited. Frequently, the SWR could not be reduced to a satisfactorily low level.

The balanced tuner that is described in this article has two, front panel adjustments, one optional, hi-Z / low-Z switch, and no clips. It uses the rarely seen, balanced version of the familiar, unbalanced, L-network. Changing bands is a piece of cake with this balanced tuner and the matching range can be made very wide by using enough L and C to handle the job.

The Trouble With Baluns

On paper, an unbalanced tuner, feeding a balun, connected to a ladder-line fed antenna should work well. In practice, it does not work well. The reason for this lies in the balun.

As a rule of thumb, a balun should have about 4 times as many reactive ohms as the resistive ohms of the load. This means that for use with a 600ohm balanced load, the balun should have a secondary winding reactance of about 2400ohm. For 80 meter operation, this works out to be more than 100µH of balun inductance! To create this much inductance on an appropriate MF/HF-rated [µ=40] ferrite core, an impracticably large number of turns of wire would be required.

The use of a balun, in a high-impedance circuit, inevitably creates two, very sticky problems: More turns means more ampere-turns of magnetic flux in the balun's core, and high magnetic flux densities can cause the ferrite-core to saturate. This distorts the RF waveform and creates harmonics. These harmonics extend well into the UHF TV band. The remaining problem with using many turns of wire is that doing so increases the winding-capacitance of the balun. The high capacitance of the winding creates unwanted reactance and/or balun imbalance. This is especially true with the commonly used 4 to 1 bifilar-wound balun, which does not have an evenly distributed winding capacitance like the trifilar-wound balun. When enough turns are placed on the 4 to 1, bifilar balun for satisfactory 80 meter operation, the inherent capacitive imbalance in the balun causes a progressively greater imbalance in the output voltage of the balun as the operating frequency increases. This imbalance causes a differential RF-current to flow through the ground wire on the tuner. The name "4 to 1 balun" is a misnomer. They are much better suited for broadband, unbalanced-to-unbalanced 4 to 1 transformer service such as would be needed in the input circuit for a grid-driven Class-AB1 amplifier, whose grid terminating resistance was 200ohm.

There is a problem with a substantial current flowing in the ground wire on any tuner: Since ALL conductors, no matter how wide, have inductive reactance, the RF current that flows through the ground wire or strap can develop a large RF-voltage on the tuner-end of the ground wire. With 1000W on 21MHz, 24MHz or 28MHz, the RF voltage on the "matches everything" tuner chassis can brilliantly light a neon-lamp, make sparks with a graphite pencil and burn fingers.

The 1 to 1, trifilar-wound balun solves the capacitive imbalance problem of the 4 to 1 balun. Unfortunately, it does not solve the problem of high capacitance in the windings themselves. And, more importantly, it does not solve the problem of core saturation due to the high magnetic flux-density created by the large number of turns required for any high-impedance balun.

The bottom-line is: high-impedance baluns are a very likely source of GRIEF no matter how carefully they are engineered and constructed.

All of these problems are easily avoided. The solution is simple: don't put the balun in the highest-impedance part of the circuit. Instead, put the balun in the lowest-impedance part of the circuit, and build a balanced L-network tuner for the balanced output of the low-impedance balun. So, why have we been putting the balun in the wrong part of the circuit for these many years? Good question. In most cases, the lowest-impedance part of the circuit is the 50ohm coax input to the antenna-tuner.

An Inexpensive, High-Performance, Ugly 50ohm-Balun

Building a no-grief 1.8MHz to 30MHz 50ohm-balun is easy. No costly ferrite-cores are needed, just a short length of 3 to 5 inch size plastic pipe, about 25 feet of 50ohm coax plus some nylon cable ties. Solid-dielectric coax is best for this application because foam-dielectric has a tendency to allow a change in the conductor to conductor spacing over a period of time if it is bent into a tight circle. This can eventually result in voltage breakdown of the internal insulation. The required length of the plastic pipe depends on the diameter and length of the coax used and the diameter of the pipe. For RG-213/U coax, about one foot of 5 inch size pipe is needed for a 1.8MHz to 30MHz balun. For 3.5MHz to 30MHz coverage, about 18 to 20 feet of coax is needed. This length of coax is also adequate for most applications on 1.8MHz. The number of turns is not critical because the inductance depends more on the length of the wire (coax) than on the number of turns, which will vary depending on the diameter of the plastic pipe that is used. The coax is single-layer close-wound on the plastic pipe. The first and last turns of the coax are secured to the plastic pipe with nylon cable ties passed through small holes drilled in the plastic pipe. The coil winding must not be placed against a conductor. The name of this simple but effective device is a choke-balun.

Some people build choke-baluns, without a plastic coil-form, by scramble-winding the coax into a coil and taping it together. The problem with scramble-winding is that the first and last turns of the coax may touch each other. This creates two complications. The distributed-capacitance of the balun is increased and the RF-lossy vinyl jacket of the coax is subjected to a high RF-voltage. The single-layer winding on the plastic coil-form construction method solves these problems since it divides the RF-voltage and capacitance evenly across each turn of the balun.

A more compact, less ugly, 1 to 1 impedance-ratio, 50ohm trifilar-wound (with wire) ferrite-core balun could also be used but there would be some tradeoffs. Ferrite cores are not cheap. Also, the air-core of the coax-balun can't saturate like the ferrite-core and, unlike ferrite-core wire-wound baluns, single-layer wound coax-baluns almost never have an insulation breakdown problem. Also, a trifilar-wound balun does not like to work into anything but a perfectly balanced load. With an imperfectly balanced load, the coax-balun will not, as does the trifilar balun, generate a differential, third RF-current on the outside of the coax that brings the RF to the input of the tuner. The choke-balun is not fussy. It will work as well into a less than perfectly balanced load as it will into a perfectly balanced load, and do so without the possibility of creating a differential RF-current on the station ground and fricasseeing the operator's fingers.

The Versatile L-Network

The L-network is THE basic RF-resistance or reactance plus resistance (impedance) [Z] transformation tool. It is also used to build high-pass, low-pass and band-pass filters. The imbalanced L-network forms the basis for all of the antenna tuners that are currently being sold. All contemporary amplifiers use two, or more, L-networks in series to match the high, anode (plate)-resistance of the amplifier-tube to the low-resistance of the coaxial output. Most amplifiers use (2) L-networks in series, which is more commonly known as a Pi ( Pi )-network. A few amplifiers use (3) L-networks in series, which is called a Pi -L-network. "No tune-up/broadband" amplifiers may use 5 (or more) fixed L-networks in series for each bandswitch position. More L-networks means more harmonic-suppression in the output of the amplifier. The complex-appearing Butterworth and Chebyshev filters are nothing more than a series of basic L-networks.

When used for matching resistive loads, an L-network consists of one capacitor and one inductor. When the L-network is used for matching loads that contain both resistance and reactance (Rohm +/- jXohm = Z), the reactance of the load may partially or sometimes completely replace one of the reactances in the L-network. Thus, in rare cases, it is possible to build an L-network with only one component, but only for a specific frequency and load Z. In some situations, cancelling the load reactance will require the use of a larger reactive component in the L-network. In more extreme situations, the load may be so reactive that the L-network must be made from two capacitors or two inductors!

There are four ways to connect the capacitor and the inductor in an L-network. See Figure 1A. When inductance is used for the series-reactance and capacitance is used for the shunt-reactance, the L-network acts as a low-pass filter as well as a resistance matching device. When capacitance is used for the series-reactance and inductance is used for the shunt-reactance, the L-network acts as a high-pass filter and a resistance-transformation device.

The resistance-matching range of the L-network is remarkably wide. It can match 50ohm to a 1ohm or to a 10Kohm load with ease and good efficiency provided that a reasonably high-Q inductor is used.

When the L-network is used for stepping up the input R, the shunt-reactance is placed across the load. For stepping down the input R, the shunt-reactance is placed across the input of the L-network. Another way to look at it is: The shunt-reactance is always connected across the highest-resistance side of the L-network. This means that, for a 50ohm input, wide-range tuner, which will match loads of more than 50ohm and less of than 50ohm, a step-up/step-down [Z] switch must be provided so that the shunt-reactance can be switched to the input side for <50ohm loads and to the output side for >50ohm loads. The same result could be accomplished by reversing the input and output connections. The switch saves time.

The T-network eliminates the need for the step-up/step-down switch by using a clever tool from AC circuit-analysis. This tool is based on the fact that, for every R-X series circuit, an equivalent R-X parallel-circuit may be calculated and substituted for the series-circuit. The equivalent circuit will act exactly the same as the original circuit. This also works in reverse.

See Figure 1, B. An additional capacitor or inductor is placed in series with the load of a resistance step-up L-network which will not normally match a load resistance that is lower than the input resistance (usually 50ohm). If the added series capacitor or inductor is adjusted so that it has a sufficiently high reactance, the resistive component of the R-X series load's parallel-equivalent circuit will be above 50ohm and the step-up L-network will be able to match the load.

For example: Given: a Z step-up, series-L / shunt-C {low-pass filter characteristic}, L-network, that will only match load resistances that are greater than 50ohms, is connected to a 1ohm load. Problem: Obtain a Z-match.

One solution is to add a +j10ohm (inductive) reactance in series with the 1ohm load { Z = 1ohm + j10ohm} This series RX circuit is electrically equivalent to a parallel-circuit of R=101ohm in parallel with an inductor whose reactance is plus j10.1ohm. Since a load resistance of 101ohm is above 50ohm, a match could be achieved if minus j10.1ohm is added to the L-network's shunt capacitor in order to cancel the parallel equivalent circuit's +j10.1ohm.

Adding more capacitive reactance to a variable capacitor is easy: simply adjust the capacitance to a lower value. This is clearly a case where less [C] gives more [ohms]!

The Balanced L-Network

The L-networks shown in figures 1A and 1B is designed to work with an unbalanced input and an unbalanced load. The balanced L-network is similar to an unbalanced L-network. The difference is that the balanced L-network's series-reactance is divided into two equal parts and both ends of the shunt-reactance must be well insulated from ground instead of one end being grounded. See Figure 1, C. The balanced L-network must be fed from a balanced source. The same formulas are used for either network.

Mechanical Considerations For Balanced L-Networks:

The tuning shafts of the variable-inductors and the variable-capacitor are hot with RF voltage and must be insulated from each other and from the outside world. Also, the insulated frame of the variable-capacitor should be kept well away from any grounded surface. This requirement is much easier to meet if the balanced tuner is built in a polyurethane-varnished plywood-box, instead of in a metal box. The (RF-hot) frame of a single-section variable capacitor should be elevated on standoff-insulators. This helps to keep the circuit capacitively balanced.

If a split-stator variable capacitor is used, it won't be necessary to insulate its tuning shaft for high-voltage RF, or to keep the capacitor well away from ground. However, an insulated tuning knob with a well recessed set-screw, instead of an all-metal tuning knob, might prevent an unpleasant surprise to the operator's fingers if the split-stator capacitor proves to be less than perfectly balanced.

The variable-inductors must have equal inductances and be driven in synchronization by one tuning-shaft. It is possible to end to end-couple the two variable-inductors with an insulated coupling that can handle minor, axial, shaft misalignment, but this does not result in good electrical symmetry or optimum inductive de-coupling between the two variable-inductors unless a shaft extension is used. Good symmetry is probably a moot point for 80 meter operation but it is a consideration on the higher-frequency bands. If you decide to use end-to-end coupling, there is a material, which is available in stores that carry drip-irrigation materials, which is ideal for end-coupling RF-components. The material is called clear (actually translucent) 1/4 inch size,polyethylene drip-tubing. It is semi-rigid and solves the axial shaft alignment problem and RF insulation problems nicely. It is a very-tight fit over 1/4 inch shafts. It can be held in place with �5/16 inch size flat spring-clamps.

Another method of coupling the variable-inductors is to use a 3/8-inch plastic timing-belt and two plastic timing-belt pulleys like the type used on xerographic copiers. This allows the variable-inductors to be placed side-by-side which results in better layout symmetry. One single-flange timing-pulley and one double-flange timing-pulley should be used so that the belt can not slip off. Small flats can be ground into the variable-inductor's tuning shafts so that the pulley set-screws will stay put.

The ends of variable-inductors that have a coil-end contact that is electrically connected to their tuning-shafts should be connected to the lower-voltage, input side, of the balanced tuner. This minimizes the RF-voltage stress on the insulated parts that synchronize and drive the variable-inductors. The roller-contact must be shorted, across the un-used turns of the inductor, to the low-voltage end of a variable-inductor. This is done to stop the Tesla-coil transformer-effect which can cause spectacular HV RF-arcing at some L settings. The courser-turns-pitch ends of variable-turns-pitch variable-inductors is placed at the (higher voltage) output side, of the tuner.

Sometimes it is more convenient to put the balanced L-network antenna tuner in a remote location so that the ladderline does not need to be brought through the wall of the house. A simplified diagram of a remote-controlled, permanent-magnet {reversible}, DC motor-driven tuner is shown in Figure 2. This simplified diagram does not show the detailed wiring of the control cable and the remote indicator/control-box/power supply.

Limitations: The balanced L-network that is illustrated is designed to work with balanced or semi-balanced loads that have a resistance greater than 50ohm. The vast majority of open-wire transmission-line fed antennas fit into this category. Rarely, it is possible to have a situation where an open-wire transmission-line fed antenna would have a resistance of less than 50ohm. This would be the case with a half-wavelength dipole, less than one-quarter wavelength above ground, that is fed with a transmission line that contains an even-number of quarter-wavelengths.

If a load resistance of less than 50ohm is to be successfully matched, the variable-capacitor must be switched to the input side of the variable-inductors, or, as in a T-network, a matched-pair of appropriate reactances can be inserted in series with the load to obtain a match.

Since the actual feedpoint Z at the bottom of a ladder-line fed, multi-band antenna, can be almost anything imaginable, it is probably prudent to include a DPDT Z-step-up / Z-step-down switch in the design of a balanced L-network antenna tuner.

Calculating The Series And Shunt -Reactances

The formulas for calculating the total series-reactance [Xser] {in +/- j ohms} and the shunt-reactance [Xsh] {in +/- j ohms} are shown in Figure 3. These ohmic values can be converted into actual values of capacitance and inductance for a specific frequency by using the two formulas at the bottom of Figure 3. Since most hams do not have access to an RF Z-Bridge, the formulas are not widely used.

The reactance formulas give exact values only for non-reactive, purely-resistive loads. If the reactances of the L-network are adjustable, a wide range of load reactances can be cancelled by adjusting the L-network to create an equal and opposite reactance. This is accomplished by tuning the L-network for zero reflected power while using a minimum power level.

The shunt and series-reactances, in ohms, that are found with the formulas can be either inductive or capacitive, but they must always be opposites for resistive loads. If a high-pass filter characteristic is desired, as is frequently the case on 160 meters where a strong, local, broadcast-station would otherwise cause receiver overload problems, the series-reactance is capacitive and the shunt-reactance is inductive.

If an L-network is to be used on just one band, only one of the two reactances usually needs to be variable in order to obtain a low-SWR over the entire band. A good use for this technique is with a half-wave antenna for the 160 meter or 80 meter band. This antenna would otherwise have a low-SWR at the middle of the band and a �5 to 1 SWR at the band edges. With a single variable component L-network tuner, the SWR will usually be less than 1.1 to 1 at the band edges. Of course, if both components are made variable, the SWR will not exceed 1 to 1.

If a band-pass filter characteristic is needed for a specific single-band operation, a high-pass L-network can be coupled to a low-pass L-network with each network contributing about 1/2 of the total resistance transformation. When this is done, only one of the 2-stage L-network's four reactances usually needs to made variable to allow covering the entire band with a low-SWR. I use this circuit in a 160m tuner. It keeps the local broadcast stations from overloading my receiver, and it attenuates the second harmonic, on 80m, to -63db.

Selecting a Suitable Peak Voltage-Rating For The Capacitor:

The RF-peak-voltage that appears across the capacitor varies widely depending on the feed-line Z and the power level used. The voltage can be calculated if the impedance and power at the feed-point are known. The formula for calculating this is E= [P x Z]^0.5 . Since P is measured in RMS-watts, the AC voltage (E) result will be in RMS-volts which can be converted to peak-volts (e) by multiplying the result by 2^0.5. For example, if Z=600ohm and P=1500W, e= [1500W x 600ohm] *2^0.5 =1341v peak.

When using a vacuum variable-capacitor, it is important to realize that the maximum RF peak-voltage rating at 30MHz is usually 60% of the rated DC / 60Hz peak-voltage rating. Thus, a "5000V" rating translates to a 3000 peak RF-volt rating.

Used or new-surplus vacuum-variable capacitors should be tested before use because they may have developed a small air-leak that renders them useless.

Constructing Ladder-Line:

Commercial open-wire transmission-lines seem to use #18 gauge copper wire exclusively. Although #18 gauge copper wire is completely satisfactory for use in transmission-lines that do not carry a high SWR, it is not adequate for the combination of high-power and very high-SWR that is the norm for multi-band-use antenna systems.

Number 14 gauge solid-copper wire is better suited for this application. High-quality, #14 gauge, 2-wire transmission-line does not seem to be commercially available so it must be constructed by the user.

For use in an open-wire transmission-line, solid-wire is better than stranded-wire because it will not twist as easily as stranded-wire and short-out. The exception is the top few inches of the transmission-line, at the center-insulator, which should be fine-stranded for flexibility. Copper-weld wire is not only dangerous to eyes when it is being handled and cut, it is too springy and unruly for use in a transmission-line. It also prematurely becomes brittle due to normal wind movement. Number 14 gauge, TW, single-conductor solid-copper house wire is commonly available in 500 foot rolls where building materials are sold. It affords a good compromise between strength and stiffness. The insulation can be removed by fastening one end of the wire to a stationary object and horizontally pulling a sharp knife between the copper and the insulation of the stretched out wire. TW wire can also be used for the antenna itself if some support is provided to hold up the center-insulator and transmission-line of the antenna.

High RF-quality, lightweight, long-lived, inexpensive and easy to fasten feedline insulators can be made from ABS thermoplastic. If you can find it, 3/8-inch ABS rod-stock or, as a second choice, 3/8 inch ABS square-stock works well. Rod-stock has the advantage of having less wind resistance. These materials can be found or at some of the larger plastic-supply houses. If you have a color choice, black is usually the most UV-resistant color. If you can't find the rod or square stock, the insulators can also be made from commonly available ABS plumbing pipe.

ABS-pipe must first be heated and flattened. This is how: Cut the ABS pipe length-ways into halves or thirds with a table-saw. The lengths should be about the same as the width of a Teflon-coated cookie-sheet that will fit into your oven. Since ABS is a thermoplastic that melts at �150ºC, the oven temperature should be set to �180ºC [350ºF]. The 1/2 or 1/3 round sections of the pipe are baked, concave-side down, on the cookie-sheet until they begin to soften. The oven is opened and a sheet of plywood, weighted with a brick, is placed on top of the pipe sections. When the ABS is flat, it is set aside to cool-down.

ABS sheet is cut, on a table-saw, into strips about 3/8-inch wide and 3 inches to 6 inches long. Or, the rod-stock is cut into appropriate lengths. The ends of these insulators are notched to a depth of �1/4 inch with a hack-saw or band-saw whose saw-cut width is less than the width of the #14 gauge solid copper wire. The lengths of the insulators is not at all critical since the characteristic Z of the ladder-line will change only slightly between 3 inches and 6 inches spacing. The main consideration is that the wires do not short-out.

The space between insulators is normally about 5 to 10 times their length. For example, 6 inch length insulators would normally be spaced about 30 to 60 inches apart. No more insulators need to be used than are necessary to prevent feedline twisting and shorting-out.

The ABS insulators can be securely fastened to the bare copper wires thusly: Clamp both parallel wires, spaced appropriately, into a vise. Stretch the wires out straight and fasten the other end of the wires to a post. With a flame from a propane-torch, heat the wire at the place where an ABS-insulator is to be fastened. When the wire is hot enough to melt the ABS, press the wire into the notch on the insulator. The heated wire will melt its way to the bottom of the notch in the ABS. The wire should be held in this position for about 15 seconds while the thermoplastic cools-down and re-hardens, which traps the wire. A damp rag can be used to speed up the cool-down time.

Notes: This operation can be done with two hands but it is much easier and faster with four hands. The wire must not be too hot or it will cause the thermoplastic to decompose and/or ignite.

The useful life of ladder-line in windy areas can be extended if a Dacron-cord or braided Dacron fishing line tether {or tethers} is fastened to an insulator about half-way up the ladder-line. The tether is pulled sideways to form an angle of �45º and fastened to a stationary object. The tether prevents the feedline from whipping around in the wind which would otherwise eventually cause the wires to break. Two or three tethers, fastened to the same feedline insulator, spread about 120º apart work better than one.

Parts

A source of variable-capacitors and variable-inductors is Cardwell-Multronics Corp. (631) 957-7200

"http://www.cardwellcondenser.com/index.html"http://www.cardwellcondenser.com/index.html

They manufacture the old, E. F. Johnson line of 5 ampere [229 series] and 10 ampere [226 series] variable-inductors. Their P/N for an 18µH, 5a, variable-inductor is 229-0202-1. The 28µH version's P/N is 229-0203-1. These P/Ns are the same as E. F. Johnson's old P/Ns.

Another source for variable-inductors is occasionally Fair Radio Sales, 419 227 6573. Fair Radio also has a few sizes of variable capacitors and a ceramic, HV, RF-switch that is capable of switching the variable-capacitor on the balanced L-network between the output of the variable-inductors, for high-resistance loads, and the input of the variable-inductors, for low-resistance loads. The P/N of this switch is 3Z9626 and the price is $2.50.

MAINTENANCE NOTE: Variable-inductor roller-bars and other sliding RF-contact surfaces should be routinely wiped clean with a lint-free cloth and re-lubricated about once a year. A suitable lubricant is GC Electronics Tunerlub, "high-frequency lubricant", Catalog #26-01. This material must be applied thinly with a lint-free cloth. More is not better.

Source of cogged belts and pulleys: Small Parts, Inc.,13980 NW 58 Court, Miami Lakes, FL 33014, telephone [305] 751 0856

The part numbers [p/n] for the 40-tooth pulleys are TBPN - 40S [single flange] and TBPN - 40/D.[double flange]. At least one of the two pulleys should be double flanged to prevent the belt from slipping off during use. The outside diameter of 40-t pulleys is 2.78". For 30-tooth pulleys, change the number after the dash from '40' to '30'. The outside diameter of a 30-t pulley is 2.14". Either 30-t or 40-t pulleys may be used but the belt does not have to bend as sharply on 40-t tooth pulleys.

The belt length equals the desired distance between the two roller coil drive shafts x 2, plus the circumference of one pulley. The belt-contact-surface circumference of the 40-t pulleys is 40/5=8" and the circumference of the 30t pulleys is 30/5=6". Thus, if the coils will be mounted 5" apart [shaft-shaft] and 40-t pulleys are used, the belt-length needed would be 5" x 2 = 10" plus 8" = 18" [p/n TB6-180]. The belt-length part of the p/n is the 2nd and 3rd digits from the right. Belt lengths from 6" to 26" are available in 1" increments.

The price of an 18" belt is $5.50 in my old Catalogue #10. The price of 40-t pulleys is under $7.00 each. The price of 30-t pulleys is under $6.00 each. I assume that the prices have probably increased.

Acknowledgements:

Congrats to Charles, W7XC, who provided valuable input during the editing of this article.

--------------

If you would like to discuss this article with me, I can usually be reached at.

AG6K SB-220 Amplifier Circuit Improvement

2026-08-20T05:15:35.164-07:00

Updated 29 August, 2001

The Heathkit SB-220, two 3-500Z, amplifier made a notable impact in the world of amateur radio. It was the first, reasonably priced, intelligently designed, SSB-CW, HF-amplifier that was sold to the amateur radio community. Unfortunately, this amplifier kit is no longer being manufactured.

The SB-220 has some excellent design features and a few design weaknesses that are easily corrected. In this article, I will discuss both topics and the cures for its weaknesses.

HV Power Supply

Before the arrival of the SB-220, there was a popular notion that a legal-limit SSB amplifier needed a heavy-duty power-supply that required two grown men to move it about. Heath engineers knew that this idea was based more on amateur radio folklore than on sound electrical engineering principles. They also knew that the average duty-cycle of a human voice was only about 15% when no carrier was present, as is the case for SSB operation. So, why build a 100% duty-cycle, AM, "lock-to-talk" power supply when one was not required? Thus, they designed a power-supply that would competently do the job that was needed. This resulted in a considerable size, weight and cost savings, which they happily passed along to the buyers of their product. At first, some people in the ham community had negative comments about the SB-220's "wimpy" power supply. With the passage of on-the-air time, it became apparent that the power supply would do the job, and do so with a low failure-rate and with no detectable on-the-air ripple. This was no accident. Heath engineers had wisely specified a HV-transformer design that had an exceptionally low secondary-resistance of only 12.2-ohms. This minimized the voltage drop under full load in the capacitor-filter fullwave-voltage-doubler rectifier circuit. Such circuits have an extremely high peak-current to average output-current ratio. So, minimizing the transformer winding resistance is essential for good voltage regulation and to minimize the I²R heat loss in the transformer's windings.

Many hams initially labeled as "inferior" the capacitor-filter fullwave-voltage-doubler rectifier circuit. They did not realize that this circuit has some advantages over the traditional, fullwave-bridge rectifier circuit. These advantages are:

1. Low ripple-voltage. This is due to the fact that, as one capacitor bank is charging, the other capacitor bank is simultaneously discharging, thus, cancelling each other's 180º out-of-phase, sawtooth waveforms.

2. It allows the transformer to have only half as many secondary turns which yields a more efficient transformer design. Here's why: Since a layer of insulating-paper is required between each layer of wires, fewer turns means fewer layers of paper. This allows the transformer to use less paper and more copper. The net result is a transformer that has a high ratio of copper to paper. This makes for a very capable transformer.

3. Excellent voltage-regulation during current-transients, due to the fact that no swinging-inductance filter is used. This is exactly what's needed for CW and SSB modes of operation.

Cooling

Since about half of the power that is consumed by a linear amplifier is converted into heat, another important amplifier design consideration is cooling. Most of the heat that a 3-500Z, or any other internal-anode tube, dissipates is carried away by radiation from its anode.

Here's how it works: during normal operation, the anode becomes so hot that it glows a bright orange color. The surrounding objects are relatively much cooler, so the anode looses heat to the surrounding objects by radiation-cooling. Unfortunately, the anode also heats up other parts of the 3-500Z that are sensitive to heat. This includes the critical glass-to-metal seals and the solder that is used to fasten the pins to the base of the tube. These heat-sensitive parts must be cooled with forced airflow.

Heath's engineers came up with a deceptively simple method of effectively cooling the 3-500Zs. They realized that the conventional, uncheap, air-system-socket / glass chimney cooling method had some serious trade-offs, such as: 1. It was difficult to force enough air through the restrictions to airflow in the "air-system" to adequately cool the filament-pins and filament-seals of the tubes. 2. The horizontal fins on the standard anode [plate]-cap coolers were obviously not designed to be cooled by the vertical airflow from the air-system chimney. 3. The restrictions to airflow with the air-system cooling method dictated that a high-pressure centrifugal blower be used. All high-pressure blowers have one thing in common: High Acoustic Noise.

What was needed was a cooling system that would quietly move high-velocity air past the hot filament-pins [1], filament-seals, the anode-seals, and the glass envelopes.

The Heath engineers knew that when horizontal air flows across vertical cylinders, such as a 3-500Z envelope and its pins, the airflow will follow the curve of the cylinders. This provides fairly uniform cooling to all areas of the cylinders, eliminating hot spots. They concluded that, with horizontal airflow, the cooling air would have a direct path the heat-sensitive parts of the tube. Horizontal airflow would also allow the anode-cap cooler's fins to lineup properly with the flow of cooling air, allowing it to function efficiently.

Since the filament-pins are below the chassis, and the filament-seals and anode-seals are above the chassis, the Heath engineers used an open-ended chassis so that a single, 15cm [6"] diameter fan blade could simultaneously blow cooling air above and below the chassis.

To position the four, hot filament-pins optimally in the under-chassis airflow, the pair of tube-sockets were mounted with the two pairs of filament-pins (#1 and #5) facing each other. This positioned the hot parts in front of the tips of the fan blades where they protruded under the chassis.

The cooling system design was brilliantly simple. It was relatively quiet and it worked well. Reports of 3-500Z-pin solder melting in SB-220 amplifiers are very rare - with the exception of cases where the fan motor bearings eventually seized because they were never oiled!

On the other hand, I have heard of many 3-500Z-pin solder melting episodes in other amplifiers that used air-system chimney/centrifugal blower cooling.

One weakness in the SB-220's cooling system was the failure to reduce the heat-reflectance of the bright aluminum surfaces that were adjacent and parallel to the anodes. This would have reduced the radiant heat that was reflected back at the hot anodes. The best color for this purpose is black. [2]

This deficiency is easily corrected: After the tubes have been removed, black, liquid shoe-polish can be applied to the vertical, aluminum surfaces, that are near the anodes. The built-in felt applicator, that is attached to the bottle cap, works well for applying the black coating.

One weakness in early models was the failure to provide oil-holes for the fan motor bearings. This problem can be corrected by drilling a small hole, no more than 1/4" [3mm] deep, above the front and above the rear oilite bearings. [3] The fan should be lubricated yearly with a thin, non-gumming [4] oil such as Hoppe's 1003. [5] / Two or three drops of oil needs to be inserted in each hole. What isn't absorbed into the felt wicks, that surround the oilite bearings, runs out the bottom. More oil is not better, just messier. The easiest way to get the desired amount of oil in the holes is to apply the oil with a disposable insulin syringe. Each "unit" on the syringe equals about one drop of oil.

Premature Filter-Capacitor Failure

Aluminum electrolytic filter-capacitors are very sensitive to heat. For every 10ºC increase above room temperature, the life expectancy of the capacitor is approximately cut in half.

The electrolytic filter-capacitors in the SB-220 are subjected to a high degree of heat during normal operation. The major source of this heat is from the eight, 30k-ohms, voltage equalizer resistors that are adjacent to the eight filter-capacitors. During transmit, a minor source of capacitor heating, results from the 60Hz ripple-current flowing through the capacitor's internal resistance.

The capacitor heating problem is compounded because cooling air does not reach the capacitors due to the molded plastic capacitor holders that surround and insulate each capacitor. In some cases, the heat present will partially melt the ends of the capacitor holders that are nearest to the 30k-ohms resistors.

The heat that is dissipated can be reduced by c.70% if 100k-ohms, 3W, metal oxide film resistors are used to replace the original 30k-ohms resistors. This simple modification will prolong the life of the (8) electrolytic filter-capacitors.

Other resistance values may be used, up to roughly 150k-ohms, provided that the resistors can withstand the voltage that is applied to them and the resistances are within ±5% of each other. I do not recommend using ancient-type 2W carbon-composition resistors for this application. They do not maintain their rated resistance tolerance with age.

Note: Increasing the equalizer resistance will also increase the capacitor bleed-down time after the amplifier is shut off.

Since this amplifier has a shorting HV-interlock, that grounds the HV positive when the perforated cover is removed, it is advisable to wait until the voltmeter indicates nearly 0V before allowing the interlock to short the HV to the chassis.

Here's why: When the HV-positive is shorted to ground, the stored energy in the filter-capacitors is applied directly to the grid-current meter shunt resistor, R3 {0.82-ohms}, which is the only HV-negative path to chassis. The peak discharge current can be substantial.

For example, if the filter-capacitors are at the 100V level when the interlock shorts, the peak-current through R3 is: I c. E/R c. 100V/0.82-ohms c.100A. If a substantial voltage exists in the filter-capacitors when the interlock shorts, R3 can be literally blown away by the discharge current-pulse. If the multimeter happens to be in the grid-current position, the meter can also be crispy-crittered.[6]

It is for this reason that I removed the interlocks from both of my Heathkit amplifiers. Another consideration was that the interlock does not prevent the operator from contacting the potentially fatal voltage from the electric-mains when the amplifier is plugged in and switched off. The interlock will not prevent the operator from being electrocuted. In other words, the safety-interlock does not make the amplifier safe.

Another advantage of removing the interlock is that it allows the perforated cover to be removed while optimizing the tuned-input circuits. This allows better access to the tuned input circuits.

>>>> There is no safe substitute for pulling the electric-mains plug before putting fingers inside any amplifier.

Intermittent Meter Readings

There are at least two problems that can cause intermittent meter readings in the SB-220. If this occurs only on the voltmeter, the most likely problem is with the (3) 4.7M-ohms, 1W voltmeter multiplier resistors [R6, R7, and R8]. These resistors, which are rated at 350v maximum per-unit, are subjected to c.1000V-per-unit in the Heathkit circuit. This can lead to resistor deterioration, which causes fluctuations and/or inaccuracies in the 0-3500V meter indication. [7] The abused resistors can be replaced with modern, 2W flameproof spiral-film resistors that were designed to handle this voltage.

The other source of trouble lies inside the meters. Here's why: Different metals are used for the various parts of the meter. These parts, which conduct current to the meter armature, are fastened together with screws. Over a prolonged period of time, moisture in the air causes electrolysis to take place at the junctions of the dissimilar metals. This increases the resistance at the junctions, which causes intermittent meter indications.

This problem can be corrected by prying off the meter face, carefully removing the meter scale, and applying small dabs of conductive-paint to all of the dissimilar metal junctions that carry current to the armature.

The conductive-paint can be thinned with acetone to facilitate penetration into the narrow areas between the parts. The conductive paint should be allowed to dry around 15-minutes before the plastic meter faces are replaced.

Transceiver Relay Contact Failure

During receive, the voltage across the Antenna Relay Jack rises to about +115V. A bypass-capacitor, C52, is connected in parallel with this jack, so the capacitor charges up to 115V during receive. During transmit, the transceiver's relay (if one is used) places a short-circuit across this jack and the fully-charged C52. The SB-220 relay coil current is only about 0.025A, but the peak discharge current, produced by placing a direct short on the charged capacitor, can be surprisingly large. This action is like an electric spotwelder. In time, the contacts in the transceiver relay can become pitted and fail to make contact, or become spotwelded together, causing the amplifier to key-up continuously.

This problem can be corrected by placing a 100-ohm to 200-ohm, 1/2W resistor [to limit I] in series with the center pin [blue wire] on the Antenna Relay Jack. C52 must be connected to the blue-wire end of the resistor. See Diagram 1.

The voltage lost in this resistor will c. 5V, which is insignificant to the 110VDC relay coil.

Filament Circuit

The most popular, published modification for the SB-220 has been filament inrush-current limiting. It is true that a large number of 3-500Zs in SB-220s suffered from filament-to-grid shorts, so some people began to theorize that excessive filament inrush current was the villain.

Another theory was that the filament-to-grid shorts were caused by a "manufacturing defect". Neither theory turned out to be true, even though Eimac®, as usual, generously made good on the tube warranties.

Curiously, none of the authors, who wrote the SB-220 inrush-current limiting articles, ever bothered to measure the actual filament inrush-current. So, I decided to measure it with my trusty old HP-1706A oscilloscope. After all, my name is Measures, so why not?

Here's what I found: The maximum, peak inrush-current through the 3-500Z filaments in an SB-220 is only 60% of what Eimac® allows. This esoteric feat was accomplished by the use of a special current-limiting core in the Heathkit filament-transformer. The core is similar to the cores used in current-limiting neon-sign transformers. Externally, this core appears to be substantially different than the core used in the HV-transformer.

About a year after I measured the inrush-current, the fact that the SB-220 used a current-limiting filament-transformer, was mentioned in QST Magazine.

The actual cause of virtually all grid-to-filament shorts in the 3-500Zs was later discovered to be a very brief, and usually very noisy, VHF parasitic-oscillation at roughly 110MHz. As will be discussed later, the large grid-current-pulse, that accompanies the parasite, was creating a large magnetic-pulse inside the 3-500Zs, which pulled the hot filament-wires off-center, causing them to touch the grid wire cage.

Another interesting feature concerning the SB-220 filament circuit is the fact that they normally use a filament-voltage at the low end of the recommended voltage range; typically about 4.85V. This may not seem important, but, according to Eimac®, each 3% reduction in filament-voltage, provided that no drop in PEP output occurs, will double the life expectancy of the amplifier-tube. Thus, the amplifier-tubes in a SB-220 can be expected to last at least 4-times longer than the tubes in some other, much more costly and prestigious brands of two 3-500Z amplifiers.

Just because something costs more does not automatically mean that it is better.8

For example, the contemporary, c.4db more expensive, Henry Radio Co. 2KD-5 was generally considered to be a better designed, higher power output, and more rugged, amplifier than the SB-220, This amplifier typically had a filament-voltage of more than 5.90V with a line voltage of 240V. 5.90 volts clearly exceeds the maximum filament-voltage rating of 5.25V and reduces the useful emission life of the two 3-500Zs to only a few-percent of what could have been realized if the same tubes had been used in an SB-220.

Another example is the Trio-Kenwood TL-922. In this amplifier, about 5.3V is applied to the 3-500Z filaments when the amplifier is operated from 120V or 240V. This reduces the useful-emission life of the tubes by more than 75%. Of course, if your electric-mains voltage is 216V, the filament-voltage will be perfect.

Obviously, the filament-circuit in the SB-220 needs no stepstart circuit to limit inrush-current, but there is still a good reason to add a stepstart circuit to the SB-220.

Here's why: If the amplifier is started up when powered from a stiff, 240V source, the inrush-current, through the Power Switch and other components, is considerable. A stepstart circuit will eliminate this potential source of trouble.

An easy-to-build, stepstart circuit is shown in Diagram 2. In this circuit, the stepstart relay can close only when the filter-capacitors in the +110V and +3000V power supplies have charged up to about 2/3 of their normal voltages. If the stepstart relay closes before the HV reaches 2/3 of its normal potential, the 2k-ohms nominal-value resistance should be increased. If the relay has trouble closing, the resistance should be decreased, which will increase the current through the relay's coil. If the circuit is functioning properly, the stepstart relay will close in about one-second, as the voltmeter passes the 2000V level. The amplifier may be operated at full-throttle 1-second after the relay closes.

The two, 20-ohms, 10W, resistors and the stepstart relay can be glued directly to the bottom of the chassis, directly under the filter-capacitor bank, using silicone-rubber adhesive. [9] The resistors should be held up off of the chassis a few mm, by the silicone-rubber. This mounting method is appropriate because drilling mounting holes in this area could harm the filter-capacitors.

Since the stepstart relay adds to the current burden on the +110V power supply, it is probably a good idea to replace the stock, halfwave rectifier [D16] with a full-wave-bridge rectifier unit. Note: The grounded red-wire, on the transformer's 80VRMS winding, must be ungrounded and connected to the input of the full-wave-bridge rectifier.

Adding A Standby-Switch

Another popular modification for the SB-220 is the addition of a standby-switch. A standby switch is really not necessary in this amplifier because it uses instant-on type tubes and a current-limiting filament-transformer.

This transformer is gentle to the filaments, so the amplifier may be switched on or off as often as you like with no problem. The exception is: if you have just made a long RTTY or FM transmission, the glass-to-metal seals in the 3-500Zs should be allowed to cool down for about 1 minute before switching off .

Silicon Rectifier "Protection"

In the early 1960s, silicon-rectifier manufacturing technology was hit and miss. There was considerable variation between individual rectifiers of the same type. The variation between rectifiers led designers to use resistor-capacitor equalizer circuits in parallel with series rectifiers.

Today, silicon-rectifier manufacturing technology has improved, so that same-type rectifiers are very uniform in their parameters. Modern silicon rectifiers do not need to be equalized. Unfortunately, old habits die a slow death and many hams are still clinging onto 1960s design techniques.

Much has been written about adding "equalizing" resistor-capacitor protection networks across the rectifiers in the SB-220's HV power supply. Unfortunately, these "protection" circuits not only do not perform as advertised, they can lead to premature rectifier failure.

Here's why: The 1/2W resistors that are typically used are rated at 250V maximum. How can a 250V resistor be trusted across a 600V or a 1000V rectifier? If anything breaks down in a series-rectifier circuit, it is like dominoes falling. One resistor failure can wipe out the remaining good parts in the circuit.

The most frequent failure mode for HV power supply rectifiers is too much reverse-current. This problem can be eliminated if the total PIV capability of the series connected rectifiers substantially exceeds the peak voltage in the circuit.

In any series circuit, the current in all of the elements is exactly equal. The rectifiers are all in series. So, the reverse-current burden is exactly the same for each rectifier unit. How is it that something which is already exactly equal needs to be "equalized"?

During the half-cycle application of reverse-voltage, it is important that all of the rectifiers in a series leg have similar junction capacitances. If they don't, then the reverse-voltage across the lower capacitance rectifiers will be excessive. Here's why: in a series circuit, smaller capacitors charge-up faster, and to a higher voltage, than larger capacitors.

Approximately 10nF [0.01µF or 10,000pF] of bypass capacitance across each rectifier is probably a good idea if, for example, 1A rectifiers are placed in series with 6A rectifiers. This is the case because of the wide difference in junction capacitance between 6A and 1A rectifiers.

If all of the rectifiers in a series leg are similar, they will all have similar junction capacitances, so no external capacitors, or resistors, are needed.

Long ago, before they knew better, some commercial high-voltage silicon rectifier-stack manufacturers used "equalizer" R/C networks. These manufacturers stopped using "equalizers" for the same reasons that were previously outlined above. I don't know of any commercial HV-rectifier manufacturer who has not abandoned the malpractice.

Silicon Rectifier Failure

When a silicon rectifier fails from excessive reverse-current, the rectifier will short-circuit. This failure mode is very rare in SB-220's because the per-leg total rectifier PIV-rating used {>4200v} is more than 1000V higher than the actual PIV {c.3100v} in the circuit. This represents a conservative design because, during a voltage surge, the eight electrolytic filter capacitors, which are rated at 3600V-max, would likely fail before a >4200PIV rectifier string.

A much more common type of rectifier failure in early production SB-220s is rectifier-opening. This is caused by a defective spotweld inside the silicon rectifier. Eventually the weld breaks and the rectifier opens. The forward voltage jumps the gap at the open weld. When this happens, a 60Hz arc can usually be heard from inside the amplifier when current is being drawn from the HV supply. It is important to switch off the amplifier immediately when this noise is heard. Here's why: In a fullwave voltage-doubler rectifier circuit, there are two, series, filter capacitors.[10] One capacitor charges on the positive half of the cycle. The other capacitor charges on the negative half of the cycle. The two capacitors discharge in series. When current is being drawn from the supply, and if one of the filter-capacitors is not being charged by its rectifiers, the capacitor that is being charged will force reverse-polarity current through the capacitor that is not being charged. If unchecked, reverse-current will cause electrolytic capacitors to discharge their corrosive electrolyte through their safety vents. In other words, reverse-polarity current will destroy polarized electrolytic capacitors in short order.

RL1, The Antenna and Bias Relay

See Diagram 1.

1. The added, reverse diode across the coil absorbs the reverse-voltage spike when current stops flowing in the coil. This prolongs the life of the coil insulation and quenches the magnetic pulse from the coil. If the magnetic pulse is unchecked, it can retrigger the transceiver's VOX circuit if a nearby, high-Z dynamic microphone is being used.

2. In its stock wiring configuration, +110V is connected to terminal #2 of the relay. During receive, the relay connects this voltage to the center-tap of the filament-transformer [terminal #8], which is the DC cathode-current path to the filaments of the 3-500Zs. The positive cathode-voltage causes the tubes to cutoff during receive. The tubes cut-off because the grids are relatively 110V more negative than the cathode's electrons. Electrons are negatively charged, and since like charges repel, no electron-current flows in the tube.

A problem arises if one of the tubes happens to develop a filament-to-grid short: Since each grid is DC-grounded, a shorted tube also short-circuits the +110V power supply. This power supply is part of the unfused filament-transformer.

Thus, if a filament-to-grid short occurs, and the amplifier is not switched off promptly, the filament-transformer will literally melt-down and short-out. The black tar that comes out of the overheating transformer makes an unpleasant mess inside the amplifier.

This potential source of grief can be eliminated if the relay is rewired as shown in Diagram 1. The new circuit uses resistor-cutoff bias, utilizing the existing 100k-ohms resistor [R27], which is rewired to relay terminal #5. The current through this resistor, during receive, is normally less than 0.25mA [P<0.007W], so the 1/2W rating is more than adequate.

3. The antenna-relay is mounted on a rubber grommet. This was intended to reduce the vibration that the relay transmits to the chassis, which acts as a sounding-board. In time, the rubber in the grommet hardens. This increases the acoustic noise that the operator hears when the relay makes or breaks.

This problem can be corrected by: removing the mounting screw and the grommet from the top of the chassis and applying a small dab of silicone-rubber adhesive through the hole.

After the silicone-rubber cures, a further noise reduction can be gained by installing U-shaped strips of thin flexible copper ribbon between the stiff wires that solder to terminal #s 4, 6, 7, and 9, and the relay terminals. The stiff wires should be shortened about 1cm before the U-links are soldered in. The flexible U-links act as shock-absorbers, and prevent the stiff copper wires from transmitting vibration from the relay to the chassis. This simple modification results in a substantial noise reduction.

4. During "barefoot" operation on 10m, when the amplifier is switched off, the bypass, or "straight-through"-SWR of the amplifier is much less than wonderful. This is due to the inductive reactance in the relay. The relay's inductive reactance [XL] can be cancelled by adding capacitive reactance [Xc] between terminal #9 of the relay and chassis-ground. The capacitance required is usually between 30pF and 43pF with a 1kV rating. Some, apparently conservatively rated, 500V micas seem to be able to handle the job. If a ceramic capacitor is to be used, the NPØ type is best, rated at 1 or 2 kV.

If you are a perfectionist, you may want to install a compression mica trimmer-capacitor, and adjust the trimmer for the lowest 10m bypass-SWR when the amplifier is connected to an accurate 50-ohms termination. The trimmer is removed, the value of capacitance measured on a C-meter, and a fixed capacitor of similar value installed permanently in the amplifier.

Parasites

The most serious design weakness in the SB-220 is its tendency for intermittent, VHF parasitic-oscillation at roughly 110MHz. This problem is far from being unique to the SB-220. I know of no model of commercial MF/HF ham band, grounded-grid amplifier that has not occasionally had a VHF parasitic problem.

Some owners will take issue with this statement because many amplifiers will play for years and never exhibit any sign of a VHF parasitic-oscillation. Other amplifiers will pop, arc, or unexpectedly go bang, often eating amplifier parts. The difference between amplifiers that "take-off" and those that don't lies in the particular tubes that are installed in the amplifier. Tubes that have above-average VHF-amplification are those that are most likely to cause grief. Below-average gain tubes, or broadcast station "pulls", will probably never be able to sustain a VHF parasitic-oscillation.

A common warning sign that any amplifier is on the verge of a full-blown parasitic-oscillation is minor arcing or spitting at the tune-capacitor or the bandswitch. The full-blown parasitic-oscillation is hard to miss. It usually makes a loud bang or pop. If the amplifier is not switched off promptly after the big bang, the filament-transformer may melt down.[1]1

The reasons for this parasitic-oscillation problem are the same as the reasons for the parasitic-oscillation problem in the Trio-Kenwood TL-922 amplifier, which was discussed at length in an article that begins on page 36 in the October 1988 issue of QST Magazine.

Briefly, the heart of the parasite problem in the SB-220 lies in the use of high VHF-Q copper-conductors between the TUNE-capacitor and the anode-connections, or "plate-caps", on the 3-500Zs. The high VHF-Q parts includes the factory-stock "parasitic-suppressors" which are made with copper wire.

This problem can be easily corrected by constructing low-VHF-Q parasitic-suppressors, using VHF-lossy, nickel-chromium-iron alloy ["nichrome-60"] wire and replacing the copper-braid between the DC blocking capacitor and the top of the HV RF-choke with a pair of unequal length nichrome-60 wires.[12] Note: The importance of the double nichrome wire replacement for the copper braid should not be discounted. This double wire suppressor/conductor may look unconventional and silly, but it has a damping effect on the VHF Q of the anode-circuit. In many amplifiers, VHF stability can not be achieved with the stock, copper braid conductor between the blocking capacitor and the top of the HV RF-choke.

If you have access to a dipmeter, and you would like to observe the before and after effect of installing improved parasitic-suppressors, before any modifications are made, unplug the amplifier from the electric-mains and check the anode-circuit's VHF self-resonance. The optimum check point is at the left end of the copper-braid that connects the HV RF-choke to the DC blocking capacitor at the rear of the TUNE-capacitor. The resonant frequency is typically 110MHz ±10MHz.

The dip in factory-stock amplifiers is so sharp that it can "suck-out" the oscillator in the dipmeter. So the dipmeter must be backed away from the conductor. This indicates that the anode-circuit has a very high VHF-Q. This is not good news unless you happen to need a self-excited VHF high power-oscillator.

After installing the low VHF-Q suppressors, check the dip again. The self-resonant frequency will usually not change appreciably, but the dip should now be smooth and broad. This indicates that you were successful at substantially reducing the anode-circuit's VHF-Q, and reducing the VHF-oscillating ability of your amplifier.

During the production life of the SB-220 and its successors, Heath made two changes in the design of the amplifier that were related to its parasitic problem. One change was to increase the spacing of the TUNE-capacitor. The other change was to decrease the C of the grid-to-ground capacitors from 200pF to 115pF.

The more reactive 115pF capacitors did cancel some of the internal grid-inductance in the 3-500Zs. This increased the VHF self-resonant frequency of the grid, making the amplifier slightly more stable. Unfortunately, this fix was not a sure cure.

Judging from numerous on-the-air and telephone conversations, Heath received many complaints from SB-220 owners who reported arcing at the TUNE-capacitor. In response to these complaints, Heath increased the voltage breakdown rating of this capacitor. Increasing the spacing of the TUNE-capacitor turned out to be a serious mistake.

Here's why: The original TUNE-capacitor already had a substantial breakdown-voltage safety factor, considering that the maximum, peak (HF) RF anode-voltage in the SB-220 is c.2600v. The arcing was not being caused by normal HF RF peak-voltage. Instead, the arcing at the capacitor was being caused by intermittent c.110MHz parasitic-oscillation voltage. Increasing the voltage rating of this capacitor did stop the arcing at the capacitor, but it shifted the arcing to the output bandswitch. In other words, the parasitic-voltage was jumping across the path of least resistance. For example, if the bandswitch's contact spacing was increased to stop the bandswitch arcing, the new wide-spaced TUNE-capacitor would probably begin arcing.

Pitting on the plates of an air-variable capacitor can be cleaned up with a file, and the capacitor will be as good as new. Arcing on the fragile contacts of a bandswitch is frequently fatal to the bandswitch.

Zener Diode Replacement

One of the more common casualties during a VHF parasitic-oscillation is the Zener cathode bias diode. Because cathode-current is the combination of the anode[plate]-current and the grid-current, the Zener gets zapped by the large grid-current-pulse that nearly always accompanies a VHF parasitic-oscillation. This is the same current-pulse that is responsible for causing the vast majority of filament-to-grid shorts in 3-500Zs. It is also responsible for blowing away the following components: R3, {0.82-ohms} the grid-current meter shunt resistor; the multimeter meter {if the multimeter switch is on the grid-current position}; the grid-to-ground 1mH RF-chokes, and the 200pF [13], mica, grid-to-chassis-ground capacitors.

Three disadvantages of a Zener diode are: They are not adjustable, they don't like high-current pulses, and they are not inexpensive. A much cheaper and more rugged, step-adjustable replacement for a 5.1V, 10W Zener [ZD1] can be made with seven rectifiers.

The replacement circuit is shown in Diagram 3. Bias voltage can be controlled, in c.0.8V steps, by adding or subtracting diodes. This allows the operator to set the zero-signal anode-current for the two 3-500Zs. The SSB-mode current should be 160mA to 200mA for best linearity.

The diodes can be mounted on a piece of perfboard and fastened in the power supply section. Note: The polarity arrows on the forward-conducting rectifiers point opposite to the arrow on the reverse-conducting Zener diode.

Full Break-in [QSK] Modification

Under 3milliSecond,QSK can be added to the SB-220 for well under $60 if you know the best places to buy the parts. The circuit is simple to construct and no exotic parts are used. The most expensive part is the high-speed, RF-output, vacuum-relay, which can be purchased new-surplus {unused} from Fair Radio Sales for c. $35. I sell the Matsushita RF input reed-relay. See Price-Info list.

Even if you do not operate much CW-QSK, this modification is still worthwhile because it makes working SSB-VOX enjoyable - and it eliminates hotswitching. . The relays are so fast and inconspicuously quiet, it's almost like talking on the telephone, or like talking in person. This is real communication.

The QSK circuit, for the TL-922 and SB-220, was published in the March 1989 issue of Ham Radio Magazine. The article begins on page 35. The magazine article is not free from errors, but it will give a basic idea of how the QSK circuit works. (see Fig. 7 on my Web site)

One of the features of this QSK circuit is that the electronic cathode bias switch [ECBS] is always in perfect sync with the RF-relays. In other QSK circuits, this is not the case.

Here's why: In some other QSK circuits, the ECBS is RF-actuated. This may sound wonderful but it is not so because RF-actuation allows the 3-500Zs to be switched in and out of their linear bias region during and between softly spoken syllables. The net result is increased IMD, or splatter, as the bias wanders between linear and non linear operation during speech.

Making the RF-relays and the ECBS RF-actuated is not a solution because this would cause the RF-relays to hot switch every time they tried to make.

A properly designed QSK circuit puts the QSK-transceiver's amplifier-control line completely in charge of the amplifier. In this way, the 3-500Z bias will always be correct for linear operation any time that the RF-relays are actuated.

To ALC, Or Not To ALC?

If you are using a Swan 500 to drive your SB-220, you definitely need to use the ALC circuit in the amplifier. I also recommend that you NOT display your call sign prominently at ham conventions or swap-meets. This may help to avoid any unpleasant situations that may result in a spontaneous tar and feathering.

If your transceiver puts out less than 140W-PEP, the use of ALC with your SB-220 is of no value since it is impossible to overdrive the 3-500Zs. It is for this reason that I removed the ALC circuit from both of my Heathkit amplifiers. [14]

Operation On The 12m And 17m Bands

One of the main problems with using older design, ham band only, amplifiers on the new 12m and 17m bands is choke-fires.

Here's what happens: When a HV RF-choke is operated on or near one of its self-resonant frequencies, an extremely high RF-voltage appears on the choke. The voltage can easily exceed four-times the supply voltage. This can cause the insulation on the choke's wire to break down and ignite.

To prevent choke-fires, all operating frequencies should be more than 5% away from any of the choke's self-resonant frequencies.

The SB-220 will operate well on the 12m/24.9MHz and 17m/18.1MHz bands because its HV RF-choke, RFC-1, does not have any self-resonance below 40MHz.

If a transistor-output radio will be used to drive the amplifier, the amplifier's tuned input circuits for the 10m and 15m bands should be optimized for this dual-purpose. More on that later.

The only possible problem associated with 12m and 17m band operation is the increased current burden on the output-bandswitch.

Here's why: In order for the amplifier to tune down to the new frequencies without increasing L, the TUNE-capacitor and the LOAD-capacitor must be adjusted for about 35% more capacitance. This increases the operating Q of the output pi-network by about 18%, which increases the RF-circulating-current through the bandswitch contacts by about 18%. Since Pc. I2R, the increase in contact dissipation will c. 1.18 x 1.18 x Rc. 1.39 x R. If R remains constant, this equals a 39% increase in the heat dissipated by the contacts. [15]

This is unlikely to be a problem for normal SSB operation without speech-processing. For higher duty-cycle operation, the amplifier should be switched to the lower-voltage CW/TUNE position in order to reduce the average heat dissipation on the output bandswitch contacts.

Improving Input SWR

The tuned-input circuits in the SB-220 have a typical SWR of about 1.5:1. This is completely satisfactory as long as tube-output radios are used to drive the amplifier.

Now days, the only type of radios that are being manufactured use transistor-output. The transistors are used in an untuned, broadband-transformer circuit. This type of amplifier will develop full power only when it is working into a nearly non-reactive load whose resistance is close to the design load of 50-ohmss. Many transistor-output radios are so fussy that they will begin to power themselves down with a reactive load whose SWR is 1.15 to 1. This means that the amplifier will not receive full drive power unless it has a very low, input SWR.. On many bands, this is the case with a stock SB-220.

Fortunately, the input-SWR of this amplifier can be easily improved if we understand what we are dealing with.

The job of the tuned-input circuit is more complicated than just matching 50-ohms to the input-resistance of the amplifier-tubes.

Here's why: The instantaneous input-resistance of a grounded-grid amplifier fluctuates wildly during the positive and negative voltage swings of the sinewave input signal.

When the input cathode-voltage swings positive, the grounded-grid looks negative with respect to the cathode, and the current is completely cut-off, making the input-resistance nearly infinite.

During the negative swing in input voltage, the grid looks more positive, and a large current flows in the tube, making the input-resistance very low.

For example: a pair of 3-500Zs. When the driving voltage is peaking at negative 117v, the anode-current is at its maximum peak, and the instantaneous anode-voltage is swinging to its lowest point of c.+250v, the total, peak cathode-current is 3.4a.16 Thus, the driving resistance at this point, Rin c. 117v/3.4a c. 34.5-ohms, and, incredibly, Ppeak c. 117v x 3.4a c. 397w.

Thus, the resistance swing is from near-infinite with positive driving voltage, all the way down to 34.5-ohms. [17] The instantaneous drive power requirement varies from 0w to 397w at the positive and negative peaks of the sinewave input voltage. This is not the type of load that makes for contented transistor-output transceivers.

During the positive swing in input voltage, there is virtually no load on the driver, so the tuned-input circuit must store the energy until it is needed the most, during the negative crest in the input voltage.

Thus, the tuned-input circuit's job is to act as a flywheel/energy storage system, and a matching transformer.

Q is like the mass of a flywheel. More Q makes for a better flywheel, which does a better job of averaging the wild swings in input-resistance, giving a lower input-SWR. The tradeoff is that more Q means less bandwidth. This means that, with a high Q, the input SWR may be near-perfect at the center of the band, but too high at the band edges. Thus, a compromise must be made.

Eimac® recommends using a Q of two for the tuned-input circuits in a grounded-grid amplifier. As I will show below, the SB-220 uses a Q of only about one. This is why the stock, input-SWR of the SB-220 is less than wonderful. Other commercial amplifiers that were also designed in the era before transistor-output transceivers used a Q of one or even less than one.

Q is defined as the ratio of the input-resistance of the tuned-input circuit, which is 50-ohms, divided by the reactance, in ohms, of the input-capacitor [XCin]. For example, in the SB-220, the 40m input-capacitor [C42] is 470pF. The reactance of C42 at 7.15MHz c. XCin c. 1÷[2¼ f C] c. 1÷[2¼ x 7.15x106Hz x 470x10-12F] c. 47.36-ohms, which is also written as minus j47.36-ohms. Thus, the Q c. 50-ohms/47.36-ohms c. 1.05.

When the Q of a tuned-input circuit is too low to start with, no amount of adjusting the inductor, L, and Cout can ever bring the SWR down to a low value.

Improving the input-SWR of an SB-220 is simple: increase the Q by decreasing the XCin in the tuned-input circuits. This means more Cin.

Since the resistance matching ratio of a tuned-input is semi-proportional to the Xc-ratio of the two capacitors, if Cin is increased, to increase Q, Cout must be increased in order to maintain the same resistance matching ratio. In this case it is 50-ohms to c.69-ohms.

Increasing both capacitances will lower the operating frequency of the tuned-input, so L must be decreased to bring the frequency back up to where it started. This can be accomplished by removing turns of wire from L, and by adjusting the inductor's tuning slug.

It is important to keep in mind that the matching ratio of a tuned input circuit can not be changed by adjusting the slug-tuned inductor alone. In order to arrive at the optimum SWR, Two components must be adjusted in order to change the matching ratio of a tuned input circuit.

Another factor that affects the input-SWR is the Q [c. XL/R(RF)] of the inductor. Any RF-resistance [R(RF)] that appears in the inductor will adversely affect the SWR. Smaller wire has more resistance than larger wire. Thus, it is important to use an adequate wire diameter.

As frequency increases, skin-effect increases, which increases the RF-resistance of the wire. To compensate for this, the wire diameter must be increased in proportion to frequency.

For example, at 1.8MHz, in a 100w tuned-input circuit, the wire diameter should be at least 0.5mm [c.#24AWG]. At 29MHz, „1.3mm [c.#16AWG] diameter wire is appropriate. You can't go wrong by choosing a larger diameter wire, unless it won't fit on the coil-form.

A Q of two is usually slightly more than optimum if you need to cover a large frequency spread with a single tuned-input circuit. Prime examples would be to cover 3.5MHz to 4MHz, and 18MHz to 21.5MHz, so that the 15m tuned-input will also cover the new 17m band. In these cases, a Q of c.1.5 should be used. This also applies to a 10m tuned-input that will be used on the 12m band.

A Q of 1.5 corresponds to a reactance of about 33.3-ohms {Xc-inc. 50-ohms/1.5 = 33.3-ohms} for Cin. At 3.75MHz this would require a 1275pF capacitor. 18 / The nearest standard value is 1300pF. Of course, capacitors can be paralleled to arrive at the desired C.

Measuring the input-SWR of an amplifier is an inexact science. For example, different models of SWR-meters will give different readings in the same circuit. Changing the distance between the SWR-meter and the amplifier can also change the indicated SWR.

Another complication is that modern transistor-output transceivers always use a set of switched, c.1.5-octave-per-filter, broadband output-filters. This is done so that their output signals will meet the FCC requirements for spectral purity.

At the extremes of an individual filter's bandpass, such as at 29MHz, the filter can introduce a reactance into the transmission line. This reactance can cause some peculiar results when you are trying to optimize the SWR of the tuned input circuits in an amplifier.

The best way to avoid this problem is to use a tube-type driver, such as a Trio-Kenwood TS-830S, when optimizing the tuned-input circuits. The driver must be tuned for maximum power into a known-to-be-accurate 50-ohms termination, and then not readjusted during the adjustment of the tuned-input's L and Cout. If the transceiver is re-tuned, it may introduce a reactance onto the transmission line that will affect the indicated SWR.

If the Q of the tuned input has been increased, by increasing Cin and decreasing L, Cout will need to be increased. The easiest way to find the new, higher, optimum value for Cout is by inserting a compression mica trimmer-capacitor in in parallel with the stock Cout.

With the maximum peak drive power applied, L and Cout are alternately adjusted for the best match at the center of the band. Cout can then be removed, its capacitance measured on a C-meter, and a fixed capacitor, of the same C, permanently installed in the amplifier.

If the amplifier is driven with a steady carrier [AØ, now called NØN], considerable stress is placed on the HV power supply and the RF-compartment will become very hot. The stress and strain can be reduced if the driver is set to the CW-mode. The driver is keyed by an electronic-keyer, set to 50 to 60 words per minute. The keyer sends a steady stream of dits, which have a 50% duty-cycle. The amplifier's current-meter readings should be approximately doubled to find the actual current.

Adjusting the tuned-inputs is much easier if the front panel is removed. The meter leads need to be lengthened to facilitate doing this. A chassis-ground wire needs to be added between the panel and the amplifier chassis if you want the multimeter to function when the panel is separated from the amplifier.

It is important, not to inadvertently contact the nearby HV feedthrough insulator while you are adjusting the tuned inputs. Doing so could result in your premature appearance in................"Silent Keys".

A reasonable procedure is to use insulated tuning tools and to stand on a plastic mat, with one hand behind your back. It is also advisable to wrap some 1" [25mm] plastic electrical tape around the nuts on the HV-feedthrough insulator, before the amplifier is plugged in.

If you would prefer not to work around lethal voltage, it is possible to adjust the tuned-inputs of the amplifier without applying high voltage to the anodes of the 3-500Zs. Here's how:

1. Make the appropriate changes in the tuned input circuits.

2. Disconnect the red, secondary wire of the HV-transformer from the rectifiers. Insulate the loose wire.

3. Re-connect the amplifier to the electric-mains, key and drive the amplifier with about 5W initially.

4. Observe the grid-current meter and apply only enough drive to obtain no more than 250mA of grid-current.

5. Adjust L and Cout for the best SWR.

This method is not as accurate as the full-power adjustment method, but it is safer.

Here are the optimum values I found for the tuned-inputs, using the full-power adjustment method:

Band - Cin pF / L turns removed / Cout pF

80m 2 x 680 / -4t / 1300

40m 820/ -4t / 680

20m 360 / -1t / 270

15m 270 / -2t / 180

10m 180 / 2t / 130 +/-

Notes:

1. The 10m and 15m values are affected by lead length and placement, so some fine tuning will usually be needed.

2. The amplifier was equipped with two, 10-ohms [5-ohms net] cathode resistors [Rc] These resistors increase the input-resistance of the 3-500Zs by about 8%.

3. The ALC circuit had been removed from this amplifier. This slightly reduces the load capacitance on the output of the tuned-input circuit.

4. The capacitors are mica, rated at „500v, or NPØ ceramic rated at 1kV.

Other experimenters have reported finding slightly different optimum values, so the best values for your amplifier may be different than those listed.

Removing The Tuned Input Circuit Inductors

The inductors are fastened to the chassis by two spring tabs in the base of each inductor. When the inductor base is pushed through its mounting hole, the spring tabs are compressed as they pass through the mounting hole. After the spring tabs have passed through the mounting hole, they spring-out and lock in the inductor base.

In order to remove an inductor, both spring tabs must be compressed. The upper spring tab can be easily compressed with a screwdriver blade. The lower spring tab is difficult to reach unless a special tool is used.

I made this tool out of 1/8" diameter piano wire, which can be purchased in 36" lengths in model airplane hobby shops. Here's one method of making the tool: Using a bench grinder, cut off about 30cm [12"] of the wire. With a pencil-point flame from a propane torch, heat a spot on the wire, about 2cm [0.8"] from the end of the wire. When the metal is glowing red, grasp the end of the wire with pliers and bend a c.85º angle in the wire.

The long end of the wire-tool is the handle. The short end is hooked under the inductor base. When the handle is pulled straight-up, the lower spring tab will depress. It is much easier to remove turns from the inductors when the inductors have been removed from the amplifier.

Cabinet Removal

The SB-220's cabinet can be removed as follows:

1. Use a book or a block of wood that is about 5-inches by 7 inches and c. 1.5-inches thick.

2. Hold the book or wood block spacer flat against the rear of the amplifier, between the air inlet screen and the right edge of the cabinet, with the c.5-inch dimension vertical.

3. Using your other hand, rock the amplifier over backwards on top of the spacer, until the amplifier is standing up. If the spacer is positioned properly, it will be supporting the full weight of the amplifier and no part of the cabinet will touch the table.

4. Remove the four screws with their rubber feet.

5. Slide the cabinet up and off.

Adding 160m Coverage

Unfortunately, a number of technically unsound, 160m conversions for the SB-220 have been published. Most of these conversions unnecessarily discard the original filament and/or HV RF-chokes and ignore RF-design rules. A better 160m conversion can be found in the January, 1989 issue of QST Magazine, beginning on page 23.

Conclusion

The Heathkit SB-220, and its younger cousins, the SB-221 and HL-2200, can provide many years of trouble free service. The only things they need from their owners are a few circuit-improvements, and a yearly cleaning and fan oiling.

>>>>> If you have any questions or comments about this article, please feel free to telephone me at 805 386 3734.

Rich, AG6K

End Notes

1 The filament-pins receive a considerable amount of heat through conduction from the c.75w filament. The amount of heat present requires that continuous forced-air cooling be directed at the filament-pins, even on standby.
2 The designers of the Trio-Kenwood TL-922 amplifier, whose circuit looks amazingly similar to the SB-220 circuit, exhibited some original thinking when they decided to incorporate black, low heat-reflective surfaces, adjacent to the 3-500Z anodes.
3 The fan does not need to be removed to be drilled.
4 "WD-40", "LPS", and similar products, are NOT non-gumming.
5 This oil can be purchased in stores that sell fishing reels and/or firearms. Ordinary, 10w or 20w SG-grade motor oil can also be used.
6 Meter damage can be avoided by connecting two, ordinary „1A, any PIV, silicon rectifiers across the terminals on each meter. The two diode arrows should point in opposite directions.
7 When the SB-220 is powered from 120.0V or 240.0V, the no-load HV is close to 3080V.
8 For an extensive treatment of this subject see: "The Emperor's New Clothes" by Hans Christian Andersen.
9 The areas to be bonded should be degreased. After the stepstart parts are in place, the amplifier should not be disturbed for at least 24-hours to allow the silicone-rubber adhesive to cure.
10 In the SB-220, each of these two capacitors is made from four, 200µF, 450V capacitors in series. Thus, the four capacitors act as a single 50µF, 1800V capacitor.
11 To be more precise, after the big-bang, if the anode [plate]-current meter indicates a current flow, when the amplifier is not in the transmit mode, a 3-500Z filament has been pushed against its {grounded} grid. This short-circuit grounds everything that is connected to the filament circuit, including the unfused, +110V relay supply. For more info see "RL1, The Antenna and Bias Relay", #2.
12 If you would like to purchase a Suppressor Retrofit-Kit for the SB-220, they are available from me
13 In later models, the grid-to-chassis capacitors were changed to 115pF.
14 For more information, see "Amplifier Driver Compatibility" which begins on page 17 in the April 1989 issue of QST.
15 In actual use, the metal in the contacts will run hotter because of the increased current burden. This will increase R. Thus, the contact dissipation will probably increase more than 39%.
16 At the instant of maximum peak-current(s), the peak grid-current, per tube, is c.0.5a and the peak anode-current, per-tube is c.1.2a. Thus, the peak cathode-current is 1.7a per-tube. This represents a meter- indicated anode-current of about 800mA for two 3-500Zs.
17 The average input-resistance for a pair of 3-500Zs is about double this value, or c.65-ohms.
18 The capacitors used should be 500V silver-mica type or „1000V ceramic, NPØ type.
19 These suppressors are supplied with the suppressor retrofit-kits.
20 For more information, see "Amplifier Driver Compatibility" which begins on page 17 in the April 1989 issue of QST.

Diagram

End

AG6K Part 4

2026-08-20T04:30:46.888-07:00

ALC

Today, the typical transceiver output is 100W to 200W. There are amplifier tubes that can be destroyed by 100W of drive. A good example is the 3CX800A7. Driving a 3CX800A7 with 100W PEP will eventually strip flakes off of the cathode. The flakes lodge between the cathode and the grid cage--creating a fatal short. Even a pair of 3CX800A7s are clearly over-driven by 100W. The fix is: connect a approx. 40 ohm cathode negative-feedback resistor in series with each 3CX800A7 cathode. As a result, the 3CX800A7s won't be driven above their maximum ratings--and into non-linearity--by a 100W transceiver. Naturally, when cathode negative-feedback resistors are added, the cathode driving impedance increases. The driving impedance for a pair of 3CX800A7s is about 25 ohms. With 40 ohm cathode resistors, the driving impedance is roughly 50 ohms.

Cathode negative-feedback resistors are better than having a matched pair of 3CX800A7s. The cathode currents automatically equalize themselves---and unlike ALC circuits, cathode feedback resistors work instantaneously--eliminating ALC's generic flaw--leading edge splatter on SSB. Amplifier-to-transceiver ALC works properly only on constant signal level modes such as RTTY and FM.

The 3-500Z is rated at approx. 60 watts drive. When a single 3-500Z is driven by 100W, it "flat-tops" and produces distortion. A 25 ohm low-L cathode feedback resistor will make a 3-500Z linear with 100W of drive. The resistor is placed in series with the cathode RF coupling capacitor.

Capacitors and RF Current

Capacitors that carry RF current are subject to two types of internal heating. Like ripple filter capacitors, the ESR in the capacitor's conductors generates heat that is proportional to I^2 x R. Due to skin-effect, R goes up with frequency. Another source of heat is RF dielectric loss. Since dielectric loss usually varies with frequency, the current carrying ability of capacitors changes with frequency. Typically, transmitting capacitors are current rated at three widely-spaced frequencies. It's a good idea to check the manufacturer's current ratings before using a transmitting capacitor in a specific application. Just because a capacitor is a transmitting-type does not mean that it will work reliably in all RF applications.

Tuned Circuit Q

Most of the tuned input and tuned output circuits in HF amplifiers are pi-networks. There are a number of ways to define the Q of a pi-network. In what follows, Q is defined as the input impedance of the pi-network divided by the reactance of the input, shunt element--typically a capacitor. This definition of Q is the one used by Eimac® in Care and Feeding of Power Grid Tubes.

Tuned Input Circuits for Class AB2 Cathode-Driven [grounded-grid]

Even though grounded-grid amplifier circuits look simple, they are not. The grounded-grid amplifier's tuned input circuit is in series with and out of phase with the anode current pulses. The RF cathode current's approx. half sine wave pulses are the sum of the anode and grid currents. Since the driver is connected to the other end of the tuned input, some of the RF cathode current finds its way back to the driver. Consequently the driver interacts with the amplifier. The Q of the amplifier's tuned input affects this interaction.

Modern solid-state output MF/HF transceivers use a broadband push-pull RF output stage. In order to meet FCC requirements, Butterworth and/or Chebyshev pass band filters are used to suppress spurious emissions. Such filters introduce inductive reactance or capacitive reactance within their pass bands. In other words, the output impedance of a modern transceiver is seldom 50 ±j0 ohms. When driving a tuned input in a grounded-grid amplifier, filter reactance interacts with the input reactance in the tuned input. The length of the coax between the driver and the tuned input affects the interaction.

When tube manufacturers state the cathode driving impedance in grounded-grid operation, they are talking about an average value. The instantaneous driving impedance fluctuates wildly during the sine wave input signal. During most of the positive half of the input cycle, the grounded-grid looks negative with respect to the cathode--so the flow of current is cut-off. Since virtually no current flows, the driving impedance is extremely high.

During the negative swing in the input cycle, the grounded-grid is relatively positive. A positive grid accelerates electrons away from the cathode, producing high anode-current and grid-current. Due to the large flow of current, the input-impedance is low during the negative half of the input cycle.

Consider a pair of 3-500Zs. When the driving voltage is peaking at negative 117v, the anode-current is at its peak, and the instantaneous anode-voltage is at its lowest point--about +250v. At this instant, the total, peak cathode-current is 3.4a. Thus, the instantaneous cathode driving impedance is 117v/3.4a = 34.5 ohm--and the peak driving power = 117v x 3.4a = 397W.

In other words, the instantaneous driving impedance swing is from near-infinite all the way down to 34.5 ohms. The instantaneous drive power requirement varies from 0w at the positive peak to 397w at the negative peak of the input sine wave. Thus, the input pi-network's job is to act as a flywheel/energy storage system and a matching transformer. That's why a simple broadband transformer can not adequately do the job of matching the driver impedance to the cathode impedance in a grounded-grid amplifier.

The Q of a tuned circuit is like the mass of a flywheel. More Q makes for a better flywheel--which does a better job of averaging the wild swings in input-Z--thereby producing a lower input-SWR. The trade-off is that more Q means less bandwidth. With a high Q, the input SWR may be near-perfect at the center of the band, but unacceptable at the band edges. Thus, a compromise is in order. Eimac® typically recommends using a pi input network Q of 2 for Class AB2 grounded-grid operation. To arrive at a Q of 2, the reactance [X] of the input capacitor, C1, is minus j50 ohm÷2=minus j25 ohm. Using C=1÷[25(2Pi x f)], approximately 220pF of input capacitance is needed for a Q of 2 on the 10m band. In actual practice, however, 220pF may be far from the value that produces a satisfactory SWR with a particular model transceiver and a particular length of coax. It may be possible to find a length of coax that would ameliorate this problem on 10m--but there are eight other bands to contend with below 30MHz. Since band switching different lengths of coax is hardly practicable, it would be useful if the input capacitors were adjustable in a grounded-grid amplifier's tuned input circuits. Adjustable coils are also useful.

Tank Circuits

When the Q of the output pi-network tank circuit is low, two problems can occur. The harmonic attenuation may be inadequate to meet FCC requirements--and the load impedance matching range decreases. In other words, when Q is low, the tank circuit may be incapable of matching even a 50 ohm load. When the Q of the tank is too high, efficiency decreases due to the increase in I^2 R circulating current losses. A compromise is in order. A Q of 10 is about minimum. A Q of 20 may cause excessive tank component heating due to high circulating current. A Q of 12 to 15 is a fair compromise.

Better tank performance can be achieved by using a pi-L tank circuit. When compared to a simple pi, the pi-L has roughly 15db better harmonic attenuation and it typically has a wider matching range. The trade-offs are that the pi-L requires an extra switch section and a tapped inductor.

Skin Effect and Current Capability

As frequency increases, progressively less current flows inside a wire--so current progressively concentrates on the surface. Since a steadily decreasing part of the conductor is being used, resistance increases as frequency increases. For example, a 12 gauge (copper) wire will carry 20A at 60Hz with very little heating. At 30MHz, the RF current carrying ability of 12 gauge wire is about 5A. Band switch contact current ratings need to be similarly de-rated as frequency increases. Paralleling contacts is a good way of increasing the current handling ability of a band switch. Directing a portion of an amplifier's cooling air flow at the band switch improves the RF current handling ability of band switch contacts.

HF tank inductors can become quite lossy unless the conductor surface area varies in proportion to frequency. Inadequate tank conductor size is the main reason for decreasing amplifier efficiency at the higher frequencies. A tank inductor made from 14 gauge wire is usually more than adequate for efficient 1.8MHz operation at 1500W PEP. For efficient operation at 29MHz, approx.10 mm o.d. copper tubing (or copper strap with an equivalent surface area) is appropriate. However, due to normal QSB--at the receiving end, even a one-third decrease in transmit power is virtually undetectable. Thus, squeezing out the last percentage of efficiency on 10m is not very important.

Calculating the RF circulating current in a tank inductor is fairly complex. A quick approximation is to multiply the maximum anode-current by Q. For example, if the anode-current is 1.2A and the tank Q is 15, the RF circulating current in the tank will be 1.2*15 =18A. At 29MHz, 18A is a formidable amount of current.

Silver

Compared to copper, silver [Ag] is cosmetically more attractive and more immune to oxidation. However, silver does not make an amplifier measurably more efficient at frequencies below about 100MHz. Copper oxidation can be prevented by polishing copper with extra fine steel wool and applying clear, gloss, polyurethane varnish.

Silver is useful as a component of solder. 95% tin [Sn], 5% silver, solder has a melting temperature of 221 degrees-C/430 degrees-F. Compared to tin-lead electronics solder, 95/5 Sn/Ag solder is about 3.5 times stronger and it has better wetability--especially on hard-to-solder materials. 95/5 Sn/Ag solder is ideal for soldering tank components, band switches, surface-mount solid state devices, loose vacuum tube pins, and low Q parasitic suppressors. When resoldering a tin-lead solder joint with tin-silver solder, first remove as much of the tin-lead solder as possible.

Anode HV RF Chokes

The basic requirements are: 1. The choke must have ample reactance at the lowest operating frequency to limit the RF current through the choke to a reasonable amount. 2. The choke can not be self-resonant near an operating frequency. 3. The wire gauge used must be able to carry the DC anode current plus the RF current at the lowest operating frequency without excessive heating.

If the HV-RFC has a self-resonance on or near an operating frequency, potentials of many times the anode supply voltage can appear on the choke. When this occurs, a choke arc and fire is likely. Choke fires can destroy more than just the choke because the rising plume of ionized gasses from the choke fire often creates a conduction path to the ceiling of the RF output compartment. If an arc occurs, pervasive damage is likely if no glitch protection resistor was used in the HV positive circuit.

Choke Design

Materials:

There are two types of wire insulation materials that are suitable for use in HV-RFCs--silicone varnish and Teflon. Modern, high-temperature electric motor wire is insulated with a tough, silicone varnish that can handle high DC voltage and high RF voltage. At room temperature, a twisted pair of #20 silicone varnished wires can withstand more than 5000VDC or 1500W in a 50 ohm circuit at 29MHz. This type of wire is sold by the pound in electric motor rewinding shops. If you want to buy some, bring your own empty spools and winding device--such as a variable-speed electric drill, with a homemade adapter to hold the spool. Due to its toughness, silicone varnish insulation requires a special method of stripping. An open flame from a butane lighter causes the silicone varnish to decompose and combust. The remaining ash residue can be removed from the copper with steel wool.

Teflon insulated magnet wire is not common. Although ordinary Teflon insulated hookup wire may be used, the extra insulation thickness requires that a longer coil form be used. One potential trade-off with Teflon insulated wire is phosgene. When Teflon burns, deadly phosgene [COCl2] gas is produced.

Due to contact with air, the current carrying ability of either type of wire is much higher in an HV-RFC than it would be in a transformer. #28 wire will easily carry 1A in a HV-RFC. #24 will carry several amperes with acceptable heating.

G10 or G11 epoxy-fiberglass tubing is RF-resistant, strong, and easy to work with. It is an ideal material for building HV RF chokes. It can be obtained from plastic supply houses. 1mm wall thickness is more than adequate. Diameters of 16 to 25 mm are typically used for building HV-RFCs. G10 tubing can be cemented to a G10 base plate with silicone rubber adhesive or epoxy. A source of G10 tubing: Plastifab,1425 Palomares, La Verne, CA 91750 818 967 9376.

It is probably a good idea to limit RF current in the HV-RFC to no more than 1 ampere. To calculate current in the choke, take roughtly 2/3 of the anode supply volts and divide it by the reactance in ohms at the lowest operating frequency -- a.k.a. Ohm's Law.

Determining Bypass C

Power supply components can be damaged by RF. Electrolytic filter capacitors are especially at risk. Thus, adequate RF bypassing on the power supply side of the HV-RFC is needed. Probably no more than 10V of RF should be allowed to appear on the +HV supply at the lowest operating frequency. Determining just how much bypass C is needed basically involves using ohm's Law. The amount of RF current flowing through the choke and the amount of bypass C need to be evaluated for the lowest operating frequency--usually 1.8MHz. For example, if the reactance of the choke is +j2000 ohms, and the AC anode voltage is 2000Vrms, then I=2000V/2000 ohm=1A of RF flows through the choke. In order to limit the RF voltage to 10V maximum at 1.8MHz, 10V/1A=10 ohm of capacitive reactance is needed for an adequate bypass. Using C=1/(Xc * 2pi * f), this equates to a HV bypass capacitance of 8842pF. Obviously, a typical 1000pF bypass C [minus j88 ohm] is not going to do the job because it would allow approx. 88V of RF to appear across the HV supply if 1A were flowing through the choke.

500pF 20kV TV-type doorknob capacitors are NOT designed to handle RF current--so they do not make satisfactory HV bypass capacitors. Disk ceramic capacitors may be used for HV bypassing. Disk ceramic capacitors are somewhat limited in the amount of RF current they can safely handle. Manufacturers typically don't publish RF current ratings for them. To find out how different capacitors react to RF current, you must test them yourself. Even a 7500WVDC, 2500pF disk ceramic capacitor becomes warm from 1A at 1.8MHz. Thus, it is often best to parallel a number of individual bypass capacitors so that the RF current will be shared among them.

Determining L

At the lowest operating frequency, the HV-RFC should have enough reactance to limit the RF circulating current through the choke to a reasonable amount. Allowing a RF current of 1A RMS through the choke usually does not create problems for the wire-lead disc-ceramic capacitors that are typically used to bypass RF on the power supply side of the HV-RFC. To minimize RF current through the choke, it would seem that more inductance is the answer. However, more inductance means more choke resonances and a greater likelihood of choke fires. A compromise is indicated.

Over the years, various schemes have been used to minimize choke resonances. Adding gaps at presumably esoteric positions in the winding was represented as a means of decoupling parts of the choke winding--allegedly ameliorating the self-resonance problem. However, when the resonances of gapped chokes are compared to similar chokes without gaps, no real improvement is observed on a dipmeter. This should not be surprising. Optimum decoupling between two coils occurs when they are mounted at a right angle. Adding end-to-end spacing with gaps is the least effective decoupling method possible. To minimize resonance problems, instead of using a single large choke, use two smaller chokes mounted at right angles.

The highest-L choke that can built that is free of self-resonances in the HF spectrum is roughly 60µH. At 1.8MHz, 60µH has a reactance of about +j679 ohm.

The RMS voltage that appears across an amplifier's HV-RFC is approximately two-thirds of the anode supply voltage. For example, an amplifier that is powered by a 3000V supply subjects its HV-RFC to about 2000V RMS. If a 60µH inductor was used in this amplifier, at 1.8MHz the RF current through the choke would be 2000V/679 ohm=2.95A RMS. Adequately bypassing approx. 3A of current on the power supply side of the choke is difficult. A typical HV disk ceramic bypass capacitor can handle only about 1A. Another problem is that at 1.8MHz 130pF [minus j679 ohm] of extra capacitance is required from the tune capacitor to cancel the +679 ohms of reactance in the choke. Adequately bypassing 3A at 1.8MHz requires a substantial amount of capacitance. To hold the voltage across the bypass capacitors to less than 10V at 1.8MHz, roughly 0.026µF [minus j3.3 ohm] is indicated. To handle this amount of current, four approx. 0.0075µF HV disc ceramic capacitors would probably be needed. All things considered, using more inductance is indicated. Limiting the HV-RFC's RF current to a maximum of 1A would make the task of bypassing a lot easier. However, increasing the inductance above 60µH is virtually certain to move choke resonances into the HF range. Unless these resonances are prudently parked between operating frequencies, a choke fire may result.

To realistically evaluate the self-resonance situation, HV-RFCs should be checked with a dipmeter after they are installed and wired in the amplifier. If a self-resonance is within about 5% of an operating frequency, there may be a problem. When re-parking resonances, it is usually best to remove turns from the choke. This will move the resonances up in frequency--and only slightly increase the maximum RF current through the choke.

In continuous coverage amplifiers, there are obviously no safe parking places for choke resonances. The only solution is to switch HV-RFCs with one or more HV vacuum relays.

HV-RFCs should be single-layer solenoid wound. To minimize wire vibration during operation, the wire should be under constant tension when winding and soldering the ends to the solder lugs. When silicone varnish insulated wire is used to wind a HV-RFC, the finished winding should be given a coat of gloss urethane varnish to hold the wire in place. Since varnish will not adhere to Teflon wire, a different method is needed to keep a Teflon winding taught. Small tensioning springs are soldered to the ends of the wire. The springs provide constant pull to minimize wire vibration during modulation. An S-shaped copper foil jumper should be connected across each tensioning spring.

DC Blocking Capacitors

Blocking high voltage DC is the least difficult part of the blocking capacitor's job. During operation on 10m, the DC blocking capacitor must be able to carry most of the RF circulating current in the tank. Here's why: The amplifier tube's anode capacitance normally provides most of the tune capacitance during 10m operation. Thus, a major portion of the tank circulating current passes through the anode capacitance and therefore through the DC blocking capacitor. In an amateur radio amplifier, blocking capacitor currents of 5 to 10 A RMS are not uncommon during operation on the 10m band.

Selection of a blocking capacitor should not be guesswork. It is advisable to select a capacitor or capacitors that is rated to carry the calculated maximum RF current present. Merely selecting an RF-type (transmitting) capacitor is not good enough. Some RF-type capacitors have rather unspectacular current capabilities. The capacitance of the DC blocking capacitor is not very critical. 1000pF seems to be more than adequate for operation at 1.8MHz. 88 ohms of Xc is relatively insigificant in comparison to the typical 1000 to 2000 ohm anode output Z.

Vacuum Components

Vacuum capacitors and vacuum relays are ideal for use in high power RF amplifiers because they can withstand high RF voltages. vacuum capacitors are able to handle more RF current than any other type of capacitor. There are some trade-offs. Vacuum components depend on their glass-to-metal or ceramic-to-metal seals to maintain their near-perfect vacuum. If a seal leaks, air molecules enter and the vacuum component is kaput. Vacuum component seals should not be subjected to unnecessary mechanical stress.

Vacuum Capacitors

Although vacuum capacitors can be mounted in any position, vertical mounting places the least stress on the soft copper plates. Vertical mounting also makes the most efficient use of chassis space. With vertical mounting, a right-angle drive is used to bring the 1/4" diameter tuning shaft to the front panel. Cardwell-Multronics® makes a compact right-angle drive mechanism that is ideal for this application. It is designed to replace the shaft-cap on a vacuum capacitor's tuning shaft. The vacuum capacitor should be set for minimum C before the drive shaft cap's setscrews are loosened.

A vacuum capacitor should not be used as a standoff-insulator to support heavy components. High G force can be fatal to a vacuum capacitor. The danger is not necessarily breakage or damage to the seals. The plates in a vacuum capacitor consist of a series of concentric, intermeshing, soft copper cylinders that almost touch each other. A vacuum capacitor can be shorted by an inertia force that is capable of bending the soft copper plates.

Vacuum Relays

To avoid stressing the seals, connections to the contact terminals of vacuum relays should be made with soft copper ribbon.

The molded-in coil terminals on vacuum relays are easily broken. Connections to the coil terminals should be made with approx. 24 gauge stranded hookup wire.

Vacuum relays generate sharp mechanical vibrations when they switch. If one is mounted securely to the chassis, the chassis acts like a speaker cone--coupling the vibrations more efficiently to the air. One way of overcoming this problem is to mount the vacuum relay on small beads of silicone rubber. To accomplish this, drill a approx. 3mm oversize mounting hole in the chassis. Use temporary L-shaped poster board spacers to prevent the relay from touching the chassis. After cleaning the surfaces with acetone, apply three small beads of silicone rubber between the relay mounting flange and the chassis. Allow the silicone rubber to cure for 2 days. Remove the spacers. The relay should float quietly on silicone rubber shock absorbers. The vacuum relay's body should be grounded to the chassis with thin copper ribbon. The ribbon may be soldered to the edge of the relay flange. To avoid overheating the seals, use a large soldering iron--and tarry ye not.

Controlling Make and Break Times

All relay coils have inductance. Since inductance delays a change in the flow of current, coil-inductance tends to increase the make-time of relays. Make-time is an important design consideration when using vacuum relays for RF switching. RF-rated vacuum relays use copper contacts to obtain high conductivity. However, copper is vulnerable to damage from hot-switching. For example, if an amplifier's RF output relay contacts are not closed and finished bouncing before the RF arrives, arcing and contact damage is likely.

Make-time can be decreased by supplying extra voltage to the coil during start-up with what is commonly called a speed-up circuit. Jennings® and Kilovac® recommend using them to accelerate relay closure. A speed-up circuit consists of a resistor in series with the relay's coil and a power supply that supplies two to three times the rated coil voltage. At turn-on, the extra voltage hastens the flow of current in the coil. The resistor limits the steady-state coil voltage to a safe value after the flow of current builds up in the coil.

DC relay coils are usually paralleled with a diode to absorb the reverse voltage spike that results when current stops flowing through the coil. If no reverse diode is used, the reverse voltage spike can exceed 20 times the rated coil voltage. The break-time of a DC relay can be controlled by adding a resistor in series with the diode. As R increases, the break-time decreases. R should probably not exceed three times the coil resistance.

Testing Vacuum Components

When a vacuum seal leaks air, the breakdown voltage decreases. This problem is easy to spot in a glass-body vacuum relay--because when electrons flow through air, blue-purple photons are emitted. With glass-body vacuum capacitors, this problem is not as obvious. In a leaky glass-body vacuum capacitor, internal ionization/arcing is often not visible since the problem usually occurs deep inside the meshed concentric plates.

It is a good idea to test all vacuum components, whether they be new or used,before constructing the amplifier.

When a vacuum component in an amplifier becomes gassy, arcing typically occurs near the crest of the RF sine wave--so a bad vacuum component typically reduces the peak power output. Since many amplifiers use more than one vacuum component, finding the bad one is difficult without individual evaluation using a breakdown tester.

Random replacement--a.k.a. "Easter-egging"--is not an efficient way to repair an amplifier that uses vacuum components. For instance--if an amplifier's RF output vacuum relay becomes gassy, it is virtually certain to divert high power RF into the (usually more delicate) RF input relay. If a thusly-damaged RF input relay is replaced, the new RF input relay may also be damaged by the gassy RF output relay. Thus, it is desirable to be able to individually test vacuum components with a breakdown tester.

Testing the quality of a vacuum is similar to testing the breakdown voltage of a diode. Connect a approx. 100M ohm HV resistor and a approx. 20 microampere meter in series with a breakdown tester. Increase the voltage until about 1 to 2 µA of leakage is detected. This voltage is the breakdown voltage. The peak RF working voltage of a vacuum component is roughly 60% of the DC breakdown voltage.

Measuring Relay Contact Resistance

There are basically two types of vacuum relays--those that are designed for hot-switching, and those that are not. Hot-switching-capable relays have tungsten contacts. Such relays are intended for use primarily in power supplies. Relays that are designed for RF have copper contacts. They should never be allowed to hot-switch. Copper-contact relays have approximately one-third the contact resistance that similar tungsten-contact relays have. For instance, the Jennings RJ-1A is the copper-contact version of the tungsten-contact RJ-1H. The rated contact resistance of the RJ-1H is 30 milli-ohms. The rated contact resistance of the RJ-1A is 10 milli-ohms. Tungsten contact relays are not rated for current RF current. However, they should work fine for RF service if they are operated at roughly two-thirds of the RF current rating for their copper-contact counterparts. Tungsten contacts are extremely hard. They are capable of more operations than copper contacts. For heavy , full break-in telegraphy use, tungsten contacts are preferable--even though they do not have the continuous RF current handling capability of copper contacts.

With vacuum relays, contact failure is not uncommon. Contacts suffer from contact erosion. This condition increases contact resistance. Eventually, an eroding contact will open completely. To test a vacuum relay, the resistance of normally open [NO] contacts and the resistance of normally closed [NC] contacts should be measured and compared with the manufacturer's specifications. Ordinary ohm-meters are not suitable for detecting contact problems other than an open circuit. The voltage drop across relay contacts should be measured with a substantial current flowing. 1A is a reasonable current to use. Measure the mV drop directly across the contact terminals using a DMM with test prod leads. Most of the vacuum relays that are designed to handle RF current have a rated contact resistance of less than15 milli ohm--so no more than 15 milli V should appear across the terminals with 1A flowing through the contacts.

Testing Vacuum Capacitors

Vacuum capacitors store energy efficiently because they have virtually zero ESR [equivalent series resistance] and internal L--thus, the peak discharge current can be astronomical. When the breakdown test voltage is high enough to create more than a few microamperes of leakage, a vacuum capacitor will normally self-discharge--producing a clearly audible tick due to the large peak discharge current and commensurately large electromagnetic force. After a vacuum capacitor self-discharges, it begins charging and the process repeats. A vacuum capacitor should not be allowed to self-discharge more than a few times unless the capacitor has been in storage for many years. During long-term storage, for some as yet unexplained reason, copper atoms tend to line up, forming whiskers on the surface of the plates. Copper whiskers initially reduce the breakdown voltage. Copper whiskers can be dislodged by self-discharge. If the breakdown voltage increases after a self-discharge, another self-discharge may be beneficial. Repeated self-discharge will cause a decrease in breakdown voltage.

Grounded-Grid Amplifier Tune-up

Linear amplifiers are like induction motors--they are designed to run fully-loaded. If your grounded-grid amplifier's instruction book says to reduce drive power during tune-up--and most of them do--it is not giving you correct information. In order to be linear, amplifiers must be tuned-up with the same peak drive-power level that they will be driven with during actual operation. Reducing drive power changes the output Z of the amplifying device to something other than the tank circuit was designed to match. Thus, the tune and load settings with low drive will be wrong when normal drive is applied.

Tune-up method #1: Set the amplifier's HV supply to the CW-Tune/low-V-tap. If you are not sure where to preset the Load control, set it to >70% of maximum loading [30% of C] to be safe. Apply the drive level that you intend to drive the amplifier with during actual use. Alternately adjust the amplifier's Tune and Load controls for maximum relative power output. The whole process should take less than 6 seconds. It may sound brutal, but this tune-up method results in good amplifier linearity and it won't damage the tubes if the maximum anode-current rating is not exceeded. If the anode-current is excessive, the resistance of the cathode, RF negative-feedback resistor needs to be increased slightly--or the PEP adjust control in the transceiver needs to be turned down.

Tune-up method #2: [not for FM, AØ, and RTTY operation] To reduce the stress on an amplifier during tune-up, use a reduced duty-cycle driving signal. This can be accomplished by keying the transceiver, on CW mode, with a CW keyer, set to send approx. 50wpm dits. CW dits have a 1/2-on, 1/2-off, or 50% duty-cycle. Using this method, the amplifier may be tuned-up, again for maximum power output, in its higher-voltage, SSB-mode. Keyers that have a weighting adjustment can be set to produce a light dit that has a duty cycle of about 30% instead of the normal 50%. Another device for reducing the duty-cycle during tune-up is a tuning-pulser.

If you want to operate with reduced power during good band conditions, first tune up your amplifier with normal drive power, then turn the microphone gain down to reduce power.

Class AB1 Design Considerations

Choosing the Optimum Value of Grid Terninating Resistance.

Tetrodes and pentodes require a peak RF drive voltage that semi-matches up to the grid bias voltage.

My strategy is to choose a value of grid termination resistance that roughly provides the needed peak RF drive V to the grid with exciters that develop 100v-p (100w rms) to 141v-p (200w rms) across 50 ohms. In other words, the goal is to match peak RF drive volts with the needed grid bias volts from the tetrode/pentode manufacturer's technical specifications.

-- For tubes that require 50v to 70v of grid bias, like the 4CX800A and 4CX1000A, a voltage-halving bifilar stepdown transformer driven 12.5 ohm grid termination is used.

-- For tubes that require 100v to 140v of grid bias, like the 4CX1500A, a directly-driven 50 ohm grid termination would be used.

-- For tubes that require 200v to 280v of grid bias, like the 4-1000A and 8169, a voltage-doubling bifilar stepup-transformer driven, 200 ohm grid termination is used.

-- For tubes that require greater than 300v grid bias, like the 8171, a trifilar stepup-transformer driven, 450 ohm grid termination is used.

However, if the peak grid V with max. drive is still a bit much, a cathode RF negative feedback reisistor (Rk) can be added to make up the difference. However, a trade-off is that the peak V drop across Rk normally subtracts from the screen to cathode V at the critical anode current peak. A workaround is to use the circuit shown in Figure 10.

Tuned Input Circuits for Neutralized, Class AB1 Grid-Driven Operation

Class AB1 grid-driven amplifiers look more complex than Class AB2 grounded-grid amplifiers. However, the tuned input circuitry for multi band Class AB1 grid-driven operation is comparatively simple.

The grid capacitance of tubes that are commonly used in Class AB1 grid-driven amateur radio power amplifier service ranges from about 15pF to 130pF. Since the capacitance of the grid is in parallel with the input, as frequency increases, input SWR worsens. This problem can be corrected by connecting a variable inductor in parallel with the grid. The inductive reactance {+j ohms} of the inductor is adjusted to cancel the capacitive reactance {minus j ohms} of the grid--thereby resonating the grid at the operating frequency. When the input SWR is tuned to minimum, the grid circuit is resonant. A simplified diagram is provided.

If the other end of the variable inductor is connected to a properly-adjusted capacitive voltage divider (connected between the anode and chassis ground), the amplifier is neutralized at whatever frequency the grid is tuned to. Obviously, this type of Class AB1 input circuit is a natural for continuous HF and MF coverage--just what's needed for operation on the 9 amateur bands below 30MHz. The ratio of the capacitances in the capacitive voltage divider equals the ratio of the feedback capacitance (the anode to grid capacitance) divided by the grid input capacitance. Typical ratios are 150 to 1 ... Achieving wide frequency coverage is not as easy in Class AB2 grounded-grid operation. A pi-network tuned input with the recommended Q of 2 has a limited bandwidth--so many, switched, tuned input circuits are required for wide frequency coverage.

Screen and Grid Supplies

There are many tetrodes and pentodes to choose from that are satisfactory for Class AB1 grid-driven operation. The essential criteria is that, with zero grid volts, the tube is capable of a peak anode-current that is at least triple its maximum (average) current rating. In most cases, this condition can only be met if near-maximum screen-voltage is applied. Relatively high screen-voltage is important because peak anode-current is a function of the screen-voltage raised to the 1.5 power.

For the best linearity, screen voltage should be regulated. For smaller tetrodes and pentodes, a Zener diode shunt regulator offers a good solution. Typically, a series of 10v to 30v, 5W Zeners are used. Screen voltage is adjusted by shorting out Zener diodes with a rotary switch. For larger tubes, an adjustable series-regulator is the best way to supply voltage to the screen. Thanks to modern power FETs and the venerable 723 IC linear regulator, building a reliable, regulated supply of 2kV or less is fairly simple.

Since the grid does not pass current in Class AB1 operation, there is no necessity to regulate the bias voltage. However, the bias supply should not have an extremely high output impedance. A maximum grid circuit R of 1k to 100k ohms is typically recommended by tube manufacturers.

Work Space

'Work-space' and 'head room' are terms that describe the range in which instantaneous anode-voltage is free to move up and down--thereby performing work. In a tetrode, at the maximum peak anode-current, to avoid excessive screen-current and a decrease in linearity, the instantaneous anode-voltage should not dip much below the screen-voltage. For example, a tetrode with a 4kV anode supply and an 700V screen supply, the work-space is approximately 4000V minus 600V = 3400V peak

In a pentode, the instantaneous anode-voltage may dip close to the suppressor-voltage--which is typically zero volts. In the above example with a screen-voltage of 800V, if the tube happened to be a pentode, the work-space would be around 3750V peak. Thus, pentodes enjoy slightly more work-space than tetrodes. As a result, pentodes are slightly more efficient than tetrodes. However, pentodes are more expensive than tetrodes because they are more complex to build. Sockets with low-L suppressor and screen bypass capacitors are needed for stable operation. Pentode sockets are not inexpensive. Another trade-off is that there are relatively few types of pentodes to choose from. A (if not the) suitable pentode for amateur radio Class AB1 grid-driven service is the 5CX1500.

Pentode Caveats

Pentodes typically have less feedback capacitance than tetrodes. This advantage theoretically makes pentodes more stable. Some designers do not neutralize pentodes because they feel the relatively low feedback capacitance between the anode and the grid is insignificant. However, for optimum linearity and stability, plus low input SWR, a pentode should be neutralized. This can easily be accomplished with the grid input circuit diagram [Figure 5] for Class AB1 tetrodes. To use this circuit with a pentode, DC-connect the suppressor to the cathode with a l0 or so ohm resistor. However, the suppressor must always be RF-bypassed to chassis ground to decrease feedback from anode to grid.

Screen Protection

Every screen type tube has a maximum screen dissipation rating in watts. If screen current times screen voltage exceeds this rating, the tube could be destroyed. This can easily happen with a no load or light load condition--so various protection schemes are used. If the anode voltage disappears while screen voltage is present, screen current will be excessive unless a means of protection is provided. Another hazard is reverse screen current. Reverse screen current can easily become a runaway condition. It happens virtually instantaneously. Reverse screen current is commonly experienced in Class AB1 operation. Unless bled off into a resistor load or into a shunt Zener voltage-regulator, reverse screen current can quickly destroy a tube. For tubes with screen voltages in the 300V to 800V range, a shunt regulator using a Zener diode string is a good solution. The Zener regulator string is connected through a high-R resistor to the anode supply. A sample circuit is provided. A suitable tube would be the 4CX1500B, or similar types.

Advantages of Shunt Zener Screen Regulation:

Limits the maximum current that can be drawn by the screen.

Protects against reverse screen current.

If the high voltage disappears, so does the screen voltage.

However, for larger tubes with higher screen current and screen voltage requirements, a Zener shunt regulator is somewhat impractical. A continuously-adjustable series-regulator screen supply is a better solution. To protect against reverse screen current, a shunt resistor/bleeder must be connected across the screen supply. A bleeder current flow of roughly 20% of the normal screen current seems to be adequate. 25% might be safer. To protect against excessive forward screen current, a fast acting fuse or magnetic-type circuit breaker is incorporated in the primary of the screen supply power transformer. An adjustable series regulator circuit is provided.

Grid-Driven Class AB1 Amplifier Neutralization and Tuneup

ßAdjusting a Class AB1 amplifier may look complicated at first, but after you have done it a few times, and you begin to understand the reason behind each step, it gets easier.

Neutralization: The goal of neutralization is to isolate the anode from the grid at the operating frequency. Neutralization discourages regeneration--oscillation. Neutralization usually needs to be adjusted only once.

1. Disconnect the amplifier from the electric-mains.

2. Temporarily disconnect the tank circuit from the HV blocking-capacitor.

3. Substitute a low-L film resistor, with the same R as the design anode-load [output] resistance, in place of the tank circuit. Typical values would be 1000 ohm to 4000 ohm, 2W. The resistor connects to the blocking-capacitor and to chassis-ground. Connect an RF-voltmeter or an oscilloscope equipped with a 10 to 1 hign impedance probe across the resistor.

4. Connect the amplifier to the electric-mains and turn on the transmit-receive relay power supply plus the grid and filament supplies. Do not turn on the screen or anode supplies.

5. Drive the amplifier with 20m or 15m RF. Tune the grid-circuit variable-inductor [L1] for minimum input SWR or minimum reflected power. If necessary, adjust the DC grid-voltage so that virtually no grid-current flows.

6. Adjust the neutralizing-capacitor (C3) for minimum RF-voltage at the anode-load resistor. If needed, readjust L1 for best input SWR followed by readjustment of C3. This completes the neutralizing procedure.

After C3 is nulled, the amplifier is neutralized for all bands. To confirm this, check the neutralization on another band. Readjust L1 for minimum SWR. The RF voltage across the output load resistor should not change appreciably. Typically, no further adjustment is necessary--even if the tube is replaced.

Remove the resistor and reconnect the tank circuit.

Tune-up.

1. Switch off the screen and HV anode supplies. Switch on the T/R relay supply, the filament supply and the grid supply.

2. Transmit on CW-mode into the amplifier and adjust L1, the grid roller-inductor, for minimum input reflected power. This tunes out the grid-reactance and simultaneously neutralizes the amplifier at the operating frequency. If you are using a transistor-output transceiver, to preclude SWR shutdown, initially tune the grid with no more than 5W of signal.

3. Apply full drive power using an electronic keyer sending dits at about 50wpm, or a use a tuning-pulser. Adjust the DC grid-voltage so that <0.1mA of grid-current flows. The grid-voltage is adjusted so that the grid is on the threshold of current flow. The grid-voltage adjustment is not used to set the zero-signal anode-current [ZSAC]--also known as 'idling current' or 'resting current'. Although the grid-voltage adjustment can discretely be used to make a small adjustment in the ZSAC, in Class AB1 operation, the primary criteria for setting the grid-voltage is that virtually NO grid-current flow with maximum drive. ZSAC is set by adjusting the screen voltage. Switch on the screen and HV supplies. Key the amplifier but do not apply drive power. Using the screen voltage adjustment, set the ZSAC as recommended by the tube manufacturer. For most tubes, the ZSAC should be about 20% of the rated anode current.

4. If a variable tank inductor and a variable tune capacitor is used, preset the tune capacitor and the tank inductor for the desired operating Q on the band in use. Preset the load capacitor and inductor by calculation. It is best to error on the side of too-little load C [heavy loading]. If too-light loading [too much load C] is used, excessive screen-current is likely. Remember that the tune C sets the operating Q. Most of the tuning should be done with the variable tank inductor. Fine tuning can be done with the tune C--but the final setting should no be very far from the setting for the correct operating Q.

5. When any amplifier is tuned-up, the anode-current must be driven to the maximum, peak, design value so that the tube's output load resistance will meet the design criteria for the pi output tank circuit. If a lesser current is used without proportionately decreasing the supply voltage, the output load resistance will be too-high and the subsequent adjustment of the tank will be for the incorrect output load resistance.

To be both linear and deliver good power output, an amplifier tube must be adjusted by loading it for the optimum peak anode-voltage swing. The indicated screen-current is an accurate way of tuning up a tetrode or pentode. If the anode-voltage swing is too great because of too-light loading, the screen-current [and distortion] will increase. This means that the instantaneous minimum anode-voltage is less than it should be--a situation which causes too many electrons to stick to the screen--thereby depriving the anode of electrons. If the screen-current is too low, the anode-voltage swing is inadequate--meaning that the loading is too heavy. This condition causes lower power-output. When the output tank circuit is tuned correctly, the screen-current meter peaks. This is done by adjusting the tune capacitor or by adjusting the tank inductor. Do not peak the screen current with the loading capacitor.

·· Thus, by using only the screen-current meter, tuning and loading can be adjusted for good linearity and good power output.

6. Set the transceiver to CW-mode. Apply full drive power. To reduce stress during tune-up, use an electronic keyer to send dits at about 50wpm. Standard dits are a 50% duty-cycle waveform, so current meter indications are roughly half of the actual value. A tuning pulser works even better. [Figure 9]

7. Peak the screen-current by tuning the tank inductor or the tune capacitor. If the screen-current begins to become excessive, stop short of the peak, increase loading and continue. If the screen-current is too low, lighter loading is needed.

--The last step is to re-peak the screen-current with the tank inductor or the tune capacitor.

Loading for slightly less screen-current increases linearity with the trade-off of slightly less power output.

·· It is useful to keep a log of the various final settings for different frequencies. This saves time during future tune-ups.

For thoriated-tungsten cathode tubes only: - - While sending dits at full power, gradually reduce the filament-voltage until the relative output just begins to decrease. Increase the filament-voltage about 2%. This is the optimum filament-voltage. This should be rechecked every few hundred operating hours. The same thing applies to grounded grid amplifiers. For indirectly heated cathode tubes, like the 8877, the ideal filament voltage for communications services is near the minimum filament voltage rating. Under no circumstance should such a tube be operated at less than the minimum filament rating.

Distortion

Perfectly linear amplification produces nothing except a larger representation of the input signal. Non-linear amplification produces mixing--and mixing creates distortion products.

Inter-modulation distortion [IMD] is the result of mixing between two or more input signals. The human voice produces many frequencies at any instant. When voice modulation is amplified non-linearly, many mixing products are produced. This is called "splatter" or, more descriptively, "rotten splatter." IMD is usually measured by simultaneously applying two equal-amplitude, not harmonically related modulation frequencies such as 2000Hz and 2200Hz. When two or more frequencies mix they produce spurious signals at their sum and their difference frequencies--in this case 4200Hz and 200Hz. The first level of mixing produces what are called "third order products." Additional products are produced by third order products mixing with the two fundamental frequencies. For instance, 2200Hz and 4200Hz mix to produce a signal at 6400Hz.

When distortion products are inside the fundamental pass band of an AM or SSB transmitter, audible distortion results. This gives voice modulation a rough, unpleasant characteristic that reduces intelligibility. Odd-order distortion products which lie outside the pass band can cause interference on adjacent frequencies.

There are two methods of referencing IMD measurements. In method A, the IMD power level is referenced to either one of two equal-amplitude input signals. The power ratio of PEP to either of two equal-amplitude sine waves is four to one [6db]. In method B, the IMD level is referenced to the PEP level. Thus, an IMD level of minus 34db using method A equals an IMD level of minus 40db using method B. Amateur radio operators tend to use method B because receiver S-meters respond to PEP. In commercial radio, the military, and the FCC--where distortion measurements are typically made with a spectrum analyzer--method A is used. When using a spectrum analyzer, distortion can be broken down further into third order products, fifth order products, seventh order products. However, total IMD referenced to PEP is a more significant number.

It is possible to measure IMD without expensive laboratory equipment. All that's needed is a receiver and some understanding of what's required to make a meaningful measurement.

By comparing the signal strength in the transmitter's fundamental pass band window with the signal strength in the adjacent pass band windows, IMD can be measured fairly accurately--even over the air. The amount of receive frequency offset is critical. If the receive pass band is too close to the transmitter's fundamental pass band, the receiver will not be able to separate the IMD energy from the fundamental energy. As a result of this overlap, the distortion measurement will be higher than the actual amount. If the receive frequency offset is too far from the fundamental pass band, the receiver's pass band will not receive all of the IMD--and the distortion measurement will be lower than the actual amount.

For a receiver with two, cascaded SSB filters, a receive offset of 3.6kHz is about right--provided that the receiver is set to the same sideband as the transmitter. For a receiver with one SSB filter, an offset of about 4.5kHz is needed. To measure the IMD level of a LSB signal, offset LSB-receive higher in frequency. For measuring the IMD from an USB signal, offset USB-receive lower in frequency.

Since very few S-meters are linear, a calibration chart of S-meter readings versus decibels is a prerequisite for making accurate measurements. A calibration chart can be made with a step-attenuator and a signal source, or with a signal generator/attenuator.

In order to measure IMD, at least two modulation frequencies are required. Human speech is a good signal source for measuring IMD because, at any instant, speech contains many fundamental frequencies and harmonics. As its name suggests, another harmonic-rich signal source is a harmonica. By simultaneously blowing into two or three adjacent holes at the low note end, a plethora of frequencies can be produced that are optimal for making distortion measurements.

Splatter Reports

Before reporting splatter, it is important to keep in mind that all SSB, DSB, and AM signals have IMD. In other words, everybody splatters. The obvious question is how many decibels down is the IMD? Minus 40db is excellent; minus 30db is objectionable; minus 20db is abundantly abominable. With one exception, FCC rules allow virtually any level of IMD inside the ham bands. The exception is when IMD causes harmful interference to emergency communications. Splattering on non-emergency communications is NOT considered to be harmful.

Before reporting a station's level of IMD, it is advisable to determine whether or not the station operator is interested in hearing your report. Although most amateur radio operators are interested in transmitting a high quality signal, some operators deliberately misadjust their equipment to maximize IMD.

Notes on Measuring Power

Since E-peak = E-rms x 2^0.5, and P = E^2 ÷ R, at its crest, the instantaneous peak power in a sine wave is double the RMS power. A common unit of measuring amplifier output is the PEP [peak envelope power] watt. Despite the name, peak envelope power watts are not peak watts--they are RMS watts at the crest of modulation. If an amplifier was powered by a regulated anode supply, there would be virtually no difference between PEP watts and AØ [NØN] watts. In a typical amplifier, the anode-voltage sags appreciably under AØ conditions--so PEP watts are typically about 20% higher than AØ watts. PEP need not be measured with voice modulation. PEP can also be measured by keying the driver at 30 pps with a steady string of pulses that approximates the duty-cycle of a human voice--roughly 30%.

Tube Ratings

Traditionally, amateur radio operators have taken a cavalier attitude toward tube manufacturer's ratings. While some ratings can be exceeded judiciously, exceeding other ratings can be costly. Examples of ratings which should not be exceeded for indirectly-heated cathode tubes are minimum filament-voltage and maximum anode-current. Violation of either rating can result in destruction of the delicate cathode. Directly-heated cathodes are more rugged. The maximum anode-current rating for directly-heated cathode tubes is a linearity issue--not a cathode destruction issue. One rating which should not be exceeded is maximum seal temperature. It has been said that the way to tell when the blower is too big is if it blows the tube out the socket.

END OF PART 4

AG6K Part 3

2026-08-20T04:30:14.406-07:00

Changing Voltage

It's nice to have the ability to reduce the output power from an amplifier. One way to do this is to reduce the anode-voltage and anode-current simultaneously so that the output load-R of the amplifier tube or tubes does not change appreciably. This allows the tank circuit to function at its design Q for both high and low power.

Switching primary taps is not an efficient method of reducing output voltage because in order to do so extra turns must be added to the primary. To make room for the extra turns, the primary's wire diameter must be decreased--and that increases R. An efficient method of reducing the DC output voltage in a HV power supply is by switching secondary taps on the transformer. If a fairly ordinary ceramic rotary switch is insulated from the chassis, it can easily perform this job. The taps should not be switched under load.

If no secondary tap is provided on a transformer, it is possible to lower the output voltage 50% by switching from fullwave doubler to fullwave bridge rectification. All that's needed is a suitable SPST vacuum relay, or well-insulated ceramic switch, two filter capacitors and four strings of rectifiers. For example, a power supply that produces 4000V for SSB could be operated at 2000V for RTTY, CW, or FM. The DC output current capability doubles when the output voltage is halved during fullwave bridge operation--just what's needed for FM's and RTTY's much higher duty-cycle. When switching the voltage output, it is best to temporarily switch the power supply off and then restart. The output voltage may be switched down without switching the supply off--provided that the amplifier is in standby.

Variable Electric-Mains Transformers

On paper, the variable auto-transformer, a.k.a. Variac® or Powerstat®, looks good. Variacs/Powerstats are intended to be used with resistive loads. When a Variac is set at or near 100% of the input voltage, it adds only a small amount of series R. However, when a Variac is set to produce a fraction of the input voltage it adds more series R. This is of little consequence with resistive loads. However, when the load is a capacitor filter DC supply, due the demand for high peak current, additional series R is most unwelcome. Although Variacs perform acceptably with resonant choke filter power supplies, using a Variac to control the output voltage from a capacitor filter supply is not good engineering practice.

A Variac can be used in place of a step-start relay. Provided that the operator always remembers to set the Variac to near-zero before switching the amplifier on, all will be well. A step-start relay offers some advantages: it is cheaper, mistake-proof, saves many kilograms, and it adds substantially less series R.

There is, however, an appropriate step-start application for a Variac. Eimac® recommends using a motor-driven Variac, feeding the filament transformer primary, to bring up and bring down the filament-voltage (over a period of two minutes each) on its tubes which incorporate water-cooled filament supports. An example is the 8973 tetrode--just what you need if you are building a 600kW linear amplifier.

Transformer Insulation

Most transformers use paper to separate and insulate each layer of windings. Paper is hygroscopic--i.e., it absorbs water vapour from the air. The presence of water reduces the insulating ability of the paper. In time, insulation breakdown is likely. The solution is to pot the windings. Plastic resins are best. Petroleum tar is next best. Since potting fills up the air spaces in the windings--and air is a poor heat conductor--potting also improves heat transfer--thereby reducing internal temperature and increasing MTBF. Potting adds very little to the initial cost of a transformer and subtracts substantially from the long-term cost. Some custom transformer manufacturers offer potting as an extra-cost option. Peter Dahl Co. has a potting option.

Transformer Potting

Commercial transformer potting is normally done in a vacuum chamber to facilitate the evacuation of air bubbles. However, with a little patience, it is possible to pot transformers satisfactorily without special equipment. Bake the transformer in an oven at a temperature of about 175 degrees F/80 degrees C. Bake for two to three hours per pound. Baking drives out internal moisture. After baking, place the transformer on a table covered with a thick layer of newspapers. Position the transformer so that the leads or lugs are down. Using masking tape, seal the end of the transformer windings opposite the leads/lugs so that liquid can not escape easily when the transformer is inverted.

Polyester fiberglass laminating resin is designed to flow into small spaces and expel air bubbles. It can be used for potting transformers.

In a clean tin-can, pour in a quantity of laminating resin that will fill up the air spaces in the bottom 5% of the transformer's windings. Using pliers, bend the rim to facilitate pouring the resin. Mix in about 5 drops of catalyst per ounce of resin. Depending on the ambient temperature and humidity, this amount of catalyst will result in a moderately fast gel time. Pour the resin slowly into the windings. Resin pouring should be done steadily and from only one area of the windings to avoid trapping air bubbles. Any leaks from the bottom can be patched by forcing raw silicone rubber into the area of the leak. When the resin gels, it forms a thin bottom plug. The bottom plug need not be more than about 5mm thick.

Pour an amount of resin into the can that will fill the remainder of the windings. For a several-kVA transformer, use about 1.5 drops of catalyst per ounce of resin. For smaller transformers, slightly more catalyst is needed. The resin must not gel before the air bubbles have had a chance to escape--so it is better to err on the light side for the amount of catalyst.

Heat increases the fluidity of the resin--hastening the exit of bubbles. However, heat tends to decrease gel-time. Internal transformer heating is accomplished by forcing current through the windings with a Variac. Connect the Variac to the highest voltage winding. Short the highest current winding with an AC ampmeter. Increase the voltage until the ampmeter indicates the rated winding current. At this level fairly normal internal heating results. As soon as the resin begins to gel, stop the current and direct a cooling fan at the transformer. Resin-gelling is an exothermic reaction.

Rectifiers

The most frequent failure mechanism for HV power supply rectifiers is too much reverse current. This problem can be virtually eliminated in 50Hz/60Hz, fullwave bridge and fullwave doubler, capacitor filter circuits if the total PIV in each string of diodes exceeds the no-load DC output voltage by at least 50%. For operation in high-temperature environments, a 100% factor may be needed.

Modern solid-state rectifiers are made differently than they were 30 years ago. In those ancient times, same-type rectifiers did not have uniform reverse characteristics. Rectifier failure was common. In an attempt to compensate for the inherent weaknesses in early solid-state devices, rectifier protection schemes were used. Resistors and capacitors were paralleled with series rectifiers--probably a take-off on the practice of using equalizer resistors on electrolytic filter capacitors. However, in any series circuit, the currents in all of the elements are exactly equal. Thus, when rectifiers are in series, the reverse current burden is exactly the same for each rectifier--provided that no parallel resistors are used. Manufacturers of series rectifier units long ago abandoned the practice of using parallel resistors and capacitors. The 1995/6/7 Radio Amateur's Handbook explains why rectifier 'equalization' is prone to cause premature rectifier failure.[page 11-9, middle column, top]

Series-connected rectifiers should be of the same type. Mixing rectifiers types in the same series string could cause a problem during the reverse half-cycle.

When a rectifier has been conducting, it takes a finite amount of recovery time for the rectifier to stop conducting after the source of forward current reaches zero. It is important that a rectifier not be conducting when the reverse voltage arrives. This can be a problem when rectifying high frequency AC or when rectifying square waves. Paralleling a capacitor with each rectifier may help the rectifiers to stop conducting sooner. If you need to rectify high frequency AC, one solution is to use fast-recovery epitaxial rectifiers. 1000PIV, 1A, 70 nanosecond recovery time units are currently priced at about 50¢ each in quantities of 100.

Packaged HV Rectifiers

Rectifiers that have a rating above 1kV PIV are typically made from a series of individual rectifiers that are entombed in an epoxy package. This arrangement makes for a neat-appearing installation--but there is a trade-off. Epoxy is a poor conductor of heat. Individual series-connected diodes mounted on perfboard and exposed to open air dispose of heat much more efficiently than do multi-diode packages.

Filter Capacitors

Filter capacitors usually have a ripple-current rating. The ripple-current rating should be at least equal to the maximum DC output current capability of the supply. Quality filter capacitors are designed to minimize equivalent series resistance [ESR]. Low ESR ohms translates into a high ripple-current rating.

Oil-filled capacitors are available in two types: filter service, for use in power supplies, and flash service for use in photography or pulsed laser applications. The flash capacitor is designed for maximum capacitance per unit volume. To reduce volume, very thin metal foil is used to make the plates of a flash capacitor. Thin plates have more ESR--so they dissipate more I-squared times R power when they are subjected to ripple-current.

For longest life in high duty-cycle applications, cool air should be allowed to circulate freely around filter capacitors.

According to some manufacturers, flash capacitors can be used in filter service if they are operated at 60% of their rated peak volts. In intermittent duty applications, it may be possible to use flash capacitors at more than 60% of their rated peak-voltage. To discover how a flash-capacitor is faring in ripple-current service, after about an hour of contesting, if the capacitor is warm to the touch, an internal heating problem is indicated. Internal heating causes expansion and stresses the capacitor's case--which may eventually come apart at the seams and begin to leak dielectric oil.

If not plainly stated on its label, there is a way of determining the intended type of service for an oil-filled capacitor. Flash-capacitors usually have a peak voltage [PV or VP] rating. Filter-service capacitors are usually rated in DC working volts [DCWV]. Capacitors can also be rated in AC working volts. To convert AC-working volts to DC-working volts, multiply the AC voltage by three.

There have been instances where surplus flash capacitors were offered for sale with altered or counterfeit markings. For example, a capacitor that was originally marked "3.5kVP" was changed to "5kV" by erasing characters. Thus, a capacitor that should have been de-rated to 0.6 x 3.5kV = 2100V for filter service would appear to be good for 5kV. A practical way of determining whether an oil-filled capacitor can withstand ripple-current is to connect it in series with an AC-ampere meter and an AC voltage source. The voltage is adjusted until the AC current is equal to the expected maximum output current of the power supply. If, after an hour, the capacitor shows little or no internal heating, you have a winner.

There is also a flash service type of aluminum electrolytic capacitor, that is not designed to handle ripple-current.

Electrolytic filter capacitors are intolerant of reverse current and heat. Electrolytic capacitor working voltage [WV] ratings should be treated with respect. The WV rating is virtually the maximum voltage rating. Despite their more delicate nature, electrolytic filter capacitors offer substantial advantages over oil-filled filter capacitors. The main advantages are more joules of energy storage per dollar, reduced weight and reduced volume.

When electrolytic capacitors are operated in series, they should share the voltage equally. In order to do this, a voltage equalizer resistor is connected across each capacitor. Equalizing resistors must have fairly equal resistance--and their resistance should not change appreciably during aging. If an equalizer resistor changes value appreciably, domino-effect destruction of an entire section of filter capacitors may result.

There is no formula for determining the optimum resistance for an equalizer resistor. Less resistance equals less bleed-down time. However, less resistance produces more heat. A compromise is in order.

Carbon-composition resistors change resistance with age. This characteristic is unacceptable for equalizer resistor service. High resistance, wire wound resistors are wound with extremely fine resistance wire. They are not remarkably reliable. Metal oxide film [MOF] resistors are more reliable. The initial resistance of a MOF resistor is typically much closer to the labeled value--and it will stay that way for many years. A Matsushita/Panasonic® 3W, 100k ohm MOF resistor makes a good equalizer resistor for 450V capacitors. It produces a reasonable bleed-down time and a reasonable amount of heat. These resistors are available from Digi-Key.

Electrolytic filter capacitors are ruined quickly by reverse current. Reverse current often occurs when a rectifier fails. To protect electrolytic capacitors from reverse current, connect a >600PIV diode across each capacitor. The cathode band of the diode connects to the capacitor's positive terminal.

Biasing

When a grounded-grid amplifier's operating bias is obtained from a single Zener diode, there is no way to compensate for tube variation. One solution is to obtain the operating bias from a series string of forward biased rectifier diodes. By switching the number of diodes in and out with a rotary switch, the bias can be changed in approximately 0.7V increments.

Traditionally, a mechanical relay has been used to switch amplifier bias between receive and transmit. An optoisolator coupled to a transistor switch, i.e., an electronic bias switch, can do this job faster, more reliably, sans-noise, and cheaper.

There are principally two means of actuating electronic bias switches--RF-actuation and coil-current actuation. Although it sounds hip, RF-actuation creates two problems. The amplifier rapidly switches between linear bias and non-linear bias during softly-spoken syllables of speech. This causes choppy-sounding audio and splatter. When the electronic bias switch is controlled by the current that passes through the RF relays' coils, it is not possible to intermittently switch the amplifier into non-linear bias during transmit. Coil current actuated bias switching can be accomplished with an optoisolator. The optoisolator's input LED is driven by the coil-current. The output of the optoisolator drives the bias switch transistor.

Class AB1 Grid Bias

Since the grid draws virtually zero current, it is easy to make the bias continuously adjustable in Class AB1 operation. Typically, the cutoff bias voltage during receive will be about 50% higher than the transmit bias voltage. An optoisolator driving a HV FET can be used to switch the bias between transmit and receive. A circuit is provided.

High Speed RF Switching

A conventional relay switches in roughly 25mS. Such relays have traditionally been used for RF and bias switching in RF amplifiers. This was acceptable when transceivers also used conventional relays. Currently manufactured transceivers are designed for AMTOR, QSK telegraphy, and unobtrusive SSB VOX operation. Modern transceivers T/R and R/T switch quietly, and do so in as little as 5mS. Such radios typically use a high-power SPDT reed relay to switch the antenna between transmit and receive. Similar relays can be used for amplifier input RF switching. Jennings and Kilovac manufacture high speed, SPDT vacuum-relays that have a continuous rating of 7A at 32MHz [2450W into 50 ohms]. The Jennings relay is the RJ-1A. Kilovac's relay is the HC-1. When used with a speedup-circuit, either relay can switch in under 2mS. Although both manufacturers make DPDT RF vacuum-relays, none are as speedy as their fastest single pole models. Thus, separate input and output relays are usually faster than a single DPDT relay.

PIN Diode Switching

Another device that can be used for high-speed RF switching is the PIN [P-Intrinsic-N] diode. PIN diodes are similar to 1000PIV rectifier diodes--i.e., they have a wide intrinsic region. PIN diodes are utilized extensively in radars as transmit-receive switches.

A PIN diode is switched off by applying DC reverse voltage to widen its intrinsic region. The PIN diode is switched on by passing DC current in the forward direction to fill its intrinsic region with current carriers. PIN diodes are extremely fast switches. Their lifetime is virtually unlimited as long as the allowable PIV is not exceeded.

The typical reverse breakdown voltage rating for a PIN diode is around 1000V. A legal-limit amateur radio amplifier produces an output voltage of about 800 volts peak-to-peak [p-p] into a 50 ohm load--so a 1000PIV PIN diode is more than adequate. When the load Z is higher, due to a somewhat less than wonderful SWR, the switching device may be exposed to more than 1000Vp-p. This poses no problem for a typical high-speed vacuum-relay. Even if a vacuum-relay's breakdown voltage is temporarily exceeded, there is little likelihood that permanent damage to the vacuum-relay's contacts will result. However, solid-state devices are not so forgiving. A single voltage-transient can destroy a PIN diode.

For 100 WPM computer-CW, the PIN diode is clearly the only choice. For 30 WPM CW, AMTOR and high-speed VOX, a vacuum-relay has advantages.

Solid-state Component Ratings

Different types of solid-state components are rated somewhat differently. Some ratings are realistic. Some ratings are not realistic. The maximum ratings of large transistors and large Zener diodes can not be realized unless drastic, extreme measures are used to keep the case temperature below the maximum allowable 25 degrees C at full ratings. In the real world, operation at 30% of a published dissipation or current rating is usually safe. Additionally, bipolar power transistors suffer from a generic weakness called secondary-breakdown phenomenon. For example, a "1500V, 8A, 150W" power transistor may only be able to safely dissipate 15W at moderate collector-to-emitter voltages. T-MOS power FETs are much more resistant to secondary-breakdown.

Wire-lead rectifier current ratings are fairly realistic when they are mounted on perfboard and cooling air is allowed to circulate freely around individual rectifiers.

Measuring PIV

There is some variation in the inverse voltage capability of solid-state rectifiers of the same type. Measuring the avalanche voltage of each rectifier diode is a not a bad idea. When a diode's inverse current is just below the rated maximum, the voltage across the diode is the avalanche voltage. Exceeding this voltage is likely to be fatal.

Breakdown Voltage Testers

A breakdown voltage tester (a.k.a. high-pot) could be described as a variable HV Ohmmeter that does not read directly in ohms. It is a useful tool. Breakdown testers are essential for testing vacuum relays, vacuum capacitors, blocking capacitors, air-variable capacitors, rectifiers, and for finding problems with insulation. Building or troubleshooting a tube-type RF amplifier without a breakdown tester is like crossing an ocean without a navigation instrument. For most amateur radio applications, the highest voltage component rating commonly encountered [with vacuum-capacitors and vacuum-relays] is 15kVDC/9kV RF peak, so a 0 to 15kV breakdown tester should suffice.

Although commercial breakdown testers are available, they are not cheap. A suitable breakdown tester can easily be constructed from mostly-surplus parts. The main parts are a 50/60Hz low-current HV transformer, a >1A, 0 - 120 V variable transformer (Variac®), a 120V incandescent bulb, some diodes, resistors, a sensitive uA meter and two HV filter capacitors.

Commercial, low-current HV DC supplies may also be used provided that they are connected to a Variac in series with a 120V incandescent light bulb to limit current. The bulb limits the short-circuit current to a safe value--obviating the need for a fuse. The wattage of the bulb is roughly proportional to the wattage of the supply. The rated [I=P/E] bulb current should be similar to the appropriate fuse rating for the HV supply primary. A multi megohm resistor is used to limit the current flow into the device under test. The uA meter should be protected with back to back 1A diodes. A circuit is provided.

Step-Start

Most power supplies benefit from something to soften the shock of start-up. A 10A DPST-NO or 10A DPDT relay and two approx..25 ohm 10W resistors are just about all that's needed to add a step-start circuit to the average 1500W amplifier. The step-start circuit goes in series with the mains fuses or circuit breakers. With this arrangement the filaments, the HV supply, the on/off switch, and the LV supplies enjoy the benefit of a gentler start-up.

VHF Stability

Every HF amplifier has at least two resonant circuits in its output circuitry. The more obvious one is the HF-resonant tank circuit. A less obvious one is the VHF-resonant circuit that is principally formed by the anode capacitance and the inductance of the conductors between the tank circuit and the anode. In 1500W amplifiers, anode resonance typically occurs around 100MHz--well within manufacturers' ratings for "Amplifier and Oscillator Service" for the tubes that are commonly used in such amplifiers.

The equivalent resistance of a high Q parallel resonant circuit is virtually infinite. A low Q parallel resonant circuit has a relatively low equivalent resistance.

The voltage gain of an amplifier tube is roughly proportional to the load resistance. High load resistance produces more gain. Low load resistance produces less gain.

If the conductors in the anode resonant circuit have a high VHF Q, the equivalent load resistance presented to the anode will be high and the tube will exhibit increased voltage gain at the VHF resonance. If the conductors in the anode resonant circuit have a low VHF Q, the load resistance presented to the anode will be low and the voltage gain at the VHF anode resonant frequency will be reduced. Of course, if no VHF energy were present, it would make no difference how much VHF gain an HF amplifier had.

When a transient current passes through a resonant circuit, the resonant circuit rings like a bell--producing a damped sine wave signal. This is how ancient spark transmitters produce RF--and the larger ones produced many kilowatts of it.

Whenever the anode-current in an HF amplifier changes, a small VHF damped sine wave signal is produced in the anode's VHF resonant circuit. This signal can be observed with a VHF oscilloscope or a spectrum analyzer. The amplitude of the RF voltage produced is proportional to the Q of the anode resonant circuit. If none of this damped wave signal were fed back to the input, there would be no problem.

In a grounded-grid amplifier, the grid appears to shield the input from the output. In a grid-driven Class AB1 amplifier, The RF-grounded screen appears to shield the input from the output. However, no grid and no screen is perfect--so some of the damped wave VHF signal at the anode is capacitively fed back to the input--and amplified.

Although it's unlikely, if the phase and amplitude of the damped wave signal happens to be just right, oscillation at the anode's VHF resonance can occur. If the VHF energy that is produced could find its way to a load, no danger would be posed by a VHF oscillation. However, the HF tank circuit is a low pass filter that effectively blocks VHF energy. Thus, the oscillator is unloaded and the resulting grid-current is very high. The unloaded condition can cause VHF voltage transients in the anode circuit. These transients may cause tune-capacitor arcing and band switch arcing across open contacts. Since they are closest to the anode resonant circuit, open tune-capacitor padder contacts, as well as open 10m contacts are most vulnerable to parasitic-instigated arcing. Band switch contacts can be melted and/or vapourized by such occurrences.

Parasitic Suppression

On page 72, the 1926 Edition of The Radio Amateur's Handbook tells us how to build an improved VHF parasitic suppressor The logic was elementary. A suppressor is supposed to dampen the anode circuit. Since low Q is synonymous with high dampening, why not decrease Q by using resistance-wire? Quoting from page 72:........ "The combination of both resistance and inductance is very effective in limiting parasitic oscillations to a negligible value of current."

After 1929, someone forgot to include this information in the Handbook. In those days, the oversight probably didn't matter very much. Large amplifier tubes generally had low VHF amplification, so VHF instability was not a major issue. During the ensuing decades, people got into the habit of using parasitic suppressors made from copper or silver-plated copper. This was an easy habit to get into since copper and silver can be soldered more easily and cheaply than nickel/chromium [nichrome] resistance wire. Meanwhile, the performance of amplifier tubes kept improving. Because of these improvements, modern high-amplification tubes appear to benefit more from 1926-vintage low VHF-Q parasitic suppressors than 1926-vintage tubes. NOTE: In 1926, a 'high-mu' triode had a mu of around 40.

Anti-parasitic Techniques

Low VHF-Q conductor material can be used to increase VHF loading in the anode resonant circuit. Nickel-chromium-iron alloys are best, Nickel-chromium (nichrome) and some types of stainless-steel are almost as good. The use of copper, aluminum, and silver should be kept to a minimum. However, good conductors are desirable beyond the tune capacitor, which marks the end of the anode VHF resonant circuit and the beginning of the HF tank circuit.

Output Z: The output impedance of most tubes is a matter of kilo-ohms--not ohms. There is no scientific reason to use "heavy duty" conductors between the anode and the tune capacitor. If good VHF stability is a design goal, it's best to use conductors that are no larger than is necessary to carry the highest current present, i.e., the 10m RF circulating current between the anode (output) capacitance and the tune capacitor. Round conductors have a lower VHF Q than flat conductors. To increase current handling ability, or to reduce inductance, two paralleled round conductors, separated by a wide air-gap, are better than a flat conductor of the same overall width.

When designing the layout for an HF amplifier, locate the tune capacitor fairly close to the anode. This reduces the inductance in the anode resonant circuit--increasing the VHF resonant frequency. If the distance from the anode to the tune capacitor is grossly excessive, the 3/4 wave anode resonance may cause instability problems--especially with tubes that are UHF-rated like the 8877, 8874, and 3CX800A7.

In order to prevent a conductor from sharing the anode VHF-resonant circuit with the HF tank circuit, connect the tank inductor directly to the tune capacitor. It is best not to connect the tank inductor directly to the blocking capacitor.

The output enclosure in an HF amplifier is a high Q VHF-resonant cavity. The output enclosure can become a player in parasitic oscillations. Cavity dampening may be needed. This is a common practice in commercial high-power amplifiers. Closed loops of resistance-wire can be used to dampen cavity resonances.

In some cases, the HV-RFC contains a VHF resonance that abets parasitic oscillations. This problem is indicated by several burned turns appearing on the HV-RFC after a glitch. To isolate a problematic HV-RFC VHF resonance, place at least one VHF attenuator-rated ferrite bead on the lead between the top of the HV-RFC and the anode circuit.

Designing VHF Parasitic Suppressors

The simplest type of parasitic suppressor is a resistor. It reduces Q by adding R. This technique is effective. However, it is mostly limited to low power applications. The traditional staggered-resonance parasitic suppressor provides two advantages over a resistor suppressor--it can handle more current, and it causes the VHF resonance to work against itself.

A staggered-resonance parasitic suppressor typically consists of a coil inductor paralleled with a low-inductance resistor. The axis of the coil inductor is parallel to the resistor. Here's how it works: The magnetic field from current flowing in the resistor is at a right angle to the direction of current flow. The magnetic field from the inductor is parallel to the direction of current flow. Because the two magnetic fields are 90 degrees apart, the inductances act independently instead of mutually. The two independent-inductances connect to one fixed-capacitance--i.e., the anode. Since the coil has more inductance than the resistor, it creates a second VHF resonance that is slightly lower in frequency than that produced by the resistor. The conflicting resonances work against each other. This technique broadbands the anode's VHF resonance, i.e., it reduces the Q--very much like stagger-tuning IF transformers to widen a receiver's band pass. Reducing VHF Q lowers the parallel equivalent VHF load resistance on the anode. This reduces the VHF voltage gain--and that reduces an amplifier's ability to oscillate at VHF.

Choosing the optimum amount of inductance for a suppressor inductor [Ls] is best determined experimentally by operating the amplifier on 10m. Since 10m is almost VHF, a device that suppresses VHF energy should get hot from 10m RF. If Ls is too small, the suppressor resistor [Rs] will not exhibit visible signs of heating during operation on 10m. If Ls is too large, there will be too much voltage drop on 10m and Rs will burn out.

Every straight conductor has inductance. The amount of inductance is fairly proportional to length. Manufactured high-power "non-inductive" resistors are long--and therefore somewhat inductive. They are too inductive for use in VHF suppressors. However, it's easy to make a suppressor resistor of sufficiently low inductance by paralleling straight nichrome wires that are separated by air-gaps.

Staggered-resonance suppressors can be built without a resistor by paralleling two unequal-inductance nichrome wires. For example, a silver plated strap in the anode circuit can be changed from a potential source of grief into a Q-reducing asset by replacing it with two, parallel, nichrome wire conductors. One of the conductors is made about 25% longer than is necessary to span the distance. Its length is shortened by winding a small 1 to 2 turn coil. The axis of the coil is parallel to the shorter wire. This arrangement decouples the magnetic field in the coil from the magnetic field in the straight conductor.

For large amplifiers, 100% nichrome staggered-resonance suppressors solve the problem of not being able to find a high-power resistor of sufficiently low inductance for use in parasitic suppression service. For very large HF amplifiers, all-nichrome staggered-resonance suppressors should be made from flat nichrome conductors in order to carry the large RF circulating current between the tune capacitor and the anode capacitance.

In two-tube amplifiers, if the two suppressors are allowed to magnetically couple to each other, a VHF parasitic oscillation may occur. In a two-tube amplifier, the suppressor coils should be positioned at a right angle. If the suppressor coils are parallel to each other, the coils should be wound in opposite directions, and separated as much as is practical.

Evaluating VHF Parasitic Suppressors

Some amateur radio operators--and some electronic engineers--do not believe that VHF oscillations can take place in an HF amplifier. This is understandable because the most common and most destructive type of VHF-parasitic-oscillation, the push-push variety, lasts only a matter of microseconds. Push-pull VHF parasitic oscillation is obviously only possible in multi-tube amplifiers. A steady oscillation between anodes is the result. Push-pull parasitic oscillation is characterized by extremely high anode dissipation, moderate grid and anode currents with zero drive, and no arcing. Push-pull parasitic oscillations can be stopped by switching the amplifier to standby. This is not possible with a push-push parasitic oscillation---wherein the event is probably over before the sound of the parasitic arc reaches the operator's ears.

VHF parasitic oscillations are not cooperative. It may take a particular sequence of anode-current transients to initiate a parasitic oscillation. Even though there is no concrete scientific evidence to prove it, the phrase "CQ contest" may have a propensity to produce the key sequence of anode-current transients--especially if the contest is one you've been waiting for, and the local radio parts emporium has just closed for the weekend.

A major factor with parasitics is the VHF-gain of the particular amplifier-tube or tubes that happen to be installed in the amplifier. Even among new amplifier-tubes from the same production lot, there is some variation in VHF gain. Tubes with below average VHF gain may never have a parasitic-oscillation--no matter how poorly their parasitic-suppressors perform. Thus, when below average gain tubes happen to be installed in an amplifier, it's easy to assume that the amplifier design is perfectly stable.

Since catching a parasitic oscillation in the act is virtually impossible with ham-type test gear, a different analytical approach must be used. It's a reasonable assumption that a resonant circuit which supports parasitic oscillation can be found and evaluated with a dipmeter.

To determine the parasitic frequency and evaluate the parasitic suppressor, unplug the HF amplifier from the electric-mains and measure the anode-resonance with a dipmeter. The best place to do this is on either side of the HV blocking capacitor. The resonant frequency typically varies inversely with the amplifier's power capability. 700W single-tube amplifiers typically resonate from 100MHz to 150MHz. 1500W amplifiers typically resonate from 80MHz to 140MHz. 100kW amplifiers typically resonate from 35MHz to 45MHz. You should be able to tune the resonance a few MHz by adjusting the tune capacitor. The VHF dip in some amplifiers' anode-circuits is so sharp that it will "suck-out" the oscillator in the dipmeter. If that is the case, the dipmeter must be backed away (decoupled) from the conductor to accurately observe the dip. A broad, smooth dip is good. A sharp dip indicates that the anode-circuit has a high VHF-Q. This is not good news unless you happen to need a VHF oscillator.

If the suppressor design is changed to (hopefully) lower the VHF Q, check the dip again. The frequency will usually not change appreciably but the dip should now be smoother, more broad, and it should be necessary to couple the dipmeter coil closer to the anode-circuit to achieve the same degree of dip.

If you make an experimental change to a parasitic suppressor, and you want to evaluate the change more precisely, use a plastic ruler to measure the distance from the anode circuit to the tip of the dipmeter coil that will result in a 20% dip on the dipmeter. If the coupling distance required for the 20% dip decreases after a change to a suppressor, a lower VHF-Q is indicated and the change was obviously an improvement. If the dip distance increases, the VHF-Q went up and the change in the suppressor was a step backward.

Amplifiers that exhibit a tuning vagary in the area of resonance on one or two bands are probably in need of better parasitic suppression. A stable amplifier usually exhibits smooth, symmetrical tuning.

Additional information on parasitic oscillation was published in the September and October 1990 issues of QST.

END OF PART 3

AG6K part 2

2026-08-20T04:29:36.938-07:00

Grid Protection in G-G Service

Transmitting AØ for long periods on 10m with the load capacitor set for maximum C would result in very high grid-current and almost no RF output. Under such a condition it might be possible to overheat a grid. However, since most people tune an amplifier for maximum output--and maximum output virtually coincides with normal grid-current--very few people are likely to overheat a grid. Thus, complex electronic grid-protection circuits are seemingly unnecessary.

A disadvantage of electronic grid-protection circuits is that they are not effective against the most common source of grid damage--intermittent VHF parasitic oscillation.

Glitch Protection

During a major problem, the anode (plate) current meter and other amplifier components can be subjected to a large current surge as the HV filter capacitors discharge. The peak discharge current can exceed 1000a if a series resistor is not used to limit the short circuit current that can be delivered by the HV filter capacitors. The current limiting resistor is placed in series with the positive output lead from the filter capacitors. A wire wound resistor with a high length to diameter ratio works best. A 10 ohm, 10W wire wound resistor is adequate for up to about 3kV & 1A. For higher voltages, additional 10 ohm, 10W resistors can be added in series to share the voltage drop during a glitch. Wire wound resistors with a high length-to-diameter ratio are best for this type of service. Since about 1985, Eimac® has recommended the use of a glitch protection resistor in the anode supply circuit. Svetlana® typically recommends using a 10 to 25 ohm glitch resistor.

A HV current limiting/glitch resistor may disintegrate during a major glitch--so it should be given a wide berth with plenty of chassis clearance. If the chassis clearance is minimal, its a good idea to cover the chassis with electrical insulating tape. Glass-coated (a.k.a. vitreous) wire wound resistors are the most suitable type of resistor for this application. If a glass-coated resistor comes apart during a major glitch, it won't be throwing chunks of shrapnel around--like a less-expensive rectangular ceramic-cased resistor often does. Metal-case power resistors should not be used in this application. If a glass-coated glitch resistor is damaged during a glitch, it should be replaced with two such resistors in series to reduce the peak V-gradient per unit of length during a problem.

If the positive HV arcs to chassis ground--due to lint, a hapless insect, a VHF parasitic oscillation, or moisture--the negative HV circuit will try to spike to several kilovolts negative in the typical 1500W amplifier. In the real world, this type of glitch is not an uncommon occurrence. Anything that gets in the way of the negative spike may be damaged. Since the grid-current meter is normally connected between chassis ground and the negative HV circuit, the meter can be exposed to kilovolts at hundreds of amperes.

The easiest way to protect a current meter is to connect a silicon rectifier diode across it, or across its shunt resistor. Usually, only one diode {cathode band to meter negative} is needed in parallel with a DC meter. In some circuits, it is best to use two diodes in parallel [anode to cathode] with the meter movement to protect against positive and negative surges.

It may take more than one diode to protect a meter shunt resistor. A silicon diode begins to conduct at a forward voltage of about 0.5V. To avoid affecting meter accuracy, the operating voltage per glitch protection diode should not exceed 0.5V. For example, a 1 ohm shunt, at a reading of 1A full-scale, has 1V across it. Thus, two protection diodes in series would be needed to preserve meter accuracy. Similarly, if the shunt resistor for a 1A full-scale meter is 1.5 ohm, the maximum shunt voltage is 1.5V--so three diodes are needed.

Glitch protection diodes should not be petite. Big, ugly diodes with a peak current rating of 200a or more are best. Smaller diodes--and the meter they were supposed to be protecting--can be destroyed during a glitch. Suitable glitch protection diodes are 1N5400 (50PIV) to 1N5408 (1000PIV). In this application, PIV is not important. The 1N5400 family of diodes is rated at 200a for 8.3mS.

During an extremely high current surge, a glitch protection diode may short out--and by so doing protect the precious parts. Replacing a shorted protection diode instead of a kaput meter is almost fun.

To reduce the chance of the negative HV circuit spiking to several kilovolts, connect a string of glitch protection diodes from the negative terminal on the HV filter capacitor to chassis. At 200a, each diode will limit the surge voltage across it to about 1.5v. Typically, three diodes are needed--thusly limiting the negative spike to about 4.5 volts. Diode polarity is: cathode band toward the negative HV. With one simple wiring change, the same string of diodes can also protect the grid I meter and the anode I meter. This dual protection technique is incorporated into the Adjustable Electronic Cathode Bias Switch on Figure 7.

Design Considerations for Indirectly-Heated Cathode Tubes

A HV arc can destroy an indirectly-heated cathode tube. Here's how it happens: In some amplifiers, one side of the filament/heater is grounded. The cathode is connected to the negative HV circuit. If the negative HV spikes to several kilovolts, the cathode will often arc to the grounded heater. At a minimum, this breaks down the insulation between the heater and the cathode. Sometimes the heater wire burns out--and sometimes the cathode arcs to the grounded grid. Either way the tube is kaput.

Grounding one side of the heater is an invitation for cathode-to-filament breakdown. Instead, let both heater wires float. If the heater is fed through an c.40micro H bifilar RFC, one side of the heater can be wired to the cathode. Even though this arrangement can not protect against cathode-to-grid breakdown, it assures that the voltage between the filament and the cathode is unlikely to rise to dangerous levels.

Safety Devices

HV-Shorting

Manufactured amplifiers typically use a safety device to automatically short the +HV supply to chassis ground when the output section cover is removed. If the cover is removed before the HV filter capacitors have discharged, the resulting positive HV to ground short can damage the amplifier. In most g-g amplifiers, the only DC current path between the negative HV circuit and chassis ground is the grid-current meter and its shunt resistor. Even if the remaining charge in the HV filter capacitors is only 200V when the short from positive to ground occurs, without glitch protection diodes, the entire 200V appears across the grid-current meter shunt and the grid-current meter. Many potentially-fatal amperes can flow into the grid meter as the HV filter capacitors finish discharging. If the amplifier is accidentally switched on with the cover removed, rectifier diodes are a common casualty.

Automatic-shorting safety devices are not only dangerous to amplifier components, they can be dangerous to operators. It is dangerous to assume that an amplifier is safe to work on because it contains a safety device. Even though an amplifier's HV supply is shorted, if the amplifier is plugged in, its electric-mains circuitry is still alive and potentially fatal. Amplifiers are inherently dangerous. They should not be worked on casually--even if they have so-called safety devices.

The safest quick method of discharging HV filter capacitors is through a paralleled pair of wire wound resistors. The resistors limit the discharge current to a safe amount. In the unlikely event that one resistor opens, the remaining resistor will do the job. For the average 1500W amplifier, a paralleled pair of non-adjustable 1k ohm to 5k ohm, 50W resistors will do the job. Its always a good idea to check the anode supply voltmeter before putting your hands inside an amplifier.

Fuses

Fuses have current and voltage ratings. For a fuse, the real test is opening safely--not operating without opening during normal operation. A fuse's maximum voltage rating is important. In some amplifiers, ordinary 250V 3AG fuses are casually used in circuits where they may be required to interrupt several kilovolts. Examples are the anode or cathode circuit. All's well until a problem occurs. When a 250V fuse attempts to interrupt a potentially-damaging flow of current in such circuits, the frangible link inside the fuse parts as it should. However, due to the available voltage, when the link melts, a metal vapour arc forms in its place. Metal vapour arcs typically have a voltage drop of around 20V--so the unsafe current will continue to flow in the circuit. At some point, the fuse will eventually explode--usually after serious damage has been done to other components.

During a glitch, circuits which normally carry low voltages can spike to several kilovolts. For example, cathode circuits normally see a maximum voltage of 30V to 100V. Thus, it might seem appropriate to use a 250V fuse to protect the cathode from excessive current. However, when a glitch occurs, several kilovolts can appear across anything that attempts to interrupt the flow of cathode current. The safest place to use ordinary 250V fuses is in the primaries of transformers.

Power Supplies

Ripple Filters

There are basically two types of DC filters: inductor-input / capacitor-output, and capacitor. Each type of filter has advantages and trade-offs.

Capacitor filters have good transient response. Since no inductor and resonating capacitor are used, the capacitor filter is simple to build, compact, cost-effective, requires no tuning and it is lightweight. The main disadvantage of a capacitor filter is that the capacitor is charged only during a small fraction of the waveform supplied by the transformer. No charging current flows until the instantaneous output voltage from the rectifiers exceeds the instantaneous voltage on the filter capacitor. This means that the transformer is either not loaded or severely loaded at different times during each cycle.

For example, with a electric-mains frequency of 60Hz, the duration of a half cycle is 8.333mS. Under load, using a capacitor filter, the capacitor charging time per half-cycle is typically only about 1mS out of the 8.333mS. This means that the ratio between output current and peak charging current can be 8 to 1. To combat I^2 R loss in a capacitor filter power supply, the transformer, all circuitry in the primary (including the electric-mains) and the filter capacitor should have low resistance. Capacitor filters are not appropriate for use with older-design transformers that were intended for use with inductor filters. Typically, such transformers have high winding resistance.

Inductor-input / capacitor filters can be of the resonated type or the non-resonated type. A non-resonated inductor tries to maintain a constant DC-current despite changes in the load current. This is the nature of any inductor. It always tries to maintain constant current by temporarily increasing the output voltage when the load current decreases suddenly, or by decreasing the output voltage when the load current increases suddenly.

When a conventional voltmeter is used to monitor the output voltage from a non-resonated inductor / capacitor filter power supply, the transient unregulation characteristic will usually not be detected because of the damped response in the meter movement. If a DC oscilloscope is used to monitor the output voltage while a string of around 5 WPM CW dashes are sent, the instantaneous output voltage swings can be easily observed. On make, the output voltage spikes downward. On break, the output voltage spikes upward. The amplitude and width of the spike depends on how much filter capacitance is used after the inductor and on the change in current. Upward and downward voltage spikes of more than ±50% are possible during a sudden load change on a non-resonated inductor / capacitor filter power supply.

Transient unregulation is probably not an important consideration unless SSB operation is used. With SSB, the PEP output and the linearity of the amplifier would be adversely affected by a non-resonant inductor DC filter.

The resonant DC filter maintains a fairly constant output voltage during rapid or slow changes in current demand--provided that a minimum current passes through the inductor. This minimum current can be the zero-signal anode-current [a.k.a. 'idling current'] of the tube itself. The inductor is resonated with a parallel capacitor. In actual practice, the value of capacitance used--as well as the bleeder resistance--is that which produces satisfactory voltage regulation. The resulting resonant frequency is usually slightly higher than double the frequency of the electric-mains. Resonant L/C pairs are available from Peter W. Dahl, Inc.

DC filter inductors come in two types, fixed-inductance and swinging-inductance. A swinging-inductor changes its inductance according to the current that is passing through it. Obviously, a swinging inductor can not stay tuned correctly with changes in current. Therefore, resonated-inductor filters can only use a fixed inductor.

The disadvantages of a resonant inductor filter are:

The resonating capacitor must have a DC-working voltage rating of about three times the DC output voltage of the supply Typical values are 0.1 to 0.15 micro F @7.5kV to 15kV.

To maintain voltage regulation during standby, a minimum 'bleeder' current must flow in the inductor. Typically, the bleeder current is 10%. Considerable heat is dissipated by the bleeder resistor(s). However, if the filter capacitor can withstand the approximately. 50% increase in voltage during receive, the 10% standby bleeder current requirement can be reduced to 0.5%.

The inductor is heavy and costly.

A resonant inductor filter usually makes an audible noise--unless the inductor has been potted in plastic resin.

The advantages of a resonant-inductor DC filter are:

Excellent voltage regulation.

Greatly reduced peak current demand on the transformer and the electric-mains. Most importantly, this reduces transformer heating.

Transformers have about double the output current capability when a resonant inductor filter is used instead of a capacitor filter. However, the output V from a capacitor filter is higher.

The resonant filter is used extensively by commercial and military amplifier manufacturers. Since a resonant filter demands much less peak power from the electric-mains than a capacitor filter demands, for 120V operation, where available power is typically much more limited than with 240V operation, a resonant filter is clearly the best choice. The resonant filter is also the best choice for high duty-cycle modes such as RTTY, FM or AM.

Rectifier Circuits

Half wave. ----Advantages: may be used where one side of the AC input is grounded. Disadvantages: requires highest filter C; causes DC current to flow in the transformer; poor voltage regulation; transient-voltage protection is needed to protect the rectifier from reverse voltage spikes.

Fullwave-centertap---- The fullwave-centertap rectifier circuit was used in ancient times when tube-type rectifiers were the only game in town. Only one rectifier filament-winding was needed to produce full wave rectification, so the center-tapped secondary winding was the norm in older transformers. If a more efficient fullwave-bridge rectifier circuit had been used, three rectifier filament windings would have been needed. Advantage: if needed, reduces output voltage to one-half or to approximately. one-fourth of the DC voltage that would be obtained with a full wave bridge or full wave voltage-doubler. Disadvantage: inefficiently utilizes only half of the secondary winding at any instant--resulting in less than optimum transformer efficiency.

Full wave bridge---- Advantages: full utilization of the transformer's capability; may be used with a resonant filter. Disadvantage: requires twice as many transformer secondary turns as the full wave voltage-doubler requires. This means more layers of insulating paper--and that takes up winding space--so smaller wire must be used.

Full wave voltage doubler---- Advantages: full utilization of the transformer's capability; to achieve a specific DC voltage, only half as many transformer secondary turns are needed compared to a fullwave-bridge circuit. This means that the power transformer secondary will have a higher ratio of copper to paper. If switched secondary taps are used to control the DC output-voltage, the voltage stress on the switch is only half of what it would be with full wave bridge rectification. Full wave doubler supplies typically have a remarkably low ripple content. This is because one half of the filter capacitance is being charged at the same time the other half is being discharged. Since the charging sawtooth waveform is similar and opposite to the discharging sawtooth waveform, the result is a fairly smooth DC output. Disadvantages: To achieve acceptable voltage regulation, the full wave doubler requires twice as much filter capacitance as a fullwave-bridge. This is the case because each of the two filter capacitor sections is charged once per cycle versus being charged twice per cycle with the fullwave-bridge circuit. Thus, in a full wave doubler, the filter-capacitors must be able to hold their charge twice as long--so twice as much filter capacitance is needed. The capacitance requirement is easily met with modern aluminum electrolytic capacitors. They provide a large amount of capacitance in a small space at a reasonable cost. Full wave voltage doublers are not practical for use with a resonant inductor filter

Transformers

Transformers are available in two basic types: E-I (conventional) core and toroidal core. The E-I core is made from a stack of thin E-shaped and I-shaped iron plates. When placed together they form a rectangle with two windows for the windings. A stack of E-I rectangles make the completed core. The toroidal core is made from a continuous tape of grain-oriented material that contains iron and silicon plus other elements that increase the permeability of the core and decrease loss. This core material is known as Hipersil. Westinghouse Corp. was the original patent and copyright holder. Their patent expired decades ago. There are different grades of Hipersil tape. Grade 5 has the highest performance. Grade 22 has the lowest performance.

Higher permeability means that fewer turns are needed to achieve the required inductance in each winding. This means that larger diameter wire can be used. The end result is a transformer with low resistance and high efficiency. The Hipersil core is so efficient that the principal loss factor is the resistive loss in the copper wire. Hipersil core transformers are capable of producing extremely high peak currents. Thus, the Hipersil core transformer is ideally suited for capacitor filter power supplies.

It is difficult and time-consuming to thread a continuous tape core through the completed transformer windings--so someone came up with a faster way of uniting the core with the windings. Here's how it's done: The tape is wound on a form of the appropriate dimensions. The tape is spot welded together, removed from the form, and annealed at about 700 degrees C to relieve internal stresses. After cooling, the core is varnished and dried. Then the core is cut in half with a machine that makes a precise square cut. The faces of the cut are then polished flat. Thus, the halves of the core can slip into the completed windings, contact each other closely--restoring nearly perfect magnetic coupling between the halves of the core. The matched halves of the core are marked so that they can not be inadvertently mixed up with other core halves. The reunited halves of the core are held together tightly by steel bands like those used for binding heavy cartons and crates. If future access to the primary and secondary windings is needed, a Hipersil transformer can be disassembled by cutting the steel bands and removing the core halves.

Hipersil is no longer the most efficient type of core material. The new amorphous core transformer is starting to come into use by electric utilities. An amorphous core transformer is so efficient that if the secondary is unloaded and the primary is disconnected from the electric-mains, the collapsing magnetic field generates a voltage spike that can destroy the transformer. To avoid this problem, one winding is paralleled with a suitable voltage surge absorber.

Transformer Power Ratings

Transformers are commonly rated in maximum "volt-amperes" [VA]. Maximum VA are roughly equal to maximum RMS watts when the rated RMS current is flowing in each transformer winding and the transformer is operated from the rated input voltage at the design frequency. If the electric-mains voltage is reduced, the VA capability of the transformer decreases.

For SSB and CW operation, a lighter transformer may do the job just as well as a much heavier and more costly transformer. Manufactured 1500W amplifiers typically use a HV transformer with a continuous capability of roughly 600W--or VA. Such transformers are completely satisfactory for normal SSB operation. Such transformers are also capable of handling brief FM and RTTY transmissions--provided that the lower voltage tap is used.

If a power supply's DC output voltage drops more than about 10% under modulation, it's a fairly safe assumption that a more capable transformer is needed. Of course, not using enough filter capacitance or excessive electric-mains resistance can also cause poor regulation.

Transformer Current Ratings

Increasing the current in any conductor causes a square-law increase in the amount of power dissipated in the conductor. Since P=I^2 x R, doubling the current causes a 4 times increase in dissipation. This is an especially important consideration with transformers because they have considerable difficulty dissipating the heat that is generated deep inside their windings. This problem is compounded because copper has a positive, resistance versus temperature, coefficient. Thus, as the copper heats up, its resistance increases--which increases the dissipation--which increases the resistance, et cetera. This can lead to thermal runaway and transformer failure.

If a transformer has a secondary rating of 1A RMS, it means 1A with a resistive load. If connected to a rectifier and DC filter, the 1A rating does not necessarily apply. For example, if a fullwave-bridge rectifier, resonant filter circuit is used, the RMS current rating can be multiplied by at least 1.2. The DC output voltage will be about 0.85 times the RMS voltage. If a fullwave-bridge rectifier, capacitor-filter circuit is used, the loaded DC output voltage will be about 1.3 times the RMS voltage. A 30% increase in voltage sounds good, but obviously you don't get something for nothing. The trade-off for the increase in voltage is a decrease in current capability. The high peak current demanded by the capacitor filter translates into a substantive current capability decrease. .

Any formula for converting a transformer's RMS current rating to a DC output current rating is bound to be problematic due to the large number of variables. Here's a rule-of-thumb that is fairly accurate. If, after about an hour of typical operation, the outside of the transformer is uncomfortably hot for one's thumb, the internal parts of the transformer are probably deteriorating. Reducing the average load current slightly will greatly reduce transformer heating because of the square-law relationship between current and power dissipation. For example, reducing the current by 30% will reduce winding dissipation by about 50%.

There is a simple, reasonably accurate, 2-step approximation for determining the safe SSB, maximum current rating for a specific transformer for use with a capacitor filter and a full wave bridge rectifier. A slightly different approximation is used for a full wave voltage-doubler. These approximations are based on the DC resistance and the AC-voltage of the transformer's secondary winding. These approximations are useful when shopping around surplus stores or swap meets. All that's needed is an ohm-meter and a clip-lead. The clip-lead is used to short the primary of the transformer. This dampens the inductive voltage spike that occurs when the ohm-meter is disconnected.

The fullwave-bridge, capacitor filter approximations are: Multiply the secondary winding resistance by 70 to find the minimum intermittent load resistance that can be placed on the power supply. To find the DC output voltage under load, multiply the secondary RMS-voltage by 1.3. To find the safe intermittent current rating for SSB service, use Ohm's law and divide the output voltage by the minimum load resistance. For a more accurate evaluation, use the appropriate graphs in this book.

For example, a 2000V RMS secondary winding has a DC-resistance of 60 ohms. A full wave bridge rectifier, capacitor filter, circuit will be used. The safe, minimum, intermittent load resistance is approximately 70 x 60 ohm = 4200 ohms. The approximate voltage delivered under load would be 1.3 x 2000V = 2600V DC. Thus, the maximum intermittent load current is 2600V ÷ 4200 ohms = 0.62a.

Another approximation can be used to find the amount of filter capacitance needed. The approximation is 50,000 divided by the minimum load resistance. In the above example this is 50,000 ÷ 4200 = 12micro F.

For a full wave voltage-doubler, capacitor filter, power supply, the SSB-service approximations are: Minimum intermittent DC-load resistance equals 300 times the winding resistance; DC-output voltage, under load, equals 2.5 times the secondary RMS voltage.

For example: A 1000VRMS transformer has a winding resistance of 10 ohms, the minimum load resistance for full wave doubler operation would be 300 x 10 ohms=3000 ohms and the output voltage would be 2.5 x 1000V=2500V. The maximum intermittent load current is 2500V ÷ 3000 ohms = 0.83A.

The amount of filter capacitance needed for each half of the full wave voltage-doubler circuit is approximately 200,000 divided by the minimum load resistance. In the above example each of the two capacitors should have a minimum of 200,000 ÷ 3000 = 67micro F.

There is more to transformer performance than secondary resistance. If a Hipersil® core is used, core loss is minimal and the maximum intermittent power capability increases. Primary resistance is another factor to consider since it is effectively in series with the electric-mains resistance. Electric-mains resistance can cause a voltage drop problem if the amplifier is a fair distance from the service entrance box and you are using a capacitor filter power supply. One solution is to use larger diameter wires than the electric code requires. Another solution is to install the power supply near the service box and bring the HV DC to the amplifier.

END OF PART 2

AG6K Part 1

2026-08-20T04:28:55.615-07:00

Preface

The following manuscript was originally written under contract for the Amplifier and Projects Chapter (13) of the 1995 ARRL Handbook. About a month before the due date on the contract, I received a phone call from the ARRL Handbook Editor. He told me to stop writing.

Introduction

Amplification can be defined as: The process of increasing the magnitude of a variable quantity--especially the magnitude of a voltage and/or current, without substantially altering any other quality.

The (pure) sine wave is the only periodic waveform that contains no harmonic energy. All of the energy in a sine wave is contained in the fundamental frequency. In other words, a pure sine wave is coherent. Maintaining the quality of sine waves during amplification is a major concern in the design and operation of amplifiers.

Even though the following discussion pertains to the operation of gridded electron tubes, some of the concepts apply to power FETs because both are voltage-driven amplifying devices.

In the discourse that follows: if a voltage is said to be negative-going, that does not necessarily mean that the voltage is negative. It could be a positive voltage that is moving in the negative direction. If a voltage is positive-going, or moving in the positive direction, it could be a negative voltage or a positive voltage that is moving in the positive direction.

In an electron tube, the cathode emits a cloud of electrons. Since electrons carry a negative charge, and unlike charges attract, electrons are strongly attracted by the positive voltage that is applied to the anode. Unless something is placed in the way, much current flows between the cathode and the anode.

The term "grid" describes appearance--not function. A grid is made from a number of closely spaced wires or bars--like a bird cage. The grid is placed close to the cathode. As a result, the grid has more influence over the cathode's electrons than does the more distant anode. Thus, a small change in grid-voltage produces a large change in the flow of electrons. Because the voltage applied to the grid controls the flow of electrons from the cathode to the anode, in function, the grid acts like a valve or gate. The grid requires virtually zero current to perform its job. As a result, the power gain of a gridded electron tube is theoretically high.

Since like charges repel, a sufficiently negative grid can stop electrons from traveling to the anode. As the grid is made less-negative, the flow of electrons to the anode steadily increases. In other words, a positive-going, albeit negative polarity, grid-voltage causes an increasing flow of electrons between the cathode and the anode. The reverse is also true: a negative-going grid-voltage causes a decreasing flow of electrons from the cathode to the anode. As long as the grid remains negative with respect to the cathode, the relationship between grid-to-cathode voltage and anode-current is fairly linear.

When an appropriate load resistor is connected between the anode and its positive voltage source, the changes in anode-current produced by changes in grid-voltage create a proportional, typically much larger, voltage change across the resistor. The ratio of changes in voltage across the load resistor to changes in grid-voltage is the voltage amplification factor. It is designated by the Greek letter Mu. Since Mu is higher at high anode-voltages than it is at low anode-voltages, average Mu is a more meaningful number than maximum Mu. Average Mu ratings vary from about 2 to 240. Mu is partly determined by the spacing between the grid wires and the distance between the grid and the cathode..

Classes of Operation

The duration of anode-current conduction per cycle determines the class of amplifier operation. A conduction angle of 360 degrees means that the anode is conducting current during 100% of the input sine wave cycle. A conduction angle of 90 degrees means that the anode is conducting current during 25% of the input sine wave cycle. Long conduction angles produce a more linear representation of the input sine wave. Short conduction angles produce more efficiency--and less linearity.

Class A is defined as a conduction angle of 360 degrees. Class B is defined as a conduction angle of 180 degrees. Class C is defined as a conduction angle of less than 180 degrees. The subscript 1 indicates that no grid-current flows. The subscript 2 indicates that grid-current flows--the result of driving the grid into the positive voltage region.

When the conduction angle is less than 360 degrees, the missing part of the sine wave must somehow be filled in. One way of filling in the missing part of the sine wave is by utilizing the flywheel effect of an output tank circuit. Another way to produce a smooth sine wave is to use a push-pull configuration. If each device in a push-pull circuit conducts for at least 180 degrees, a smooth sine wave can be produced.

Class A Operation

Class A is the most linear class of amplifier operation. Class A amplifiers produce only about 1/100,000 part, or minus 50dB, distortion. The theoretical efficiency of a Class A amplifier is 50%. The practical efficiency is slightly lower. Class A is used mainly in low level amplifiers--where efficiency is not much of an issue. Since Class A operates with continuous (360 degrees) conduction, no tank circuit is needed to complete the sine wave. Class A is ideal for wide band amplification.

The zero-signal anode-current [ZSAC] in Class A is set to roughly half of the electron tube's maximum anode-current rating. Although the meter-indicated anode-current remains constant from zero signal to maximum signal, the instantaneous anode-current typically varies from just above zero to c. 6-times the meter-indicated anode-current.

The maximum available power in Class A is roughly equal to the anode-dissipation rating of the electron tube.

The Class A amplifier can be compared with a gas turbine engine. Both have a smooth, continuous power stroke--and neither one is very efficient.

Class AB1 Operation

Class AB1 amplifiers are roughly 60% efficient. The trade-off for increased efficiency is slightly more distortion--roughly 1/10,000 part, or minus 40db. Since most transceivers produce more than minus 36db of IMD, such an amplifier would not add significant distortion.

The anode-current in Class AB1 varies in proportion to the grid-voltage for about 60% of the input sine wave cycle. Thus, the anode-current is off for about 40% of each input sine wave cycle. The missing 40% in the output sine wave is filled in by the flywheel effect of the output tank circuit.

In Class A or Class AB1 a somewhat unusual relationship exists between the work being performed by the tube and the grid voltage. The grid is operated in the zero to negative voltage region. Maximum instantaneous anode-current, maximum anode voltage swing and maximum peak power output coincide with an instantaneous grid-voltage of zero--i.e., maximum stoke equals zero grid volts.

The grid must not be allowed to become positive. If the grid became positive, electrons from the cathode would begin flowing into the grid. Whenever grid-current flows, the linear relationship between grid-voltage and anode-current deteriorates.

Since there is zero grid-current in Class A or Class AB1 operation--and any voltage multiplied by zero amperes is zero watts--the driving power is usually stated as zero on the tube manufacturer's technical information sheet. However, because charging and discharging a capacitor requires current flow, in the real world of conductor RF resistance, charging/discharging the grid capacitance at an RF rate consumes some power. In a typical HF amplifier, the drive power required for Class AB1 MF/HF operation is roughly 1% to 2% of the output power. Thus, the typical power gain is roughly 50 to 100. As frequency increases, conductor resistance increases due to skin-effect. More R causes more I^2 R loss. As frequency increases, the amount of current needed to charge and discharge the grid capacitance also increases--causing even more I^2 R loss. These losses can only be compensated for by adding more drive power.

Drivers (usually a transceiver) require a resistive load--which the capacitive grid does not provide--so a suitable resistance must be connected from the grid to RF-ground..

The zero-signal anode-current [ZSAC] in a Class AB1 amplifier is normally set to about 20% of the maximum-signal single-tone anode-current.

Tubes that are designed for Class A and Class AB1 service produce high peak anode-current when the instantaneous grid-voltage is zero. Typically, the peak anode-current is about three times the maximum rated (average) anode-current. Most of the tubes that are used in Class A and Class AB1 RF amplifier service are tetrodes and pentodes--devices that have the advantage of grid-to-screen amplification. Triodes are seldom used because the only grid designs that can produce high anode-current with zero grid volts are those that have a Mu of 2 to 5. Low Mu triodes require much more driving voltage than a comparable tube with a screen requires. Since imposing a high RF voltage across the capacitive grid is difficult, low Mu triode Class A and Class AB1 power amplifiers are only practical up to a few hundred kHz.

Class AB1 Cathode-Driven Operation

The most common configuration for Class AB1 operation is grid-driven. Since grid amplification as well as grid-to-screen amplification takes place, the resulting power gain is high. Class AB1 amplifiers can also be cathode-driven if the tube is a tetrode or pentode. The grid is tied to the cathode. Thus, the grid-voltage is always 0V--so no grid-current flows. The screen is grounded. The input signal is applied to the cathode/grid. Because input signal voltage is applied between the grid and the screen, grid-to-screen amplification takes place. However, since the grid is tied to the cathode, no grid amplification takes place. Although the power gain is relatively low, linearity is excellent. The Collins 30S-1 is an example of a cathode-driven Class AB1 amplifier.

The Class AB1 amplifier is like a 2-cycle, single-cylinder engine. The power stroke is roughly half of each crankshaft revolution--it has a flywheel (the tank circuit) that supplies power between power strokes--and it is more efficient than a gas turbine (Class A).

Class AB2 Operation

Class AB2 is similar to Class AB1 except that the grid is driven into the positive voltage region during a part of the anode conduction period. A Class AB2 amplifier can be grid-driven or cathode-driven.

Grid-Driven Class AB2 and Negative-feedback

When the grid is driven positive, it attracts and accelerates the cathode's electrons. Some of the electrons stick to the grid, resulting in grid-current. Electrons that miss the grid travel to the anode. The accelerated head start causes a sharp increase in instantaneous anode-current--and a sharp decrease in linearity. The distortion products from a single-ended, Class AB2 grid-driven amplifier are roughly 1/100 part [minus 20db]. In SSB service, this level of distortion is virtually certain to cause interference to other stations using adjacent frequencies. However, by limiting grid-current and by adding an unbypassed, low-L cathode feedback resistor (to develop an out-of-phase {negative} feedback voltage) it is possible to achieve acceptable linearity in grid-driven Class AB2 operation--but only if the grid current produced is small.

An unbypassed cathode resistor is also useful for improving linearity in Class AB1. For example, the 4CX250B has a somewhat objectionable distortion level in Class AB1, SSB service. Adding a 25 Ohm resistor between the unbypassed cathode and chassis ground improves linearity. The trade-off is that slightly more grid drive voltage is needed to achieve the same output level. Cathode negative-feedback is also useful with TV sweep tubes--devices that were originally designed for switching--the opposite of linear amplification. When an appropriate cathode resistor is used with a sweep tube, reasonable linearity can be achieved.

Class AB2 Grounded-Grid Operation

Even though Class AB2 cathode-driven/grounded-grid operation produces grid-current, it is never the less fairly linear due to the laundering effect of negative-feedback. This is the result of the input and output signals being in series with each other and out of phase. Due to the negative-feedback, the distortion level in Class AB2 grounded-grid service is low--typically about 40db below PEP.

High-Mu triodes work well in Class AB2 grounded-grid operation. Medium-Mu triodes can be used, but they have less power gain. Tetrodes and pentodes usually work well in grounded-grid operation. Since tetrodes and pentodes typically have a grid-to-screen amplification factor of about 5, its easy to assume that they offer an advantage over triodes in Class AB2 grounded-grid operation. However, RF-grounding the grid and the screen stops grid-to-screen amplification. Applying DC screen-voltage does NOT increase gain because grid-to-screen amplification can not take place unless input signal voltage is applied between the grid and the screen.

The maximum available power in Class AB2 is roughly double the anode-dissipation rating.

Class B Operation

Class B is defined as a conduction angle of 180 degrees. Class B RF amplifiers produce unacceptable distortion in SSB operation.

Class C Operation

Class C is defined as an anode conduction angle of less than 180 degrees. In Class C, the amplifying device is deliberately not operated linearly. Instead, it is operated as a switch in order to reduce resistance loss. The anode conduction angle in Class C operation is usually made as short as is possible. In effect, the tank circuit makes the RF output sine wave--like a bell that is struck at a constant rate by a hammer. This is similar to the principal behind the spark transmitter.

The efficiency of a typical Class C amplifier is high. When compared to a Class AB1 or Class AB2 amplifier operating at the same power input, a Class C amplifier will deliver a received signal increase of about 1db--in other words, 1/6 of 1 S-unit. However, significant trade-offs are required to achieve that 1/6 of 1 S-unit. As is the case with Class B operation, the distortion from Class C operation is so high that SSB operation is precluded. Only CW, FM or FSK operation is practical. The harmonic output level from a Class C amplifier is substantial. Extra filtering is usually needed to control harmonic radiation.

The maximum available power in Class C operation is roughly three to four times the anode-dissipation rating of the electron tube.

Class D Operation

Class D is used up to about 1.6MHz--mostly in AM broadcast service. In Class D operation, the amplifying device rapidly switches on and off at a fixed rate--like a switching power supply--except that the output voltage varies at an RF rate. The amplitude of the RF is controlled by varying the on period of the switch. Smoothing is accomplished by a complicated filter that converts some of the odd-harmonic energy from the rectangular waves back to fundamental frequency energy. Class D is highly efficient--but it is limited in frequency capability and frequency agility.

Amplifier Design Considerations

Arriving at an amplifier design that will give years of surprise-free service involves many considerations. Merely copying circuits from published amplifier designs or commercial amplifiers is not necessarily the best approach. Doing so may result in copying someone's mistakes. The best approach is to learn what you can about each section of an amplifier, discuss it with others--then reach your own conclusions.

Basic prerequisites for getting a handle on what's going on inside an amplifier are an understanding of Ohm's Law, inductive and capacitive reactance, impedance, resonance, how gridded electron tubes function, and some knowledge of L and pi networks.

A useful book on amplifier design is Eimac®'s Care and Feeding of Power Grid Tubes. I will minimize the discussion of topics that are covered adequately in this book.

Tubes vs. FETs

When the first RF power FETs were introduced, it was commonly thought that FETs would eventually replace bipolar transistors and gridded electron tubes in HF power amplifiers. Since RF power FETs work better at 50V than they do at 12V, FETs have not replaced bipolar transistors in 12V mobile applications. Another difficulty with FETs is cost. A pair of FETs that can produce 1200W PEP at 29MHz cost about six times more than an electron tube, or tubes, that can do the same job. The FETs' input power requirements are 50V at 50A, i.e., 2500W--so there's considerable heat to dispose of. Meeting the cooling requirement is not nearly as easy as it is with tubes because tubes operate quite happily at surface temperatures that destroy silicon devices.

In low power applications at room-temperature, solid-state devices can last 100 years. However, at the junction temperatures encountered in high power applications, the P and N doped layers slowly diffuse into each other--thereby steadily eroding the device's amplifying ability. A relatively-large rigorous cooling system is needed to achieve a reasonable operating life from high power solid-state RF amplifying devices.

Another difficulty with solid-state high power RF amplifiers is their power supply requirements. Tubes are quite tolerant of moderate variations in their anode supply voltage. Transistors, however, are fatally sensitive to over-voltage. It is much easier to build a 3000V, 0.8A unregulated supply for a tube than to build a regulated 50V, 50A power supply with over-voltage, over-current and over-temperature protection circuitry for high-power solid-state devices.

The bottom-line is that 1500W, HF, gridded electron tube amplifiers are more efficient, more forgiving, easier to cool, more compact, weigh less, are more tolerant of high SWR and are less costly than 1500W HF semiconductor amplifiers. For instance, a pair of legal-limit FETs cost about $800 from Motorola®. The efficiency is about 10% less that what one can achieve with gridded electron tubes.

Grounded-Grid Versus Grid-Driven

For at least the last three decades, the vast majority of amateur radio amplifier designs have been Class AB2 cathode-driven--a.k.a. 'grounded-grid'. One reason for this is simplicity--or at least the appearance of simplicity. Ground the grid(s), drive the cathode. Only three supplies are needed--the T-R switching supply, the filament supply and the anode supply. Neutralization is theoretically not needed because the grounded grid(s) shield the output element, the anode, from the input element, the cathode. This theory works almost perfectly. Grounded-grid amplifiers are virtually always stable at the operating frequency because the reactance of the feedback C is too high at HF to allow regeneration. This is fortunate because there is no way of neutralizing a single ended grounded-grid amplifier. Another advantage is flexibility. Almost any tetrode, pentode, or high-Mu triode from the junk box will work. Linearity is usually good and the typical power gain--10db to 14db--is acceptable. So far, so good. Now for the trade-offs.

What goes on inside a grounded-grid amplifier is not as simple as it looks. The AC component of the anode-current and the grid-current, i.e., the RF cathode current, passes entirely through the cathode coupling capacitor and the tuned-input circuit--so the input circuit is in series with (and out of phase with) the output circuit. The components in the tuned circuit must be able to handle a substantial amount of RF current. Manufacturers of tubes that are designed for grounded-grid operation typically recommend using a tuned input pi-network with a Q of 2 to 5. To maintain an acceptable SWR and Q when the operating frequency changes appreciably, all three reactances in the tuned input must change proportionally. However, if Q is allowed to change, L can be left as is providing that C1 and C2 are retuned. [For more information on this problem, see the section titled "Tuned Input Circuits."]

Even though HF grounded-grid amplifiers are stable at their operating frequency, at VHF the grid looses its ability to shield the input from the output. HF grounded-grid amplifiers have a less-than-pristine reputation for VHF stability.

For wide frequency coverage, the Class AB1 grid-driven amplifier requires a much simpler tuned input than a grounded-grid amplifier requires. Typically, grid-driven amplifiers have more power gain than grounded-grid amplifiers. One Class AB1 grid-driven amplifier has about as much gain as two Class AB2 grounded-grid amplifiers in series. The trade-off is two additional DC supplies--a grid bias supply and a screen supply. Both of these supplies need to be adjustable. HV power FETs make this task easy.

Cathodes

There are two types of cathodes--directly-heated and indirectly-heated. In a directly-heated cathode, a ditungsten carbide layer on the hot (c.1800 degrees K) tungsten, alloyed with about 1.5% thorium--a.k.a. 'thoriated-tungsten', filament wire emits electrons. In an indirectly-heated cathode, the filament (a.k.a. heater) heats a metal cylinder that is coated with strontium oxide and barium oxide. This coating is relatively frangible--but highly emissive.

Ditungsten carbide is commonly formed by heating tungsten in an atmosphere of acetylene (C2H2) gas. Carbon atoms in the gas break their electron bonds with hydrogen atoms and bond with tungsten atoms to form ditungsten carbide on the surface of the filament wire. Since it is atomically linked to the underlying tungsten, the ditungsten carbide layer is very durable. During use, the process reverses. Ditungsten carbide gradually looses carbon and changes back to tungsten. Extra heat exponentially accelerates this process. A cathode is worn out when the carbon is mostly used up.

After their cathodes grow tired of emitting electrons, large external-anode amplifier tubes are commonly "recarburized" with acetylene, vacuum-pumped and resealed. This restores full emission. Although it is possible to recarburize a 3-500Z, doing so is not economically feasible. The smallest tube that is currently being recarburized is the 3CX1000A7.

Each type of cathode has advantages and disadvantages. Indirectly-heated cylinder [8877] and planar [3CX100A5] cathodes have much less inductance than a directly-heated cathode made from wires. Thus, indirectly-heated cathodes are more frequency-capable. Some indirectly-heated cathode tubes can perform satisfactorily at 2500MHz. The 3CX100A5 is an example.

Directly-heated/thoriated-tungsten cathodes are more resistant to damage from electrons that bounce off the anode. It's possible to use up to 22kV with the larger thoriated-tungsten cathode tubes. Electrons that have been accelerated by such voltages move at very high velocities. When they strike the anode, they produce X-rays.

A thoriated-tungsten cathode typically warms up in one second, while few indirectly-heated cathodes can warm up safely in one minute--and three to five minutes is not uncommon. For HF operation, indirectly-heated cathode tubes have a much higher cost to watt ratio than thoriated-tungsten cathode tubes. For VHF and especially for UHF operation, indirectly-heated cathode tubes are often the only choice. For super-power HF operation, thoriated-tungsten cathode tubes are the only choice.

Cathodes deserve respect. Filament-voltage and filament inrush current are the prime areas for concern.

Filament / Heater Considerations

For optimum life from a thoriated-tungsten cathode, the filament-voltage should be just above the voltage where PEP output begins to decrease. As a thoriated-tungsten cathode ages, filament-voltage needs to be increased incrementally to restore full PEP. By using this technique, commercial broadcasters typically achieve an operating life of more than 20,000 hours from thoriated-tungsten cathode tubes.

According to Eimac®'s Care and Feeding of Power Grid Tubes, "every 3% rise in thoriated-tungsten cathode filament-voltage results in a 50% decrease in life due to carbon loss." Each additional 3% rise in filament-voltage decreases the life by half. Thus, cathode life is proportional to [E1/E2]^23.4 where E1 is the lowest filament-voltage at which normal PEP output is realized--and E2 is the increased filament-voltage. However, for heater-type oxide cathodes, if the heater potential is allowed to fall below the specified level, the emissive material may flake off of the cathode, and cause a cathode-grid short. On the other hand, excessive heater potential causes barium migration to the grid - which results in primary grid emission.

Controlling Filament-Voltage

It's simple to make the filament voltage adjustable when the filament is powered by its own transformer. All that's needed is a small rheostat in series with the primary. For dual voltage, dual primary transformers, a dual ganged rheostat is required. However, when the filament is powered by a winding on the HV transformer, making the filament-voltage adjustable is more difficult since a dual ganged, very low resistance, high current rheostat must be connected to the low-voltage high-current secondary winding. The typical value needed for a pair of 3-500Zs would be (2) adjustable 0.01 Ohm @ 30A--most definitely not a common rheostat. A reasonable substitute can be made from a double pole, c.10 position, 30A rotary switch and short lengths of resistance wire or ribbon bridging the fixed contacts.

An indirectly-heated cathode can be ruined by operating it below the rated minimum filament-voltage. When operated above its maximum filament-voltage rating, an indirectly-heated cathode boils off emissive material (principally barium) onto the grid and other parts. This results in decreased cathode life and undesirable grid-emission when the grid warms up during transmit. This condition is indicated when the output power steadily drops off in AØ (max. signal, key-down, a.k.a. NØN) operation. The decrease in power normally begins within two seconds.

For maximum cathode life in HF communications service, an indirectly-heated cathode should be operated at the rated minimum filament-voltage. This can be accomplished best with a regulated DC supply.

Controlling Filament-Voltage During Receive

In a typical amateur radio amplifier, the filament-voltage rises about 5% during receive due to decreased load on the electric-mains. Not only is a 5% increase in filament-voltage during receive useless, it uses up thoriated-tungsten cathode emission life about three times as fast as during transmit. However, if an appropriate resistor is placed in series with the filament transformer primary, and shorted out by a relay during transmit, the filament-voltage will not change appreciably between transmit and receive. An appropriate type of relay for shorting the resistor on transmit is a 'power' reed-relay.

Filament Inrush Current

Thoriated-tungsten filaments commonly consist of two vertical intermeshing helices (coils) of tungsten wire that are suspended by their ends. (see Sept. 1990 QST, p.15) The conductance of tungsten at room temperature is about 8.33 times the conductance at the normal operating temperature. Thus, the start-up current for a 15-ampere filament can exceed 100 amperes. Needless to say, 100 amperes makes for a dandy electromagnet.

In a high amplification triode such as the 3-500Z, the filament helices clear the grid cage by a matter of thousandths of an inch. If the position of the filament changes, a grid-to-filament short may result. Therefore, it is prudent to limit filament inrush current in order to minimize thermal and magnetic stress.

Since the grid-to-cathode clearance in an indirectly-heated cathode is not affected by movement of the heater inside the rigid cathode cylinder, indirectly-heated cathodes are not affected by inrush current.

For many of its smaller thoriated-tungsten cathode amplifier tubes--such as the 3-400Z and 3-500Z--Eimac® recommends that filament inrush current be limited to no more than double the normal current. This rating is easily exceeded unless a special current-limiting filament transformer or a step-start circuit is used.

END PART 1

WHAT ARE THE DIODES D16 AND D18 IN THE SB-200 SCHEMATIC

2026-08-19T17:02:00.000-07:00

Based on the Heathkit SB-200 schematic and parts list, the diodes labeled D16 and D18 are 1N4007 (or originally specified as 1N2071 / 1N1319 in early manuals, which are equivalent general-purpose silicon rectifiers).

Function: These diodes are part of the ALC (Automatic Level Control) and metering protection circuitry.
- D18 specifically monitors plate current for the ALC system, developing a DC voltage proportional to the RF output to feed back to the transceiver.
- They are typically located near the meter switch and the 20nF (C12) and 2nF (often part of the meter calibration or bypass network) capacitors on the main chassis or meter board.
Specifications:
- Type: Silicon Rectifier
- PIV (Peak Inverse Voltage): 1000V (for 1N4007)
- Current: 1A
- Replacement: A modern 1N4007 is the standard direct replacement.

If your unit has been modified (as many SB-200s have), verify the orientation: the cathode band (stripe) usually faces toward the meter or the ALC output line, depending on the specific schematic revision.

572B Tube Operating Temperatures and Data

2026-08-05T02:53:00.000-07:00

572B Tube Operating Temperatures and Data

The 572B tube is a high-mu power triode designed for RF and audio amplification, relying primarily on radiation and forced-air convection for cooling.

Key Thermal and Electrical Data:

Maximum Envelope (Glass) Temperature: 250°C. Exceeding this limit risks softening the glass seal and causing vacuum failure.
Plate Dissipation:
- Continuous (ICAS): 160 Watts per tube (some modern import versions are conservatively rated at 65–125 Watts due to thinner anode material).
- Peak (CCS): Higher short-term dissipation is possible, but modern tubes have reduced thermal mass compared to vintage US-made versions, limiting overload tolerance to seconds rather than minutes.
Cooling Method: Radiation and Forced Air. The tube is not designed for conduction cooling through the base.
Base Type: Ceramic (to withstand high temperatures), but the seal between the ceramic and glass envelope is the thermal weak point.

Cooling Guide for 572B Tubes

Because the 572B dissipates heat primarily through infrared radiation from the anode to the glass envelope, and then from the glass to the air, effective cooling requires managing both airflow and radiant heat absorption.

1. Airflow Requirements

Direction: Air must flow directly across the glass envelope of the tubes. In amplifiers like the Heathkit SB-200, tubes are mounted horizontally, and the fan pushes air from the rear toward the front (or vice versa) directly over the bulbs.
Volume: High volume is critical. At 600W output, a pair of tubes may dissipate ~400W of heat. Inadequate airflow causes heat to soak into the glass, raising the seal temperature beyond 250°C.
Obstructions: Ensure no components block the direct path of air to the tube glass. Dust accumulation on the glass acts as an insulator; keep tubes clean.

2. Radiant Heat Management

Blackening: Some builders paint the interior metal chassis surfaces facing the tubes with flat black high-temperature paint. This increases the absorption of infrared radiation emitted by the anode, helping to transfer heat away from the tube and into the chassis, which is then cooled by the airflow.
Chassis Ventilation: The amplifier chassis must have adequate exhaust vents. Hot air trapped inside the cabinet will recirculate over the tubes, reducing cooling efficiency.

3. Operational Limits & Duty Cycle

Modern Tube Caution: Modern import 572B tubes often have folded or thinner anodes with less thermal mass than original US tubes. They can reach critical hot spots in 15–20 seconds at high dissipation (200W+), whereas vintage tubes could tolerate this for over a minute.
Duty Cycle:
- SSB/CW: Generally safe with stock cooling.
- RTTY/Data: Not recommended for stock 572B amplifiers without significant cooling upgrades or power derating. The continuous carrier causes heat to build up faster than radiation/convection can remove it, leading to seal failure.
Fan Interlock: The amplifier should never be operated without the cooling fan running. Many operators add a relay to disable high voltage if the fan fails.

Elaborate

Elaborated 572B Thermal Data and Cooling Guide

1. Critical Temperature Limits

The 572B tube's lifespan is dictated by the temperature of the glass-to-ceramic seal, not just the anode.

Maximum Seal Temperature: 250°C. This is the absolute limit for the hard glass envelope. Exceeding this causes the glass to soften, leading to air leaks and immediate tube failure.
Anode Temperature: While the graphite anode can withstand higher temperatures (glowing dull red is acceptable briefly), the heat must radiate to the glass and then to the air. If the glass exceeds 250°C, the tube fails regardless of the anode's condition.
Modern Tube Variance: Modern import 572B tubes often use thinner, folded anodes with less thermal mass. They reach critical hot spots in 15–20 seconds at 200W dissipation, compared to 60–100 seconds for vintage US-made tubes with solid U-channel anodes.

2. Airflow Requirements (CFM)

Effective cooling relies on moving enough air to keep the glass envelope below 250°C.

Minimum Airflow: 35 CFM (Cubic Feet per Minute) is generally sufficient for a pair of 572B tubes operating at legal limit SSB/CW power (~500–600W output).
Recommended Airflow: 50–60 CFM provides a safer margin, especially for modern tubes or higher duty cycles.
Fan Selection:
- Standard "muffin" fans (120mm) often come in 35 CFM (quiet) or 110+ CFM (loud).
- For the Heathkit SB-200, a 35–40 CFM fan is adequate and quieter. If using a high-flow (110 CFM) fan, it is common practice to add a series resistor or diode to reduce voltage and noise, dropping the effective flow to ~45–50 CFM.
Air Direction: Air must flow directly across the glass bulbs. In horizontal mounts, ensure the airflow path is unobstructed by chassis walls or components.

3. Radiant Heat Management

Since the 572B cools primarily by radiation from the anode to the glass, and then convection from the glass to the air:

Chassis Painting: Painting the metal chassis surfaces facing the tubes with flat black high-temperature paint significantly improves cooling. The black surface absorbs infrared radiation from the anode more efficiently than shiny metal, transferring heat to the chassis where the airflow removes it.
Glass Cleanliness: Dust and grime on the glass act as thermal insulators. Keep the tubes clean to ensure maximum heat transfer to the airflow.
Ventilation: Exhaust vents must be large and unobstructed. Recirculating hot air inside the cabinet drastically reduces cooling efficiency.

4. Operational Guidelines & Duty Cycle

SSB/Voice: Safe with stock or modestly upgraded cooling (35+ CFM).
CW: Safe, but monitor tube color. A faint dull red glow on the anode is acceptable; bright orange/red indicates overheating.
RTTY/Data/FT8: High Risk. These modes create continuous carrier conditions that generate heat faster than radiation/convection can remove it in a 572B.
- Derating: Reduce power to 200–250W output (approx. 80–100W dissipation per tube) for data modes.
- Cooling Upgrade: For sustained data operation, increase airflow to 60+ CFM and ensure the chassis is painted black.
Fan Interlock: Always operate with a fan interlock relay. Running high voltage without active airflow can destroy modern 572B tubes in under a minute.

SB-200 AMPLIFIER ALL PARTS BREAKDOWN

2026-07-22T17:44:00.000-07:00

1. High Voltage Power Supply (Main Filter & Metering)

Location: Mounted on the original phenolic PCB or the replacement Harbach PM-200 board, located in the rear left corner of the chassis (viewed from front) near the power transformer. These capacitors are arranged in a series stack to handle ~2000V DC.

Ref Des	Physical Location	Original Value	Modern Equivalent Part Number	Est. Cost (Each)	Notes
C1 – C6	HV Filter Bank (6 caps in series on PCB)	125 µF, 450V Electrolytic (Axial)	Nichicon UHE2W121MHD (120µF, 450V, Radial w/adapter) or Cornell Dubilier 380LX121M450H022 (Axial). Upgrade: Nichicon UHE2W221MHD (220µF, 450V).	$8.50 – $12.00	Critical: Must replace all 6. Total capacitance is ~21µF (stock) or ~36µF (220µF mods). Polarity is critical (+ to - in series chain).
C7	HV Metering Filter (Across meter switch)	0.05 µF, 600V Paper/Plastic	WIMA FKP2D024701F00KSSD (0.047 µF, 630V Film, Axial)	$1.50	Standard value is 0.047 µF. Located near the meter switch wafer.

2. Anode (Plate) Tank & RF Output Circuit

Location: Mounted on the band switch wafer, tube socket pins, and the output tank coil assembly (center-right of chassis). These components handle high RF voltage and DC anode voltage (~2000V).

Ref Des	Physical Location	Original Value	Modern Equivalent Part Number	Est. Cost (Each)	Notes
C8	Anode Choke Bypass (Bottom of choke to chassis)	1 nF (0.001 µF), 3kV Ceramic Disc	Vishay 561R30TCC102 (1nF, 3kV) or CDR31BX102BKSR. Mod: Vishay 561R30TCC562 (5.6nF, 3kV).	$3.50 – $6.00	Key Mod: Replace with 5.6nF for 80m improvement. Located under chassis near anode choke.
C9	HV Blocking/Feedthrough (Plate to Tank)	500 pF, 5kV Doorknob	CDR32BX501BKSR or RF Parts C500-5 (500pF, 5kV Ceramic Doorknob).	$8.00 – $12.00	Large ceramic "doorknob" style. Connects plate compartment to tank coil.
C10	80m Plate Padder (Switched Fixed)	100 pF, 5kV Mica	Cornell Dubilier DM19FA101FO3 (100pF, 5kV Mica) or CDR31BX101BKSR (Must verify 5kV rating).	$6.00 – $9.00	Switched in only on 80m. Often fails; inspect closely.
C25	Plate Tuning Capacitor (Variable)	150 pF, 5kV Variable Air	Johnson 154-150-5 (150pF, 5kV Variable).	$45.00 – $65.00	Large variable capacitor on front panel shaft. Rarely needs replacement unless plates are warped.
C26	Plate Loading Capacitor (Variable Dual)	457 pF (Dual Section), 3kV Variable	Johnson 154-250-3 (250pF/section, 3kV) wired in series/parallel or Hammarlund HF-250-X.	$55.00 – $75.00	Dual-section variable. If one section shorts, the whole unit often needs replacement.
C27	80m Load Padder (Switched Fixed)	500 pF, 3kV Mica	Cornell Dubilier DM19FA501FO3 (500pF, 3kV Mica).	$6.00 – $9.00	Switched in on 80m Load circuit.

3. Input Tuned Circuit (Grid/Cathode)

Location: Mounted directly on the input band switch wafer (front left under chassis) and near the tube grid pins. Critical for matching input SWR.

Ref Des	Physical Location	Original Value	Modern Equivalent Part Number	Est. Cost (Each)	Notes
C29	ALC Sample Coupling	5 pF, 1kV Mica	CDR30BX5R0BKS (5pF, 1kV Mica).	$1.20	Small mica disc near ALC rectifier diode.
C30	80m Input Series	470 pF, 1kV Mica	CDR31BX471BKSR (470pF, 1kV). Mod: 910 pF for better SWR.	$1.50	On 80m switch wafer.
C31	80m Input Shunt (Filament)	(Part of L-network, often stray/structural)	N/A	N/A	Sometimes omitted or structural in L-net config.
C32	40m Input Series	310 pF, 1kV Mica	CDR31BX301BKSR (300pF, 1kV) or 330pF. Mod: 240 pF (with new choke).	$1.50	On 40m switch wafer.
C33	40m Input Shunt	510 pF, 1kV Mica	CDR31BX511BKSR (510pF, 1kV). Mod: 200 pF (swapped with C34).	$1.50	On 40m switch wafer.
C34	20m Input Series	200 pF, 1kV Mica	CDR31BX201BKSR (200pF, 1kV). Mod: 100 pF to reduce coil heat.	$1.50	On 20m switch wafer.
C35	20m Input Shunt	360 pF, 1kV Mica	CDR31BX361BKSR (360pF, 1kV). Mod: 470 pF (swapped with C34).	$1.50	On 20m switch wafer.
C36	15m Input Series	75 pF, 1kV Mica	CDR31BX750BKSR (75pF, 1kV). Mod: Replace with Johnson 142-0101-801 (Trimmer 2-10pF + fixed).	$1.50 (Cap) / $4.00 (Trimmer)	On 15m switch wafer. Often replaced with trimmer for tuning.
C37	10m Input Series	66 pF, 1kV Mica	CDR31BX680BKSR (68pF, 1kV). Mod: Replace with Trimmer.	$1.50	On 10m switch wafer (if 10m mod installed).
C38	Grid Bypass (Per Tube, Pin 1 to Ground)	200 pF, 1kV Ceramic	CDR31BX201BKSR. Critical Mod: CDR31BX102BKSR (1nF, 1kV) or 1.5nF.	$1.50	Two required. Directly on tube socket pins. Increase to 1-1.5nF for stability.
C39	10m Input Shunt	(Varies, often ~100 pF)	CDR31BX101BKSR (100pF). Mod: Trimmer.	$1.50	On 10m switch wafer.

4. Bias, Filament, Relay, and Low Voltage Circuits

Location: Scattered across the chassis: bias rectifier board (rear), filament choke (near tubes), relay assembly (front right), and meter lamp circuit.

Ref Des	Physical Location	Original Value	Modern Equivalent Part Number	Est. Cost (Each)	Notes
C12	Filament Choke Bypass	0.02 µF (20 nF), 50V+ Ceramic	Vishay K5050203Z (0.022 µF, 50V Disc).	$0.80	Mod: Relocate from choke to directly across tube grid pins for better RF ground.
C13	Bias Rectifier Filter	20 µF, 150V Electrolytic (Axial)	Nichicon UHE2E220MHD (22µF, 250V, Radial w/adapter) or Cornell Dubilier 380LX220M160A022 (Axial).	$2.50	Located on bias rectifier board.
C14	Bias Output Filter	20 µF, 150V Electrolytic (Axial)	Nichicon UHE2E220MHD (22µF, 250V).	$2.50	Located on bias rectifier board.
C15	Relay Delay/Filter	20 µF, 150V Electrolytic (Axial)	Nichicon UHE2E220MHD (22µF, 250V).	$2.50	Near relay assembly.
C16	RF Input Coupling	0.02 µF (20 nF), 600V Paper/Plastic	WIMA FKP2D022001F00KSSD (0.022 µF, 630V Film, Axial).	$1.50	Input SO-239 connector to band switch.
C17	ALC Filter	0.1 µF, 50V Electrolytic	Nichicon UHE1H100MDD (10µF, 50V) or 0.1µF Film.	$0.80	Near ALC diode. Original was likely electrolytic; film is more reliable.
C18	Meter Lamp Bypass	0.02 µF, 50V Ceramic	Vishay K5050203Z (0.022 µF, 50V).	$0.80	Across meter lamp terminals.
C19	Relay Supply Filter	2.2 µF, 250V Electrolytic (Axial)	Nichicon UHE2E2R2MHD (2.2µF, 250V Axial) or Harbach CAP-221.	$3.75	Critical for relay timing. Often leaks in old units.
C20	Relay Supply Filter (Second)	2.2 µF, 250V Electrolytic (Axial)	Nichicon UHE2E2R2MHD (2.2µF, 250V Axial).	$3.75	Parallel with C19 or separate stage depending on revision.

Total Estimated Cost for Full Capacitor Rebuild

DIY (Individual Parts): Approximately $250 – $350 depending on whether you choose standard replacements or high-end upgrades (e.g., 220µF HV caps, doorknob replacements).
Kit Approach:
- Harbach PM-200 Kit (HV Supply only): ~$125 (includes C1-C6, diodes, resistors).
- Harbach Master Kit: ~$548 (includes HV kit, fan, soft-start, input caps, grid caps, suppressors, etc.). This is often more cost-effective than sourcing every individual part.

Simple ALC Monitor for the Heathkit SB-200

2026-07-21T02:14:00.000-07:00

Simple ALC Monitor for the Heathkit SB-200

The stock Heathkit SB-200 ALC circuit samples RF from one grid, causing imbalance, and can generate excessive negative voltage (up to -35V) that damages modern transceivers (which often require only -8V to -10V).

Here is a simple, safe ALC Monitor/Clamp design you can build externally or internally.

1. The Problem

Voltage Mismatch: The SB-200 can output -30V to -35V on the ALC line when overdriven.
Radio Limit: Most solid-state radios (Icom, Kenwood, Yaesu) are damaged or shut down by anything below -10V.
Imbalance: The stock sampling capacitor (C29) connects to only one tube grid.

2. Simple External ALC Clamp Circuit

This circuit connects between the SB-200's rear ALC jack and your transceiver. It clamps the voltage to a safe level while allowing you to monitor the ALC activity with an LED or meter.

Components:

D1: 1N4735A Zener Diode (6.2V) or 1N4737A (7.5V). Note: Connect Cathode to Ground, Anode to ALC Line.
D2: 1N4148 Signal Diode (Protection).
R1: 1 kΩ, ½ W Resistor (Current Limiter).
LED (Optional): Red LED + 4.7 kΩ Resistor (Visual ALC indicator).
Connector: RCA or 3.5mm plug (matching your radio).

ASCII Schematic:

       FROM SB-200 ALC JACK
            |
            |
          [ R1 ]  1 kΩ
            |
            +----------> TO TRANSCEIVER ALC INPUT
            |
          [ D1 ]  Zener (6.2V - 7.5V)
            | (Cathode to Ground, Anode to Line)
            |
          [ D2 ]  1N4148 (Anode to Ground, Cathode to Line)
            |
          GROUND

How it works: When the SB-200 generates negative ALC voltage, D1 (Zener) conducts once the voltage exceeds its rating (e.g., -6.2V), clamping the line to a safe level. D2 protects against any positive voltage spikes.
Visual Monitor: To add an LED, connect the LED (with a 4.7 kΩ series resistor) from the ALC line (after R1) to a small positive voltage source (like a 9V battery) or use a dual-LED setup to show activity. Simpler: Just use a high-impedance voltmeter across the ALC line to monitor.

3. Internal Modification (Recommended)

For best performance, modify the internal sampling point to balance the tubes and limit voltage at the source.

Balance the Sample: Move the stock sampling capacitor C29 (5 pF) from the grid of one tube to the junction of the two 33 Ω grid resistors (R21/R22). This samples both tubes equally.
Add Internal Zener: Solder a 1N4737A (7.5V) Zener diode directly across the internal ALC output terminal and ground (Cathode to Ground, Anode to ALC line). This prevents the voltage from ever exceeding -7.5V internally.
Adjust Threshold: The stock ALC potentiometer can then be adjusted to set the "turn-on" point without fear of over-voltage.

4. Usage Tips

Setting ALC: Tune the amplifier to full power (600W). Increase drive until grid current is ~80-100mA. Adjust the ALC pot on the SB-200 until the ALC voltage just begins to appear (or your radio's ALC meter shows slight deflection).
Monitoring: Use a multimeter on the ALC line. If it exceeds -10V, reduce drive or adjust the ALC pot.
Safety: If your radio does not have an ALC input, do not connect the ALC line. Instead, carefully monitor grid current manually to avoid overdrive.

Robust VHF Parasitic Suppressor Design (Based on QST Oct 1990)

2026-07-20T09:00:00.000-07:00

Robust VHF Parasitic Suppressor Design (Based on QST Oct 1990)

Based on Richard L. Measures, AG6K’s article "Parasitics Revisited" (QST, Oct 1990, pp. 32–35), the most robust method for suppressing high-VHF oscillations (common in tubes like 3-500Z, 8874, and 572B) is the Low-Q "U-Inductor" (Hairpin) Suppressor.

Unlike traditional coiled suppressors that can magnetically couple to their resistors (raising Q and narrowing bandwidth), the U-design ensures magnetic decoupling, creating a broad, low-Q loss network that effectively swamps oscillations from 100 MHz to over 500 MHz.

1. Design Theory & Mathematics

The goal is to lower the Q of the anode circuit's self-resonance without introducing significant loss at HF (3–30 MHz).

Inductance ($L_s$): The U-loop acts as a low-Q inductor.
- Target: 60 nH to 80 nH.
- Formula approximation for a U-loop: $L \approx 0.02 \cdot l \cdot [\ln(2l/r) - 0.75]$ (where $l$ is length in inches, $r$ is wire radius).
- Measures specifies a 3.5-inch span for 60 nH (3-500Z) and 4.0-inch span for 80 nH (lower voltage tubes like 572B).
Resistance ($R_s$): Placed across the open end of the U.
- Target: 100 Ω to 150 Ω total.
- Configuration: Two or three 100 Ω, 2W metal-film resistors in parallel (yielding 50 Ω or 33 Ω) OR a single 100 Ω depending on the specific tube's VHF gain. Measures suggests starting with two 100 Ω resistors in parallel (50 Ω) or just one 100 Ω if space is tight, but explicitly cites two or three 100-Ω, 2-W metal-film resistors soldered across the U. Correction based on text: The text says "two or three 100-Ω... resistors soldered across the U." If placed in parallel, this lowers resistance. If placed in series/parallel combo, it adjusts wattage. The standard implementation is often two 100Ω resistors in parallel to handle power and lower resistance to ~50Ω, or simply one 100Ω if the text implies a single equivalent. "consists of two or three 100-Ω, 2-W metal-film resistors soldered across the U." This implies a parallel arrangement to increase power handling and lower the net resistance to swamp the high gain. Let's assume a net 50 Ω to 33 Ω load across the inductor for high-gain tubes.
Material: Nichrome 60 (NiCr) wire (No. 18 AWG).
- Why? Nichrome has higher resistivity than copper, lowering the Q of the inductor itself. It also withstands high heat.
Geometry: The "U" shape minimizes magnetic coupling between the inductor legs and the resistor, preventing the formation of a high-Q transformer effect.

2. Component List

Wire: No. 18 AWG Nichrome 60 wire (approx. 4.5 inches total length).
Resistors: Two (2) 100 Ω, 2 Watt Metal-Film resistors (1% tolerance).
- Note: Do not use carbon composition here; metal film is non-inductive and stable.
Solder: High-temperature silver solder (e.g., Stay-Brite) and appropriate flux, as nichrome is difficult to solder with standard rosin.

3. ASCII Schematic

       TO TUBE PLATE (Anode)
            |
            |  (No. 18 Nichrome Wire)
            +-----------------------+
            |                       |
            |                       |  <-- Leg Length ~ 1.75" each
            |                       |      (Total Span A-B = 3.5")
            |                       |
            |                       |
            +-----------+-----------+
            |           |           |
          [ R1 ]      [ R2 ]      [ R3 ]  (Optional 3rd resistor)
        (100 Ω)     (100 Ω)     (100 Ω)
            |           |           |
            +-----------+-----------+
            |
            |  (No. 18 Nichrome Wire)
            |
       TO TANK CIRCUIT (Output Capacitor/Choke)

Dimensions:
- Span A to B (Resistor mounting points): 3.5 inches (for 3-500Z/8874)
- Leg Height: ~1.0 to 1.5 inches (determined by resistor length)
- Wire Gauge: #18 AWG Nichrome 60

4. Construction Steps

Cut Wire: Cut a 4.5-inch piece of No. 18 Nichrome wire.
Form U-Shape: Bend the wire into a "U" shape. The distance between the two parallel legs (Points A and B) should be 3.5 inches for high-voltage tubes (3-500Z, 8874) or 4.0 inches for lower voltage tubes (572B, 811A).
Mount Resistors: Solder the two 100 Ω metal-film resistors across the open ends of the U (Points A and B).
- Tip: Keep the resistor leads short to minimize stray inductance.
- Power: Using two resistors in parallel yields 50 Ω at 4 Watts total dissipation capability.
Install: Connect one leg of the U to the tube's plate pin (or anode cap) and the other leg to the tank circuit input.
Orientation: In multi-tube amplifiers, orient adjacent U-suppressors at 90° angles to each other to prevent magnetic coupling and push-pull parasitics.

5. Why This Works Better

Decoupling: The straight legs of the U and the perpendicular resistor placement ensure the magnetic field of the inductor does not couple into the resistor, maintaining a low Q.
Broadband: Unlike a coiled suppressor which might only dampen a narrow frequency, the U-inductor provides effective suppression over a wide VHF/UHF range (100 MHz – 1 GHz).
Thermal Stability: Nichrome wire and metal-film resistors can withstand the heat generated during 10-meter operation or a parasitic event without drifting in value.

Citations

Measures, R. L., AG6K. "Parasitics Revisited." QST, October 1990, pp. 32–35. (Specifically Fig. 7 and associated text).
Measures, R. L., AG6K. "Improved Anode Parasitic Suppression for Modern Amplifier Tubes." QST, October 1988, pp. 36–38, 66, 89.
Measures, R. L., AG6K. "Calculating Power Dissipation in Parasitic-Suppressor Resistors." QST, March 1989, pp. 25–28.

Technical Breakdown: The 3-500Z High-Mu Power Triode

2026-07-17T21:02:44.834-07:00

Technical Breakdown: The 3-500Z High-Mu Power Triode

The 3-500Z is a legendary high-mu (high amplification factor) power triode developed by Eimac (now part of CPI). It revolutionized amateur radio amplifiers in the 1970s by enabling high-power, zero-bias operation in a compact, radiation-cooled envelope. It remains a standard for grounded-grid linear amplifiers today.

1. Core Specifications

Parameter	Value	Notes
Type	Thoriated Tungsten Filament Triode	Directly heated cathode
Amplification Factor ($\mu$)	130 (Average)	High mu allows zero-bias operation
Maximum Plate Dissipation	500 Watts	Continuous duty
Maximum Plate Voltage	4,000 VDC	Absolute maximum rating
Maximum Plate Current	400 mA	DC average
Maximum Grid Dissipation	20 Watts	Critical limit for grounded-grid
Filament Voltage ($V_f$)	5.0 VAC ($\pm$0.25V)	Must be regulated at socket pins
Filament Current ($I_f$)	14.6 Amps	High inrush current potential
Max Operating Frequency	110 MHz	Usable well into 6 meters
Interelectrode Capacitance	$C_{in}$: 8.3 pF, $C_{out}$: 4.7 pF, $C_{gp}$: 0.07 pF	Grounded filament configuration
Base Type	5-Pin Special (Giant)	Requires SK-410 or equivalent socket
Cooling	Radiation & Forced Air	Air must flow over plate seal
Dimensions	6.10" Length x 3.44" Diameter	~155mm x 87mm

2. Operational Characteristics

Zero-Bias Design

The 3-500Z is designed for Class AB2 operation with zero fixed bias.

Simplification: Eliminates the need for a negative bias power supply.
Grounded Grid: Ideally suited for grounded-grid circuits where the cathode is driven. This configuration provides a power gain of approximately 15x to 20x.
Idling Current: Typically idles at 50–70 mA per tube at 3,000V.

Filament Management

The thoriated tungsten filament is sensitive to voltage variations.

Voltage Tolerance: Strictly 4.75V to 5.25V measured at the socket pins.
- Over-voltage (>5.25V): Drastically shortens tube life due to evaporation of the thorium layer.
- Under-voltage (<4.75V): Causes cathode poisoning and reduced emission.
Inrush Current: Cold filament resistance is very low. Inrush current can exceed 60–80 Amps initially.
- Requirement: A step-start circuit (thermistor or resistor/relay delay) is mandatory to limit inrush to <2x operating current (~30A) to prevent filament stress and grid-to-filament shorts.
Warm-up Time: Being directly heated, it is ready for RF application in 30–60 seconds, though 2 minutes is recommended for thermal stabilization.

3. Typical Amplifier Circuit Values (Single Tube)

For a grounded-grid linear amplifier operating at 3,000 VDC:

Plate Voltage ($E_b$): 3,000 – 3,500 VDC
Idle Plate Current ($I_q$): 60 – 70 mA
Peak Plate Current: ~750 mA (SSB)
Input Impedance: ~50 – 75 $\Omega$ (Non-linear, requires broadband transformer or pi-network)
Output Load Impedance ($R_L$): ~4,000 – 5,000 $\Omega$
Drive Power Required: ~60 – 80 Watts for ~1,000 Watts PEP output.
Parasitic Suppression: Essential due to high $\mu$ and low $C_{gp}$.
- Plate: U-inductor (hairpin) with 100$\Omega$ resistor (as per AG6K designs).
- Grid/Cathode: 10–30$\Omega$ non-inductive resistor in series with grid pin to ground (if not directly grounded) or filament choke upgrades (20–40 $\mu$H).

4. Mechanical & Thermal Considerations

Cooling Airflow: Although "radiation cooled," reliable operation at legal limits requires forced air.
- Direction: Air should be blown upward through the socket chimney, across the glass envelope, and over the plate seal/anode connector.
- Flow Rate: Approx. 20–30 CFM for a pair of tubes in a confined chassis.
Socket: The SK-410 (or PSK410) ceramic socket is standard. It provides proper pin spacing and a chimney mount for airflow.
Orientation: Can be operated base-up or base-down, but base-down (anode up) is preferred for natural convection if forced air fails.

5. Common Failure Modes & Mitigation

Grid-to-Filament Shorts: Often caused by excessive filament inrush current or mechanical shock.
- Fix: Implement a soft-start circuit; handle tubes gently.
Plate Seal Overheating: The glass-to-metal seal at the anode is the thermal bottleneck.
- Fix: Ensure airflow is directed specifically over the plate cap/seal, not just the glass bulb.
VHF Parasitics: The high $\mu$ makes the tube prone to oscillation above 100 MHz.
- Fix: Install low-Q plate suppressors (Nichrome hairpins) and keep grid leads extremely short.
Cathode Stripping: Applying high voltage before the filament is hot.
- Fix: Use a time-delay relay to interlock HV until filaments have warmed for at least 30 seconds.

Citations

Eimac Care and Feeding of Power Grid Tubes, CPI Electron Tube Business Unit.
3-500Z High-Mu Power Triode Data Sheet, Eimac Division of Varian/CPI.
Measures, R. L., AG6K. "Parasitics Revisited." QST, October 1990.
W7BRS. "3-500Z Data Sheet and Amplifier Construction Notes." w7brs.com.
The ARRL Handbook for Radio Communications, various editions (Chapter on Power Amplifiers).

Conductivity Comparison Table

2026-07-17T20:58:05.545-07:00

Conductivity Comparison Table

Material	Resistivity ($\Omega\cdot$m)	Relative to Copper	Conductivity Rating
Silver	$1.59 \times 10^{-8}$	0.95	Best Conductor
Copper (Annealed)	$1.72 \times 10^{-8}$	1.00	Standard Baseline
Gold	$2.44 \times 10^{-8}$	1.42	Excellent (Corrosion Resistant)
Aluminum	$2.82 \times 10^{-8}$	1.64	Very Good (Lightweight)
Calcium	$3.36 \times 10^{-8}$	1.95	Good (Reactive)
Tungsten	$5.60 \times 10^{-8}$	3.26	Moderate (High Heat)
Zinc	$5.90 \times 10^{-8}$	3.43	Moderate
Nickel	$6.99 \times 10^{-8}$	4.06	Moderate
Iron	$9.70 \times 10^{-8}$	5.64	Fair
Platinum	$1.06 \times 10^{-7}$	6.16	Fair
Tin	$1.09 \times 10^{-7}$	6.34	Fair
Carbon Steel	$1.43 \times 10^{-7}$	8.31	Poor
Lead	$2.20 \times 10^{-7}$	12.79	Poor
Titanium	$4.20 \times 10^{-7}$	24.42	Very Poor
Manganin	$4.82 \times 10^{-7}$	28.02	Resistor Alloy
Constantan	$4.90 \times 10^{-7}$	28.49	Resistor Alloy
Stainless Steel	$6.90 \times 10^{-7}$	40.12	Very Poor
Mercury	$9.80 \times 10^{-7}$	56.98	Liquid/Poor
Nichrome	$1.10 \times 10^{-6}$	63.95	Heating Element
Graphite (Parallel)	$2.5 \times 10^{-6}$	~145	Semiconductor/Resistive
Graphite (Perp.)	$3.0 \times 10^{-3}$	~174,000	Highly Resistive

How to Remove the SWR Bridge from Your Heathkit SB-200

2026-07-16T23:25:32.270-07:00

How to Remove the SWR Bridge from Your Heathkit SB-200

The internal SWR bridge in the Heathkit SB-200 is a convenient feature, but it can introduce insertion loss, become a point of failure, or limit peak power handling. Removing it creates a direct, low-loss path from the Pi-network to your antenna.

This guide covers the removal process with ASCII schematics, formatted for easy reading on your blog.

⚠️ High Voltage Warning: The SB-200 contains lethal voltages. Always unplug the unit, wait at least 5 minutes for capacitors to discharge, and short the high-voltage capacitor terminals to the chassis with an insulated screwdriver before touching any internal components.

1. Understanding the Stock Configuration

In the stock SB-200, the RF path flows from the Pi-network output capacitors, through the directional coupler (SWR bridge), and finally to the rear SO-239 antenna connector.

Stock RF Path:

[ Pi-Network Output ] -----> [ SWR Bridge ] -----> [ Antenna Connector ]
                                  |
                                  +-----> [ Detector Diodes ] -----> [ Front Meter ]

2. Tools and Parts Needed

Soldering Iron & Solder Sucker: For desoldering connections.
Heavy Gauge Wire or Short Coax: RG-142, RG-400, or #12/#14 solid copper wire for the jumper.
Screwdrivers: To remove the chassis covers and mounting hardware.
Multimeter: To verify continuity after modification.

3. Step-by-Step Removal Guide

Step 1: Access the Chassis

Remove the outer cabinet and the top inner shield. Locate the SWR Bridge assembly near the rear of the chassis, situated between the large output tuning capacitors and the rear panel SO-239 connector.

Step 2: Disconnect the Detector Wires

The bridge has two small coaxial cables (usually RG-316) running from the internal diodes to the front-panel meter switch.

Desolder or unplug these two small coax connectors from the circuit board or terminal strip.
Tape the ends of these wires to prevent them from shorting against the chassis.

Step 3: Remove the Bridge Assembly

Input Side: Desolder or unbolt the connection coming from the Pi-Network Output (usually from the junction of the loading capacitors and the band switch).
Output Side: Desolder or unbolt the connection going to the Antenna Connector.
Unscrew the mounting hardware holding the bridge box to the chassis and remove it completely.

Step 4: Install the Direct Jumper

You must now create a direct link between the Pi-network and the antenna connector.

Modified RF Path:

[ Pi-Network Output ] =================================== [ Antenna Connector ]
           |                                                 (SO-239)
           |
      (No Bridge)

Measure: Cut a piece of heavy wire or short coax just long enough to span the gap between the Pi-network output lug and the center pin of the SO-239 connector.
Solder: Solder one end to the Pi-network output point (where the bridge input was).
Connect: Solder the other end directly to the center lug of the rear SO-239 antenna connector.
Insulate: Ensure the jumper does not touch the chassis or other components. Use heat-shrink tubing if necessary.

4. Verification and Testing

Continuity Check: Set your multimeter to resistance/continuity. Place one probe on the Pi-network output and the other on the center pin of the SO-239. You should read 0 Ω (or very close to it).
Ground Check: Ensure there is no continuity between your new jumper and the chassis ground.
Reassemble: Replace the top shield and cabinet.
Operational Test:
- Connect an external SWR meter between the amplifier and your antenna/dummy load.
- Tune the amplifier on a low band (40m) and then 10m.
- Verify that power output is stable and SWR readings on your external meter are consistent.

5. What You Lose (and Gain)

Gain: You eliminate ~0.1–0.2 dB of insertion loss and remove a potential point of arcing or failure. The amplifier is now slightly more robust for high-power operation.
Loss: The front panel meter will no longer display SWR or Relative Power. You must rely on an external meter or your transceiver's internal readings for tuning and monitoring.

Pro Tip: If you ever want to restore the amp to stock condition, keep the removed SWR bridge and the small detector coax cables in a labeled bag. Future owners often appreciate original parts!

Disclaimer: Modifications to high-voltage equipment are performed at your own risk. Ensure all connections are secure and insulated before applying power.

How-To: VHF Stability & Grid Protection for the Heathkit SB-200

2026-07-16T14:24:22.190-07:00

Stabilizing the Heathkit SB-200: A Guide to VHF Parasitic Suppression & Grid Protection

The Heathkit SB-200 is a legendary amplifier, but like many vintage designs, it can suffer from VHF instability (parasitics) when using modern 572B tubes or operating on the 10-meter band.

The following guide adapts proven engineering techniques—specifically cathode input swamping and grid fuse protection—to ensure your amplifier remains stable and your tubes stay safe.

⚠️ Safety Warning: This project involves high voltage and internal amplifier modifications. Always disconnect power, wait for capacitors to discharge, and short the high-voltage points to ground before touching internal components. If you are unsure, consult a qualified technician.

1. The Theory: Why Modify?

Modern 572B tubes can exhibit different internal capacitance characteristics than the originals. At Very High Frequencies (VHF), energy can feed back from the plate to the cathode through the tube's internal capacitance, causing self-oscillation (parasitics).

To stop this, we create a low-resistance path to ground for VHF signals at the cathode and grid, effectively "swamping" the unwanted energy before it can build up.

2. Modification A: Cathode Input Swamping

This technique adds a small RC (Resistor-Capacitor) network from the filament (cathode) to the chassis. While large tubes (like the 3-1000Z) use tiny values, the SB-200's 572B tubes require slightly different components.

Recommended Values for SB-200

Capacitor: 1.0 nF to 1.5 nF (Ceramic Disc or Mica, 1KV+ rating). Note: The 25pF value cited for large tubes is too small for 572Bs.
Resistor: 1 Ω to 10 Ω (Non-inductive metal film, ½ W).
Location: Directly at the tube socket pins.

Installation Steps

Locate the Filament Pins: On the 572B socket, identify the filament pins (typically Pins 1 & 4, but verify with a tube manual).
Mount the Components: Solder the capacitor from each filament pin to the nearest chassis ground.
Add the Resistor: Place the small resistor in series with the capacitor if you notice self-resonance issues, though often the capacitor alone works if leads are kept extremely short.
Tuning Check: Adding capacitance here changes the input impedance. You may need to slightly adjust the Input Tuning Capacitor on the front panel to restore a low SWR on 10 meters.

3. Modification B: Grid Fuse Resistor

This is a critical safety mod. It protects the grid from melting if a parasitic oscillation occurs and dampens grid resonance.

Parts List

Resistor: 24 Ω to 30 Ω, ½ W (Carbon Composition or Metal Film).
Bypass Capacitor: 1.8 nF to 2.7 nF (Ceramic or Mica).

Installation Steps

Identify the Grid Pin: For the 572B, this is usually Pin 3.
Connect to Ground: Solder the resistor from the Grid Pin directly to the chassis ground. Keep leads as short as possible.
Bypass for RF: Solder the bypass capacitor directly across the resistor (from Grid Pin to Chassis). This ensures the resistor doesn't burn up during normal operation but still acts during VHF events.

4. Schematic Diagram

Below is a simplified schematic of the protection network.

       TUBE SOCKET (572B)
      +----------------+
      |                |
      |  [Cathode]     |
      |  [Filament]    |---||---+ (To Chassis)
      |                |   1nF  |
      |  [Grid] o------+--------+------[ R_grid ]------+------||------+
      |  (Pin 3)       |        |    (24-30 Ohm)      |      C_byp    |
      |                |        |   (Non-Inductive)   |   (1.8nF+)    |
      |  [Plate]       |        |                     |               |
      +----------------+        |                     |               |
                                |                     |               |
                              CHASSIS GROUND (Frame) <---------------+

5. Tuning and Verification

Because the parasitic frequency depends on internal tube inductance, you cannot measure it with a standard meter. You must use trial and error.

Initial Test: Install the components with the values listed above.
Low Power Check: Apply low drive power from your transceiver. Sweep across 10 meters and 15 meters.
Monitor: Watch for sudden jumps in grid current or output power drops, which indicate instability.
Adjust:
- If unstable, try shortening the leads on your new components.
- If SWR is high on 10m, adjust the amplifier's input tuning capacitor slightly.

Pro Tip: Many SB-200 owners also upgrade the Input Network capacitors on 10/15 meters to compensate for stray capacitance. If you still have high SWR after these mods, consider rewinding the input coils or upgrading the silver mica capacitors in the input bank.

Why Suppressors Heat Up on 10 Meters

The heating occurs because the 10-meter band (28–29.7 MHz) is the closest in frequency to the amplifier's natural VHF parasitic oscillation frequency (typically above 30 MHz). The parasitic suppressor is a low-Q network (usually a resistor in parallel with a small inductor) designed specifically to dissipate energy at these high frequencies to prevent instability.

Mechanism: When the amplifier operates near the suppressor's effective frequency range, the network absorbs RF energy that would otherwise cause oscillation. This absorbed energy is converted into heat by the resistors.
Visual Confirmation: It is common for the nichrome wire or resistors in the suppressor to glow a slight orange when keyed up on 10 meters. This indicates the suppressor is functioning correctly by dumping excess VHF energy.
Design Trade-off: An effective VHF parasitic suppressor must dissipate power to be effective. If it remains cool on 10 meters, it may indicate the suppressor is not effectively suppressing parasitics, which could lead to arcing or tube damage.

Adjusting Suppressors for 10 Meters

If the suppressors are heating up excessively early in the 10-meter band (e.g., below 28.3 MHz) or causing significant power loss, the inductance of the suppressor coil may need adjustment.

Hairpin or U-Inductor Types: The inductance ( $L_{s}$ ) can be lowered by squeezing the sides of the U-shaped inductor closer together. Lowering the inductance raises the cut-off frequency, moving the peak dissipation higher up the band or beyond it.
Conventional Coils: Spreading the turns of the coil lowers the inductance.
Goal: The objective is to raise the frequency where maximum suppression (and thus maximum heating) occurs, ensuring the amplifier remains stable across the entire 10-meter band without excessive heating at the lower end.

Common Issues and Solutions

While heating is normal, component failure related to the suppressors is a known issue in older or modified SB-200 units.

Resistor Failure: The carbon composition resistors originally used in the suppressors often drift in value or fail due to heat cycling. A broken resistor can arc to adjacent components. Many restorers replace these with metal-film resistors or install updated kits (such as those from Harbach Electronics or Rich Measures/AG6K).
Power Loss: It is inherent to the SB-200 design to produce 100 to 150 watts less output on 10 meters compared to 40 meters. If the loss exceeds this, check the suppressor resistors for value changes before oscillation damages the plate tuning capacitor or band switch.
Cooling: Because the suppressors and tubes generate significant heat on lower bands (80m/40m) and VHF frequencies, adequate cooling is essential. Some operators upgrade the original fans to higher-flow models to prevent the chassis from becoming too hot to touch.

Heathkit SB-200 power supply Rebuild notes ( for 8-10 Capacitors not suggested just notes)

2026-07-12T14:01:14.345-07:00

PLEASE BE AWARE THAT THIS POST IS SAVED NOTES AND NOT THE BIBLE OF WHAT TO USE OR DO.

The original string used lower-rated diodes. For reliability with 220µF capacitors (and especially for Turbo mode), upgrade to 1000V+ 3A diodes. You need 10 diodes total (5 per leg of the doubler).

Quantity: 10

Part Number: 1N5408 (1000V, 3A, Axial)

Alternative: BY225 (1300V, 3A) for extra voltage headroom.

Mouser: 512-BY225 or 625-1N5408

DigiKey: 1N5408E3/54GICT-ND

2. Bleeder / Equalizing Resistors

Replace the original high-wattage wirewound resistors with modern 2W Metal Oxide or Carbon Film resistors. With 220µF caps, 82kΩ or 100kΩ provides perfect voltage balancing and safe discharge (~2 minutes) without excessive heat.

Quantity: 8 (One across each capacitor)

Resistance: 100 kΩ (Preferred for cooler operation) or 82 kΩ

Power Rating: 2 Watt (Minimum), 3 Watt (Preferred for Turbo mode)

Part Number: CFR-200JB-100K (2W, 5%) or MOX-200JB-100K (2W Metal Oxide)

Mouser: 299-100K-RC (Ohmite 2W)

DigiKey: 100KQBK-ND

3. HV Output "Glitch" Resistor (Highly Recommended)

While not originally on the doubler PCB, it is standard practice to add a surge-limiting resistor in the HV output line to protect tubes and the board during flashovers.

Quantity: 1

Resistance: 15 Ω to 47 Ω

Power Rating: 10 Watt (Wirewound or Ceramic Composition)

Part Number: OH150E (Ohmite 15Ω 10W Ceramic) or RW01015R00 (Vishay 15Ω 10W Non-Inductive)

Mouser: 290-15-RC

DigiKey: 15F10-ND

4. HV Fuse Holder & Fuse (Safety Upgrade)

Replace the original hard-wired fuse link with a panel-mount holder for serviceability.

Quantity: 1 Holder, 1 Fuse

Holder: BFN-1000 (Bussmann 10kV Fuse Holder) or equivalent ceramic HV holder.

Fuse: 2.0 Amps, 10 kV, Fast-Blow, Ceramic Body.

Part Number (Fuse): 50200 (Bussmann)

Mouser: 504-50200 (Fuse), 504-BFN-1000 (Holder)

5. RF Bypass Capacitors (Optional but Recommended)

Add these directly to the HV output point on the board to prevent RF feedback.

Quantity: 2

Value: 0.0022 µF (2200 pF)

Voltage: 3 kV to 5 kV (Ceramic Disk)

Part Number: 580-5KV222M (5kV, 2200pF)

Mouser: 580-5KV222M

DigiKey: 493-15715-ND (Similar Vishay)

Summary Shopping List

Component Value / Spec Qty Example Part Number

Rectifier Diode 1000V, 3A, Axial 10 1N5408

Bleeder Resistor 100kΩ, 2W (or 3W) 8 CFR-200JB-100K

Glitch Resistor 15Ω, 10W, Wirewound 1 OH150E

HV Fuse 2A, 10kV, Ceramic 1 Bussmann 50200

HV Fuse Holder 10kV Rating 1 Bussmann BFN-1000

RF Bypass Cap 2200pF, 3kV, Ceramic 2 580-5KV222M

Installation Note: When installing the 10 bleeder resistors (8 across caps + 2 extra if modifying the string for Turbo), ensure leads are kept short and away from high-voltage arcs. If performing the 8-capacitor Turbo mod, you will need 8 bleeder resistors total (one for each cap). The list above assumes the standard 6-cap board plus 2 added elsewhere, or a full refresh of the 6 original positions plus spares. If you are doing the 8-cap mod, simply ensure you have 8 resistors installed (one per cap).

Diagnosing a bad Harbach soft key modification in a Heathkit SB-200 amplifier

2026-07-12T13:58:32.110-07:00

Diagnosing a bad Harbach soft key modification in a Heathkit SB-200 amplifier involves verifying whether the modification is present, checking for correct wiring, and testing the keying behavior with and without the mod engaged. The Harbach soft key was designed to protect modern solid-state transceivers from the SB-200’s high negative keying voltage (approximately –120 VDC) by interposing a relay or FET. A faulty installation or failed components can cause keying issues such as the amp staying keyed, not keying at all, or damaging the transceiver.

Step-by-step instructions

Tools and Materials

Multimeter (capable of measuring DC voltage and continuity)

Dummy load (for safe testing)

Shorting plug or mechanical switch for keying jack

SB-200 schematic (preferably with Harbach mod annotations)

Insulated tools and safety equipment (high voltage is present)

Prerequisites

Ensure the amplifier is unplugged and discharged before opening the chassis.

Confirm whether the Harbach soft key mod is installed (look for a small PCB near the keying jack or antenna relay connections).

Have access to the original SB-200 schematic and, if possible, Harbach mod wiring instructions.

Diagnostic Steps

Visual Inspection

Locate the Harbach board inside the SB-200 chassis.

Check for obvious signs of damage: burnt components, cold solder joints, loose wires.

Verify that the keying jack (“Ant Relay”) is wired to the Harbach board and not directly to the original keying circuit.

Check Keying Voltage

With the amplifier powered on and warmed up, measure the DC voltage at the keying jack (RCA “Ant Relay” jack).

Without the Harbach mod, you should see approximately –120 VDC.

With a functioning Harbach mod, the voltage presented to the transceiver should be near 0 V, and the internal relay/FET should switch the high voltage internally.

Test Keying Function

Use a shorting plug or mechanical switch to simulate keying.

Observe whether the amplifier keys and unkeys properly:

If the amp stays keyed, the Harbach relay or FET may be stuck closed or miswired.

If the amp does not key, the control side of the Harbach circuit (transistor/FET) may be failed or incorrectly connected.

Bypass Test

Temporarily disconnect the Harbach board:

Disconnect the wire from resistor R16 from the Harbach board.

Disconnect the wire from the “Ant Relay” jack to the Harbach board.

Connect R16 directly to the “Ant Relay” jack (splice or extend wire if needed).

Use a mechanical switch or shorting plug to key the amp (do not use a solid-state transceiver in this configuration).

If the amp keys correctly in this configuration, the issue lies with the Harbach board.

Check Harbach Board Power and Ground

Verify that the Harbach board is receiving its supply voltage (often derived from the filament choke or another low-voltage source).

Check ground connections, especially the green wire to the filament choke, as poor grounding can prevent proper operation.

Component Testing

Test the relay or FET on the Harbach board for proper switching.

Check polarized components (diodes, capacitors) for correct orientation and functionality.

Replace any suspect components, especially if both Harbach boards (if multiple mods exist) show identical failures.

Reintegrate and Retest

After repairs, reconnect the Harbach board.

Test keying again with a shorting plug before connecting a transceiver.

Once confirmed working, test with a solid-state transceiver, monitoring for proper keying and absence of high voltage at the key line.

⚠️ Caution: Never use a modern solid-state transceiver to key an SB-200 unless you are certain the soft key mod is functioning correctly. The –120 VDC present on the key line can destroy transceiver keying circuits.

Heathkit SB-200 Harbach soft key modification troubleshooting

What parts are in a Harbach soft key?

How do I test the Harbach board relay?

Where to buy SB-200 soft key replacement parts?

/////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////

Primary Cause: Insufficient RF Bypassing

The most likely culprit is a lack of adequate RF bypass capacitors on the Harbach board wiring. When RF enters the keying circuit, it can rectify in the FET or control transistor, creating a DC bias that keeps the switch closed even after the keying signal is removed.

Specific Components to Check and Modify

1. Missing or Inadequate Bypass Capacitors The most common fix reported by users involves adding 0.01 µF (10nF) ceramic disc capacitors:

Install a 0.01 µF capacitor from each signal wire (excluding the ground wire) to the chassis ground, mounted as close to the Harbach board as possible.

Install an additional 0.01 µF capacitor directly across the Drain and Source terminals (the two outer legs) of the FET on the Harbach board. This prevents RF from sustaining the FET in the "on" state.

2. The FET (Field Effect Transistor)

The FET itself may be marginal or damaged by previous RF exposure. If adding bypass capacitors does not resolve the issue, the FET (typically an N-channel enhancement mode MOSFET) may need replacement.

RF energy can cause the FET to enter thermal runaway or latch up, effectively welding the circuit closed electronically.

3. Main T/R Relay Contacts (Secondary Possibility)

While the symptom points to the soft key, do not rule out the main T/R relay (K1) inside the SB-200.

If the T/R relay contacts have carbon buildup or pitting, RF can arc across the open contacts, creating a conductive path that mimics a "stuck" relay.

Action: Clean the main T/R relay contacts using a strip of printer paper with a drop of 91% isopropyl alcohol, or replace the relay if heavily pitted.

Recommended Troubleshooting Steps

Add Bypass Capacitors: Before replacing components, solder 0.01 µF ceramic capacitors from each wire (except ground) entering the Harbach board to the nearest ground point. Add one directly across the FET's outer terminals. This resolves the issue in the majority of reported cases.

Inspect Grounding: Ensure the green ground wire from the Harbach board is securely connected to the filament choke or a solid chassis ground. A poor ground prevents RF from draining properly.

Clean Main Relay: If the problem persists, clean the contacts of the internal SB-200 T/R relay to rule out contact welding or arcing.

SB-200 Plate Meter Design with GC9A01 & Arduino Nano

2026-07-12T13:55:14.914-07:00

SB-200 Plate Meter Design with GC9A01 & Arduino Nano

This design replaces the analog meter in your Heathkit SB-200 with a digital gauge on a 1.28" GC9A01 Round TFT. It measures plate current via a shunt resistor and displays it on a colorful, animated dial.

1. Hardware Connections

Critical Warning: The GC9A01 is a 3.3V logic device. The Arduino Nano is 5V. You must use a logic level converter or series resistors (220Ω–330Ω) on the SPI lines (MOSI, SCK, CS, DC) to prevent damage.

GC9A01 Pin Arduino Nano Pin Note

VCC 3.3V Do NOT use 5V

GND GND Common Ground

CS D10 Chip Select

DC D9 Data/Command

SDA (MOSI) D11 Data In (Use 220Ω resistor)

SCL (SCK) D13 Clock (Use 220Ω resistor)

RST D8 Reset (Optional, can tie to 3.3V if not used in code)

BL 3.3V Backlight (Or PWM pin for dimming)

Current Sensing Circuit (Shunt): The SB-200 cathode current (Plate + Grid) flows to ground. Insert a low-value shunt resistor in the cathode ground path.

Shunt Value: 0.1Ω to 0.5Ω (5W+ Wirewound).

At 500mA (0.5A), a 0.1Ω shunt drops 0.05V (50mV).

At 500mA, a 0.5Ω shunt drops 0.25V (250mV).

Recommendation: Use 0.22Ω @ 5W.

Max Drop: $0.5A \times 0.22\Omega = 0.11V$ (110mV). This is safe for the tube bias but requires amplification or careful ADC reading.

Wiring: Connect the shunt between the tube cathodes and the amplifier chassis ground. Connect the Arduino A0 to the tube side of the shunt and GND to the chassis side.

Note: The Arduino reads 0–1023 for 0–5V. 110mV is only ~22 counts. For better resolution, use the Internal 1.1V Reference (analogReference(INTERNAL)), which gives ~930 counts for 110mV.

Arduino Nano current shunt sensing circuit

View all

2. Required Libraries

Install these via the Arduino Library Manager:

Adafruit GFX Library

Adafruit GC9A01A Library (or GC9A01 by Bodmer if preferred)

3. Arduino Code

#include <SPI.h>

#include <Adafruit_GFX.h>

#include <Adafruit_GC9A01A.h>

// --- Pin Definitions ---

#define TFT_CS 10

#define TFT_DC 9

#define TFT_RST 8

#define SHUNT_PIN A0

// --- Configuration ---

#define SHUNT_RESISTANCE 0.22 // Ohms (Change to match your shunt)

#define MAX_CURRENT_MA 600 // Full scale deflection (mA)

#define REF_VOLTAGE 1.1 // Internal reference voltage (1.1V for better resolution)

// Initialize Display

Adafruit_GC9A01A tft = Adafruit_GC9A01A(TFT_CS, TFT_DC, TFT_RST);

// Gauge Geometry

#define CENTER_X 120

#define CENTER_Y 120

#define RADIUS 100

#define GAUGE_START_ANGLE 150 // Degrees (Left bottom)

#define GAUGE_END_ANGLE 390 // Degrees (Right bottom, wraps around)

void setup() {

Serial.begin(9600);

// Use Internal 1.1V Reference for higher resolution on small voltages

analogReference(INTERNAL);

tft.begin();

tft.fillScreen(GC9A01A_BLACK);

tft.setRotation(2); // Adjust based on USB port orientation

// Draw Static Gauge Background

drawGaugeBackground();

}

void loop() {

// Read ADC (0-1023 for 0-1.1V)

int sensorValue = analogRead(SHUNT_PIN);

// Convert to Voltage (1.1V ref)

float voltage = sensorValue * (REF_VOLTAGE / 1023.0);

// Convert to Current (Ohm's Law: I = V/R)

float currentA = voltage / SHUNT_RESISTANCE;

float currentMA = currentA * 1000.0;

// Clamp values

if (currentMA < 0) currentMA = 0;

if (currentMA > MAX_CURRENT_MA) currentMA = MAX_CURRENT_MA;

// Draw Needle

drawNeedle(currentMA);

// Display Numeric Value

displayNumericValue(currentMA);

delay(100); // Update rate

}

void drawGaugeBackground() {

// Outer Ring

tft.drawCircle(CENTER_X, CENTER_Y, RADIUS, GC9A01A_WHITE);

tft.drawCircle(CENTER_X, CENTER_Y, RADIUS - 5, GC9A01A_WHITE);

// Scale Ticks

for (int i = 0; i <= MAX_CURRENT_MA; i += 50) {

float angle = map(i, 0, MAX_CURRENT_MA, GAUGE_START_ANGLE, GAUGE_END_ANGLE);

int x1 = CENTER_X + (RADIUS - 10) * cos(radians(angle));

int y1 = CENTER_Y + (RADIUS - 10) * sin(radians(angle));

int x2 = CENTER_X + (RADIUS - 20) * cos(radians(angle));

int y2 = CENTER_Y + (RADIUS - 20) * sin(radians(angle));

if (i % 100 == 0) {

tft.drawLine(x1, y1, x2, y2, GC9A01A_WHITE); // Major tick

// Draw Numbers (Simplified positioning)

tft.setCursor(CENTER_X + (RADIUS - 35) * cos(radians(angle)),

CENTER_Y + (RADIUS - 35) * sin(radians(angle)));

tft.setTextColor(GC9A01A_WHITE);

tft.setTextSize(1);

tft.print(i/100); // Print 0, 1, 2...

} else {

tft.drawLine(x1, y1, x2, y2, GC9A01A_GRAY); // Minor tick

}

// Center Label

tft.setCursor(CENTER_X - 20, CENTER_Y + 40);

tft.setTextColor(GC9A01A_GREEN);

tft.setTextSize(2);

tft.print("mA");

}

void drawNeedle(float currentMA) {

static float lastAngle = -1;

float angle = map(currentMA, 0, MAX_CURRENT_MA, GAUGE_START_ANGLE, GAUGE_END_ANGLE);

// Simple erase/redraw (could be optimized with XOR or buffer)

// Clear previous needle (draw over with background color)

if (lastAngle >= 0) {

int x1 = CENTER_X + (RADIUS - 25) * cos(radians(lastAngle));

int y1 = CENTER_Y + (RADIUS - 25) * sin(radians(lastAngle));

tft.drawLine(CENTER_X, CENTER_Y, x1, y1, GC9A01A_BLACK);

}

// Draw new needle

int x2 = CENTER_X + (RADIUS - 25) * cos(radians(angle));

int y2 = CENTER_Y + (RADIUS - 25) * sin(radians(angle));

// Color changes based on current (Green -> Yellow -> Red)

uint16_t needleColor = GC9A01A_GREEN;

if (currentMA > 400) needleColor = GC9A01A_YELLOW;

if (currentMA > 550) needleColor = GC9A01A_RED;

tft.drawLine(CENTER_X, CENTER_Y, x2, y2, needleColor);

tft.fillCircle(CENTER_X, CENTER_Y, 5, GC9A01A_WHITE); // Center cap

lastAngle = angle;

}

void displayNumericValue(float currentMA) {

// Clear old number area

tft.fillRect(80, 160, 80, 30, GC9A01A_BLACK);

tft.setCursor(90, 165);

tft.setTextColor(GC9A01A_CYAN);

tft.setTextSize(2);

tft.print(currentMA, 0); // Print integer

}

4. Calibration & Safety

Calibration: Transmit a carrier into a dummy load. Adjust the SHUNT_RESISTANCE value in the code until the display matches a trusted external multimeter reading.

Safety: The shunt resistor is at RF Ground, but ensure the Arduino is powered by an isolated USB source (laptop battery or isolated supply) to prevent ground loops or RF feedback into the USB port.

Filtering: Add a 0.1µF capacitor across the shunt resistor to filter RF noise from the ADC reading.

This design utilizes the Hosyond 3.5" ESP32 (ST7796U) to replace the SB-200's mechanical meter with a digital multi-function display. It measures High Voltage (0–3000V), Plate Current (0–600mA), and Grid Current (0–150mA) using isolated sensing circuits safe for the ESP32's 3.3V ADC.

2026-07-12T13:53:16.349-07:00

This design utilizes the Hosyond 3.5" ESP32 (ST7796U) to replace the SB-200's mechanical meter with a digital multi-function display. It measures High Voltage (0–3000V), Plate Current (0–600mA), and Grid Current (0–150mA) using isolated sensing circuits safe for the ESP32's 3.3V ADC.

⚠️ Critical Safety Warning

Lethal Voltage: The SB-200 contains 2,400V+. The voltage divider must use high-voltage rated resistors (series string) to prevent arcing and resistor failure.

Isolation: The ESP32 ground MUST be connected to the SB-200 chassis ground (Common Ground) for these resistive dividers to work. If you cannot ensure a solid common ground, use Hall Effect sensors (e.g., ACS712/ACS758) or opto-isolators instead of direct resistive dividers.

RF Protection: All input lines to the ESP32 must have RF bypass capacitors and clamping diodes to prevent RF feedback from damaging the microcontroller.

1. Hardware Interface Design

The Hosyond module uses the ST7796U driver. We will use the TFT_eSPI library for the display and XPT2046_Touchscreen for input.

Sensing Circuit Schematics

A. High Voltage Meter (0–3000V DC)

The ESP32 ADC maxes at ~3.3V. We need a divider ratio of roughly 1000:1.

Input: HV Line (after glitch resistor).

Divider:

R1 (HV): Three 4.7 MΩ, 1W resistors in series (Total 14.1 MΩ, Rating ~9kV). Do not use a single resistor.

R2 (Low): 15 kΩ, 0.25W precision resistor.

Formula: $V_{out} = V_{in} \times \frac{15k}{14.1M + 15k}$. At 3000V, $V_{out} \approx 3.18V$.

Protection: Add a 3.3V Zener diode (cathode to signal, anode to GND) at the ESP32 input pin.

B. Plate Current Meter (0–600mA)

The SB-200 measures plate current across a 1.0Ω internal shunt (R12/R14 network).

Input: Across the existing 1.0Ω shunt resistor in the power supply return.

Scaling: At 600mA, voltage drop is $0.6V$. This is safe for ESP32 directly, but we add gain for resolution.

Op-Amp Buffer (Optional but Recommended): Use an LM358 (Single Supply) in non-inverting mode with Gain = 5.

$0.6V \times 5 = 3.0V$ (Full Scale).

Direct Connect (Simpler): Connect across the 1Ω shunt via a 1kΩ series resistor and 0.1µF cap to ground.

Max voltage ~0.6V. ESP32 reads this easily (approx. 400–500 ADC counts).

C. Grid Current Meter (0–150mA)

The SB-200 measures grid current across a 1.5Ω shunt.

Input: Across the 1.5Ω grid shunt resistor.

Scaling: At 150mA, voltage drop is $0.225V$.

Connection: Direct to ADC pin with 1kΩ series resistor and 0.1µF cap.

Wiring Diagram (ASCII)

SB-200 AMPLIFIER CHASSIS (HIGH VOLTAGE ZONE)

--------------------------------------------

[HV LINE +2400V]

.-----+-----.

| R1a 4.7M | \

'-----+-----' |

| | R1b 4.7M (Series String for HV Safety)

.-----+-----. |

| R1b 4.7M | /

'-----+-----'

.-----+-----.

| R1c 4.7M |

'-----+-----'

+-----------------------+

| |

[R2 15k] [Zener 3.3V]

| |

GND (Chassis) GND

+-------> To ESP32 GPIO 34 (HV_SENSE)

[PLATE SHUNT 1.0Ω] (In Power Supply Return)

+-------> To ESP32 GPIO 35 (PLATE_SENSE) via 1k Resistor + 0.1uF Cap to GND

GND

[GRID SHUNT 1.5Ω] (In Grid Circuit)

+-------> To ESP32 GPIO 32 (GRID_SENSE) via 1k Resistor + 0.1uF Cap to GND

GND

HOSYOND ESP32 MODULE

--------------------

GPIO 34 (VP) --> HV_SENSE

GPIO 35 (VN) --> PLATE_SENSE

GPIO 32 --> GRID_SENSE

GND --> SB-200 Chassis Ground (CRITICAL)

5V --> External 5V Supply (Do not power from SB-200 low-voltage rails if noisy)

2. Software Implementation

Required Libraries

Install these in Arduino IDE:

TFT_eSPI (by Bodmer) - Configure User_Setup_Select.h for ST7796 and ESP32.

XPT2046_Touchscreen (by Paul Stoffregen).

User_Setup.h Configuration (for TFT_eSPI)

Edit the library folder TFT_eSPI/User_Setups/Setup25_ST7796.h (or create a new setup):

#define ST7796_DRIVER

#define TFT_WIDTH 320

#define TFT_HEIGHT 480

#define TFT_MISO 12

#define TFT_MOSI 13

#define TFT_SCLK 14

#define TFT_CS 15

#define TFT_DC 2

#define TFT_RST 4

#define TOUCH_CS 33

#define LOAD_GLCD

#define LOAD_FONT2

#define LOAD_FONT4

#define LOAD_FONT6

#define LOAD_FONT7

#define LOAD_FONT8

#define SMOOTH_FONT

Main Code (SB200_Meter.ino)

#include <TFT_eSPI.h>

#include <XPT2046_Touchscreen.h>

#include <SPI.h>

// --- Pin Definitions ---

#define HV_PIN 34

#define PLATE_PIN 35

#define GRID_PIN 32

// --- Calibration Constants (Adjust based on multimeter) ---

// ADC Range: 0-4095 (3.3V).

// HV: 3000V = ~3.18V -> ADC ~3930. Factor = 3000/3930 ≈ 0.763

const float HV_FACTOR = 0.763;

// Plate: 1.0 Ohm shunt. 0.6V (600mA) = ADC ~743. Factor = 600/743 ≈ 0.807 (mA per count)

const float PLATE_FACTOR = 0.807;

// Grid: 1.5 Ohm shunt. 0.225V (150mA) = ADC ~278. Factor = 150/278 ≈ 0.539

const float GRID_FACTOR = 0.539;

// --- Objects ---

TFT_eSPI tft = TFT_eSPI();

TS_Point tp;

void setup() {

Serial.begin(115200);

tft.init();

tft.setRotation(1); // Landscape

tft.fillScreen(TFT_BLACK);

// Draw UI Layout

drawGauge("HV", 0, 0, 160, 240, TFT_RED, 3000);

drawGauge("PLATE mA", 160, 0, 160, 240, TFT_GREEN, 600);

drawGauge("GRID mA", 0, 240, 320, 240, TFT_BLUE, 150);

pinMode(HV_PIN, INPUT);

pinMode(PLATE_PIN, INPUT);

pinMode(GRID_PIN, INPUT);

}

void loop() {

// Read ADC (Average 10 samples for stability)

int hv_raw = analogReadAvg(HV_PIN, 10);

int plate_raw = analogReadAvg(PLATE_PIN, 10);

int grid_raw = analogReadAvg(GRID_PIN, 10);

// Convert to Engineering Units

float hv_val = hv_raw * HV_FACTOR;

float plate_val = plate_raw * PLATE_FACTOR;

float grid_val = grid_raw * GRID_FACTOR;

// Update Displays

updateGauge(0, 0, 160, 240, hv_val, "kV", 100); // Scale HV to 30.00 kV display? No, keep Volts.

updateGaugeValue(0, 0, 160, 240, (int)hv_val, "V", TFT_RED);

updateGaugeValue(160, 0, 160, 240, (int)plate_val, "mA", TFT_GREEN);

updateGaugeValue(0, 240, 320, 240, (int)grid_val, "mA", TFT_BLUE);

delay(200); // 5Hz update rate

}

// --- Helper Functions ---

int analogReadAvg(int pin, int count) {

long sum = 0;

for (int i = 0; i < count; i++) {

sum += analogRead(pin);

delay(2);

}

return sum / count;

}

void drawGauge(String label, int x, int y, int w, int h, uint16_t color, int maxVal) {

tft.fillRect(x, y, w, h, TFT_DARKGREY);

tft.setTextColor(TFT_WHITE);

tft.setTextSize(2);

tft.setCursor(x + 10, y + 10);

tft.print(label);

// Draw Max Scale Line

tft.drawLine(x + 10, y + h - 30, x + w - 10, y + h - 30, TFT_WHITE);

tft.setTextSize(1);

tft.setCursor(x + w - 40, y + h - 50);

tft.print(maxVal);

}

void updateGaugeValue(int x, int y, int w, int h, int val, String unit, uint16_t color) {

// Clear Value Area

tft.fillRect(x + 10, y + 40, w - 20, h - 80, TFT_BLACK);

tft.setTextColor(color);

tft.setTextSize(4);

tft.setCursor(x + 10, y + 60);

tft.print(val);

tft.setTextSize(2);

tft.setCursor(x + 10 + tft.width(), y + 60); // Approximate unit pos

tft.print(" " + unit);

// Simple Bar Graph

int barW = map(val, 0, (unit == "V" ? 3000 : (unit == "mA" && x==160 ? 600 : 150)), 0, w - 20);

tft.fillRect(x + 10, y + h - 30, barW, 10, color);

tft.fillRect(x + 10 + barW, y + h - 30, (w - 20) - barW, 10, TFT_DARKGREY);

}

3. Assembly & Calibration Steps

Enclosure: Mount the Hosyond display on the SB-200 rear apron or in a custom external box. Do not place the ESP32 board inside the HV RF cage without shielding.

Wiring:

Run shielded cables from the sensing points (HV divider, shunts) to the ESP32.

Connect the ESP32 GND to the SB-200 Chassis Ground at a single point (star ground) to avoid ground loops.

Calibration:

Power up the SB-200 in Standby (Filament only). HV should read 0V.

Switch to Operate (No Drive). Adjust HV_FACTOR in code until the display matches a reliable HV probe multimeter (Target: ~2200V).

Apply drive to reach 500mA plate current. Adjust PLATE_FACTOR to match the stock meter or a clamp meter (if DC capable).

Touch Functionality (Optional):

The code includes the touch library. You can expand the loop() to detect touches (ts.touched()) to toggle between "SWR" mode (if you add an SWR bridge sensor) or "Peak Hold" modes.

High Voltage Resistor 4.7M 1W

ESP32 ST7796 Hosyond pinout configuration

Updated Guide: SB-200 Digital Meter with SWR & Peak Hold

This guide expands the previous ESP32 design to include SWR monitoring (via an external directional coupler) and PEP Peak Hold functionality. The system now displays HV, Plate Current, Grid Current, Forward/Reflected Power, SWR, and Peak Power.

1. Hardware Additions: SWR & Peak Sensing

To measure SWR and Peak Power, you cannot connect the RF directly to the ESP32. You must use a Directional Coupler (e.g., Monimatch) and RF Detectors.

A. Directional Coupler (External or Internal)

Placement: Install a dual-directional coupler (e.g., Bruene or Monimatch type) between the SB-200 output and the antenna connector.

Outputs: Provides two DC voltages proportional to Forward Power (FWD) and Reflected Power (REF).

Scaling: Ensure the coupler's output voltage does not exceed 3.0V at maximum amplifier power (600W). Use a potentiometer or voltage divider if necessary.

B. Peak Hold Circuit (Hardware vs. Software)

Software Peak Hold (Recommended): The ESP32 samples the ADC rapidly (kHz range). You can implement a "decay" algorithm in code to simulate a peak-hold meter without extra components. This is more flexible for SSB/CW.

Hardware Peak Hold (Optional): If you want to smooth the RF envelope before the ADC:

Use a Schottky Diode (1N5711) -> 1nF Capacitor -> 100kΩ Resistor to Ground.

Connect the junction of the Cap/Resistor to the ESP32 ADC.

C. Updated Wiring Diagram

DIRECTIONAL COUPLER (Installed at Amp Output)

---------------------------------------------

[RF OUT] ---> To Antenna

[RF IN] <--- From Amp Tank Circuit

Coupler Outputs (DC Voltages proportional to Power):

FWD Output o----[1kΩ]----+-----> ESP32 GPIO 33 (FWD_SENSE)

[0.1µF]

GND

REF Output o----[1kΩ]----+-----> ESP32 GPIO 32 (REF_SENSE)

[0.1µF]

GND

NOTE: Ensure FWD/REF voltages are < 3.3V at max power.

Use a 10k Potentiometer as a voltage divider if needed.

Updated ESP32 Pinout:

GPIO 34: HV Sense

GPIO 35: Plate Current Sense

GPIO 32: Grid Current Sense (Moved from 32 to GPIO 33 if using 32 for REF, or use GPIO 33 for FWD and GPIO 25 for REF).

Correction: Let's use GPIO 33 for FWD and GPIO 25 for REF to keep ADC1 channel consistency.

GPIO 33: Forward Power Sense

GPIO 25: Reflected Power Sense

2. Software Implementation (SWR & Peak Hold)

The code now includes:

SWR Calculation: $SWR = \frac{1 + \sqrt{P_{ref}/P_{fwd}}}{1 - \sqrt{P_{ref}/P_{fwd}}}$

Peak Hold Algorithm: Captures the maximum value and decays it slowly over time.

Touch Control: Touch the screen to reset the Peak Hold or toggle display modes.

Updated Code (SB200_Meter_Advanced.ino)

#include <TFT_eSPI.h>

#include <XPT2046_Touchscreen.h>

#include <SPI.h>

#include <math.h>

// --- Pin Definitions ---

#define HV_PIN 34

#define PLATE_PIN 35

#define GRID_PIN 32

#define FWD_PIN 33

#define REF_PIN 25

#define TOUCH_CS 33 // Note: Conflict with FWD_PIN if on same SPI. Use GPIO 21 for TOUCH_CS if needed.

// Fix: Reassign Touch CS to GPIO 21 in User_Setup.h or wire physically to 21.

// For this example, assume TOUCH_CS is on GPIO 21 (modify wiring/setup)

#define TOUCH_CS_PIN 21

// --- Calibration Constants (ADJUST THESE) ---

const float HV_FACTOR = 0.763;

const float PLATE_FACTOR = 0.807;

const float GRID_FACTOR = 0.539;

const float PWR_FACTOR = 0.05; // Adjust based on coupler (Volts to Watts)

// --- Peak Hold Variables ---

float peakFwd = 0;

float peakTime = 0;

const float DECAY_TIME = 1500; // ms for peak to decay

// --- Objects ---

TFT_eSPI tft = TFT_eSPI();

XPT2046_Touchscreen ts(TOUCH_CS_PIN);

void setup() {

Serial.begin(115200);

tft.init();

tft.setRotation(1);

tft.fillScreen(TFT_BLACK);

ts.begin();

// Draw Layout

drawPanel("HV", 0, 0, 160, 120, TFT_RED);

drawPanel("PLATE mA", 160, 0, 160, 120, TFT_GREEN);

drawPanel("GRID mA", 0, 120, 160, 120, TFT_BLUE);

drawPanel("SWR", 160, 120, 160, 120, TFT_YELLOW);

drawPanel("PEP W", 0, 240, 320, 240, TFT_ORANGE); // Wide bar for PEP

pinMode(HV_PIN, INPUT);

pinMode(PLATE_PIN, INPUT);

pinMode(GRID_PIN, INPUT);

pinMode(FWD_PIN, INPUT);

pinMode(REF_PIN, INPUT);

}

void loop() {

// 1. Read Sensors

float hv = analogReadAvg(HV_PIN, 5) * HV_FACTOR;

float plate = analogReadAvg(PLATE_PIN, 5) * PLATE_FACTOR;

float grid = analogReadAvg(GRID_PIN, 5) * GRID_FACTOR;

int fwdRaw = analogReadAvg(FWD_PIN, 5);

int refRaw = analogReadAvg(REF_PIN, 5);

// Convert to Power (Simplified linear approx for demo, square law for diodes)

float fwdVolts = (fwdRaw / 4095.0) * 3.3;

float refVolts = (refRaw / 4095.0) * 3.3;

// Diode correction (approx 0.3V drop for Schottky)

if(fwdVolts < 0.3) fwdVolts = 0; else fwdVolts -= 0.3;

if(refVolts < 0.3) refVolts = 0; else refVolts -= 0.3;

float fwdPwr = sq(fwdVolts) * 100; // Calibration factor needed here

float refPwr = sq(refVolts) * 100;

// 2. Peak Hold Logic

if (fwdPwr > peakFwd) {

peakFwd = fwdPwr;

peakTime = millis();

} else if (millis() - peakTime > DECAY_TIME) {

peakFwd *= 0.95; // Decay

if(peakFwd < 1) peakFwd = 0;

}

// 3. SWR Calculation

float swr = 1.0;

if (fwdPwr > 5 && refPwr > 0) {

float ratio = sqrt(refPwr / fwdPwr);

if (ratio < 1.0) swr = (1.0 + ratio) / (1.0 - ratio);

if (swr > 9.9) swr = 9.9;

} else if (fwdPwr < 5) {

swr = 0; // No power

}

// 4. Update Display

updateValue(0, 0, 160, 120, (int)hv, "V", TFT_RED);

updateValue(160, 0, 160, 120, (int)plate, "mA", TFT_GREEN);

updateValue(0, 120, 160, 120, (int)grid, "mA", TFT_BLUE);

// SWR Color Code

uint16_t swrColor = TFT_GREEN;

if(swr > 1.5) swrColor = TFT_YELLOW;

if(swr > 2.5) swrColor = TFT_RED;

updateValueFloat(160, 120, 160, 120, swr, "", swrColor);

// PEP Display

updateValueWide(0, 240, 320, 240, (int)peakFwd, "W (PEAK)", TFT_ORANGE);

// 5. Touch Reset (Optional)

if (ts.touched()) {

peakFwd = 0; // Reset Peak on touch

}

delay(100);

}

// --- Helper Functions (Same as before + new ones) ---

int analogReadAvg(int pin, int count) {

long sum = 0;

for (int i = 0; i < count; i++) sum += analogRead(pin);

return sum / count;

}

void drawPanel(String label, int x, int y, int w, int h, uint16_t color) {

tft.fillRect(x, y, w, h, TFT_DARKGREY);

tft.setTextColor(TFT_WHITE);

tft.setTextSize(2);

tft.setCursor(x + 5, y + 5);

tft.print(label);

tft.drawLine(x+5, y+h-25, x+w-5, y+h-25, TFT_WHITE);

}

void updateValue(int x, int y, int w, int h, int val, String unit, uint16_t color) {

tft.fillRect(x + 5, y + 25, w - 10, h - 50, TFT_BLACK);

tft.setTextColor(color);

tft.setTextSize(4);

tft.setCursor(x + 5, y + 40);

tft.print(val);

tft.setTextSize(2);

tft.print(" " + unit);

}

void updateValueFloat(int x, int y, int w, int h, float val, String unit, uint16_t color) {

tft.fillRect(x + 5, y + 25, w - 10, h - 50, TFT_BLACK);

tft.setTextColor(color);

tft.setTextSize(4);

tft.setCursor(x + 5, y + 40);

tft.print(val, 1);

tft.setTextSize(2);

tft.print(unit);

}

void updateValueWide(int x, int y, int w, int h, int val, String unit, uint16_t color) {

tft.fillRect(x + 5, y + 25, w - 10, h - 50, TFT_BLACK);

tft.setTextColor(color);

tft.setTextSize(6);

tft.setCursor(x + 10, y + 40);

tft.print(val);

tft.setTextSize(3);

tft.print(" " + unit);

// Bar Graph

int maxP = 1000; // Scale to 1000W

int barW = map(val, 0, maxP, 0, w - 20);

tft.fillRect(x + 10, y + h - 20, barW, 10, color);

tft.fillRect(x + 10 + barW, y + h - 20, (w - 20) - barW, 10, TFT_DARKGREY);

}

3. Calibration Steps

SWR Calibration:

Connect a dummy load (SWR=1.0). Transmit 100W. Adjust PWR_FACTOR until FWD reads 100W and SWR reads 1.0.

Introduce a known mismatch (if possible) or verify REF reads near 0.

Peak Hold Tuning:

Transmit SSB voice. Watch the PEP display. It should jump quickly and decay slowly (1.5s). Adjust DECAY_TIME in code to taste.

Touch Reset:

Tap the screen anytime to zero the Peak Hold reading.

4. Safety & RF Proofing

Shielding: Place the ESP32 in a metal enclosure. Run shielded cables for FWD/REF inputs.

Clamping: Add 3.3V Zener diodes on FWD and REF lines to protect the ESP32 from high-voltage spikes if the coupler fails.

Grounding: Ensure the coupler ground is solidly connected to the SB-200 chassis.

Heathkit SB-200 Complete modern parts list

2026-07-12T13:09:20.105-07:00

SB-200 Complete modern parts list

1. High-Voltage Power Supply Components

Component Original Spec Modern Replacement Spec Mouser Part Number DigiKey Part Number Notes

Filter Capacitors (C1–C6) 125 µF / 450 V 220–330 µF / 450 V, 105°C, Snap-in 598-331A450M2PH (Cornell Dubilier, 330µF) 493-15716-ND (Nichicon, 330µF) Upgrade to 330µF requires Soft-Start mod.

Bleeder Resistors (R1–R6) 30 kΩ / 10 W 82 kΩ / 3 W or 2x 220 kΩ / 1 W 283-82K-RC (Ohmite, 3W) 82KQT-ND (2x 220k in parallel) Reduces heat from 26W to <1W total.

Rectifier Diodes (D1–D16) Silicon Rectifier 1N4007 (1000V/1A) or BY225 (1300V/3A) 512-1N4007 (Vishay) 1N4007FSCT-ND (Diodes Inc.) Use 10x in series for safety margin.

Bias Filter Capacitor ~100 µF / 6.3 V 100 µF / 16 V Aluminum Electrolytic 667-ECA-1CM101 (Panasonic) P1235-ND (Panasonic) Low voltage rating is sufficient.

HV Fuse N/A (Add-on) 2 A / 10 kV Ceramic High-Speed 504-50200 (Littelfuse) F10200-ND Essential for tube flashover protection.

Inrush Thermistor N/A (Add-on) CL-90 or MS32 10002 (10Ω, 5A) 527-CL90 (Ametherm) 2156-CL90-ND Limits transformer/cap inrush current.

Glitch Resistor N/A (Add-on) 15 Ω / 10 W Wirewound (Non-inductive) 290-15-RC (Ohmite OX series) 15F10-ND Absorbs energy during catastrophic failure.

Heathkit SB-200 power supply capacitor kit

2. RF Deck & Tube Circuit Components

Component Original Spec Modern Replacement Spec Mouser Part Number DigiKey Part Number Notes

Plate Resistors (Suppressor) 47 Ω / 2 W Comp 56 Ω / 2 W Carbon Composition (Non-inductive) 290-56-RC (Ohmite OX series) 56F2-ND (Ohmite) Critical: Must be Carbon Comp, not film.

Suppressor Wire Bare Copper 12-14 AWG Teflon Insulated Stranded 566-1240-ND (Belden, 14 AWG) AE1014-ND Cut 115mm length for hairpin loop.

Grid Resistors (R21, R22) 33 Ω / 2 W 33 Ω / 2 W Carbon Composition 290-33-RC (Ohmite) 33F2-ND Match pair within 1% if possible.

Grid Capacitors (C14, C15) 200 pF / 500 V 200 pF / 500 V Silver Mica (5% tol) 594-CD15FD201JO3 (Cornell Dubilier) CD15FD201JO3-ND Keep leads <6mm. Some use 510pF.

Filament Choke (RFC2) ~10 µH 20–25 µH RF Choke (High Current) 546-190-253J15 (Miller) M8660-ND Ensure current rating > 3A.

Filament Bypass (C12) 0.02 µF 0.022 µF / 630 V Film (Polypropylene) 667-ECQ-U2223KL (Panasonic) EF1126CT-ND Non-inductive film type preferred.

Plate Choke (RFC3) ~50–100 µH 60 µH / 500 mA RF Choke (J-W type) 436-1025-ND (J.W. Miller) M4563-ND Verify DC resistance is low (<10Ω).

Anode Blocking Cap 500 pF / 5 kV 500 pF / 7.5 kV Ceramic Doorknob 80-C941U501KZS (Cornell Dubilier) 399-11455-ND High voltage rating is critical.

Tube Sockets Ceramic 4-Pin Ceramic 4-Pin (572B/811A compatible) 520-1423 (Amphenol) AE1023-ND Replace if original is plastic/cracked.

Tubes (V1, V2) 572B 572B (Shuguang/Taylor) N/A (RF Parts Co.) N/A (RF Parts Co.) Not typically stocked by Mouser/DigiKey.

Heathkit SB-200 parasitic suppressor assembly

3. Input Network & Metering Components

Component Original Spec Modern Replacement Spec Mouser Part Number DigiKey Part Number Notes

Input Caps (80m) ~510–910 pF 560 pF / 1 kV Silver Mica 594-CD15FD561JO3 CD15FD561JO3-ND Values vary; verify with analyzer.

Input Caps (40m) ~360–510 pF 390 pF / 1 kV Silver Mica 594-CD15FD391JO3 CD15FD391JO3-ND

Input Caps (20m) ~200–360 pF 220 pF / 1 kV Silver Mica 594-CD15FD221JO3 CD15FD221JO3-ND

Input Caps (15/10m) ~47–68 pF 56 pF / 1 kV Silver Mica 594-CD15FD560JO3 CD15FD560JO3-ND

HV Divider Resistors 4.7 MΩ (x3) 1.5 MΩ / 1 kV Metal Film (x10) 299-1.5M-RC (Vishay) 1.5MQBK-ND Series chain totals 15 MΩ.

Meter Protection Diodes N/A 1N5408 (1000V/3A) or 1N4148 512-1N5408 1N5408FSCT-ND Install inverse-parallel across meter.

Meter Shunt ~1 Ω 1 Ω / 5 W Wirewound (Precision) 283-1.0-RC (Ohmite) 1.0QT-ND Ensure low temperature coefficient.

Meter Lamp 6.3 V Incandescent 6.3 V LED with 1kΩ Resistor 512-LED63V (If avail) or Generic LED5MM-ND + 299-1K-RC Reduces heat inside cabinet.

4. Relays, Connectors, and Mechanical

Component Original Spec Modern Replacement Spec Mouser Part Number DigiKey Part Number Notes

Antenna Relay (K1) DPDT 120VAC Vacuum Relay or High Power RF Relay 546-JEN1212 (Jennings) JEN1212-ND Lower loss on 10m/6m than original.

Input/Output Connector Proprietary SO-239 (UHF) or BNC Bulkhead 523-142-0201-202 (Amphenol) ARF1055-ND Use Teflon insulated (RG-142) coax.

Cooling Fan Blower Motor 4" (120mm) 115/230V AC Muffin Fan 527-109P0412 (Sanyo Denki) 109P0412-ND Ensure ≥20 CFM airflow.

Line Cord 2-Wire Hardwired 3-Wire Grounded Cord + Strain Relief 536-701960000 (Schurter) WK2030-ND Mandatory for safety grounding.

Wire (Internal) Cloth Covered 22/18/12 AWG Teflon (Silver Plated) 566-1420-ND (Belden 1420) AE1020-ND High temp, low loss dielectric.

Coax (Internal) RG-58 RG-142 (Silver/Teflon) or RG-316 566-1421-ND (Belden 1421) AE1021-ND Same OD as RG-58, higher temp rating.

Implementation Notes

Resistor Types: For parasitic suppressors, carbon composition is mandatory. Metal film or wirewound resistors will act as inductors at VHF and fail to suppress oscillations.

Capacitor Voltage: Always exceed the working voltage by at least 20%. For the input network, 1 kV rating is recommended despite lower operating voltages to handle SWR spikes.

Sourcing Tubes: 572B tubes are not typically stocked by general distributors like Mouser/DigiKey. Purchase from specialized RF suppliers like RF Parts Company or Tube Depot.

Soft-Start Requirement: If upgrading filter capacitors above 220 µF, the inrush thermistor (CL-90) is mandatory to prevent tripping breakers and damaging the transformer.

6-tube 572B HF amplifier

2026-07-12T13:02:50.347-07:00

Building a 6-tube 572B HF amplifier is a significant homebrew project typically aimed at achieving 1.5 kW to 2 kW PEP output. While commercial amplifiers like the Ameritron AL-82 use four tubes (3-500Z or similar), a 6-tube 572B design is less common commercially but feasible for high-power linear operation.

Here are the critical design parameters and steps for constructing such an amplifier:

1. Core Design Specifications

Tube Configuration: Six 572B tubes in parallel in a grounded-grid configuration.

Operating Class: Class AB2 or B linear (SSB/CW).

Expected Output: Approximately 1800 W PEP (assuming ~300 W per tube under ideal conditions).

Plate Voltage: Typically 2000 V to 2500 V DC. The 572B is rated for 2750 V maximum, but 2200–2400 V is a common sweet spot for longevity and power.

Plate Current: Total static current ~270 mA (45 mA/tube); peak SSB current ~1.5 A (250 mA/tube).

Filament Supply: Critical. Requires 6.3 V AC at 24 Amps (4 A per tube × 6). Heavy gauge wiring and a massive transformer are essential to prevent voltage drop, which kills thoriated-tungsten filaments.

2. Key Circuit Blocks

Power Supply

You will need a heavy-duty high-voltage transformer capable of delivering ~2200–2500 V at 1.5–2 A (for SSB duty cycle).

Rectification: Full-wave bridge using high-voltage, high-current diodes (e.g., 10 kV, 1 A stacks) or vacuum tube rectifiers.

Filtering: A capacitor-input filter (approx. 20–40 µF at 3000 V rating) with equalizing resistors.

Bleeder Resistor: Essential for safety to discharge HV capacitors when off.

RF Deck and Tank Circuit

Input Matching: A broadband transformer or switched pi-network to match the low cathode impedance (~4–6 ohms for 6 tubes) to 50 ohms.

Output Tank: A standard Pi-network (or Pi-L for better harmonic suppression) capable of handling 2 kW.

Plate Capacitor: High voltage, high capacitance (e.g., 250–500 pF, 5000 V rating).

Load Capacitor: Similar rating, often higher capacitance (e.g., 500–1000 pF).

Plate Choke: Must be resonant-free across 160m–10m (or 6m). A "silver bullet" or wound choke designed for high power is required.

Band Switching: Heavy-duty ceramic wafer switches with short leads to minimize inductance.

Cooling and Mechanical

Airflow: The 572B relies on radiation and forced air. Six tubes generate immense heat (~1 kW dissipation at full load). You need a high-CFM blower (squirrel cage) pushing air from base to plate cap.

Chassis: Large, well-ventilated metal enclosure. Tubes are often mounted horizontally or vertically with ample spacing.

3. Construction Resources & Schematics

No single "off-the-shelf" schematic for a 6-tube 572B exists in standard manuals (most are 2 or 4 tubes). You must adapt existing designs:

Adaptation Strategy: Take a proven 2-tube schematic (like the Ameritron AL-572 or Heathkit SB-200) and scale the power supply, filament transformer, and tank components for 3x the tube count.

Reference Designs:

Look for "Homebrew 1.5 kW Linear Amplifier" plans in amateur radio journals (e.g., QST, CQ Amateur Radio).

The W8JI website (Tom Rauch) offers extensive technical articles on designing high-power amplifiers, including tank circuit calculations and stability tips.

Search for "4-572B amplifier schematic" (e.g., Ameritron AL-82 or Dentron GLA-1000B) and add a third pair of tubes, ensuring the filament and HV supplies are upgraded.

572B 6 tube amplifier schematic homebrew

building 572B tube amplifier

View all

⚠️ Safety Warning

This project involves lethal voltages (2500+ V) and high currents.

Always use safety interlocks on cabinet doors.

Install bleeder resistors on all HV capacitors.

Ensure proper grounding and use a fuse or circuit breaker rated for the input current.

Verify filament voltage under load; low filament voltage drastically shortens tube life.

Building a 6-tube 572B amplifier requires sourcing specialized high-voltage components that are rarely found in standard catalogs like Mouser or DigiKey. While those distributors excel at low-voltage electronics, the specific HV capacitors, plate chokes, and high-current filament transformers needed for a 2 kW RF amplifier are typically sourced from ham radio specialty suppliers (e.g., RF Parts, Max-Gain Systems, Surplus Sales) or industrial surplus.

Below is a comprehensive Parts List categorized by function, with estimated costs and primary sources based on current availability.

1. Active Components (Tubes & Rectifiers)

The core of the amplifier. You need six matched tubes to ensure even current distribution.

RF Power Triodes (572B): 6 units.

Specs: 572B/T160L, Thoriated Tungsten Filament.

Source: RF Parts, DX Engineering, Ham Radio Outlet.

Cost: ~$40–$125 each. Matched sets (pairs/trios) are preferred.

Est. Total: $300 – $750.

HV Rectifier Diodes: 4–8 units (for full-wave bridge).

Specs: 10 kV PIV, 1 A (e.g., HVM10, 10A10 stacks).

Source: Island Amplifier, eBay, Surplus Sales.

Note: Often bought as pre-built stacks or kits to avoid series balancing issues.

Est. Total: $20 – $60.

572B vacuum tube matched set

2. Power Supply Components

This is the heaviest and most expensive section. Mouser/DigiKey can supply some capacitors, but the transformer is custom/surplus.

HV Power Transformer: 1 unit.

Specs: 2200–2500 VAC secondary @ 1.5–2 A (SSB duty). ~5–6 kVA.

Source: Surplus (e.g., Surplus Sales of Nebraska), eBay, or custom wind from Antek (multiple units in series/parallel, though difficult for HV).

Cost: $300 – $800 (New custom units exceed $1,500).

HV Filter Capacitors: 2–4 units.

Specs: 20–40 µF total, 3000–4000 VDC rating. (Often wired in series/parallel).

Source: Alibaba (for new electrolytics like Pluspark), eBay (surplus Cornell Dubilier), Mouser (limited selection >2kV).

Cost: $50 – $150.

HV Bleeder Resistor: 1 assembly.

Specs: ~220kΩ – 470kΩ, 50–100 W total rating.

Source: Mouser/DigiKey (series chain of 5W metal oxide resistors).

Cost: $20.

3. RF Deck & Tank Circuit

Critical for tuning and efficiency. Variable capacitors must be vacuum types to handle high voltage without arcing.

Plate Capacitor (Variable): 1 unit.

Specs: 5–500 pF, 5000–7000 V peak.

Source: Max-Gain Systems, Surplus Sales. (e.g., Comet CV05C-500 series).

Cost: $350 – $500.

Load Capacitor (Variable): 1 unit.

Specs: 10–1000 pF, 3000–5000 V peak.

Source: Max-Gain Systems, eBay.

Cost: $200 – $400.

Plate Choke (RFC): 1 unit.

Specs: 2.5 mH – 5 mH, self-resonant free across 160m–10m, 2 kW rating.

Source: RF Parts (e.g., RFC-245 or equivalent "Silver Bullet" chokes).

Cost: $80 – $150.

Band Switch: 1 unit.

Specs: 6-position, ceramic wafer, high voltage rating (e.g., Centralab 2000 series).

Source: Surplus, eBay.

Cost: $40 – $80.

Input Matching Transformer: 1 unit.

Specs: Broadband 1:9 or 1:16 impedance ratio (cathode to 50Ω).

Source: Homebrew (FT-240-43 cores) or RF Parts.

Cost: $20 – $50.

high voltage vacuum variable capacitor 500pF 5kV

4. Filament & Control Supply

The 6.3V supply is critical; voltage drop kills tubes.

Filament Transformer: 1 unit (or parallel combination).

Specs: 6.3 VAC @ 24–30 Amps (4A/tube × 6 + margin). ~150–200 VA.

Source: Antek Products (e.g., AN-0206M is only 2A; you need massive custom or parallel industrial transformers), Surplus.

Note: Standard audio transformers (Hammond) are too small (max ~5A). You may need to parallel three 10A units or find a large industrial control transformer.

Cost: $100 – $300.

Relays: 2–3 units.

Specs: HV vacuum relay (plate), 12V/24V DPDT (antenna bypass).

Source: Mouser (Jennings, Kilovac), DX Engineering.

Cost: $50 – $100.

Cooling Fan: 1–2 units.

Specs: High CFM (100+), low static pressure, 120VAC.

Source: Mouser/DigiKey (e.g., NMB, Papst), DigiKey.

Cost: $30 – $60.

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5. Chassis & Hardware

Chassis: Large aluminum or steel enclosure (e.g., 19" rack mount or custom sheet metal).

Source: Hammond Manufacturing, eBay.

Cost: $100 – $200.

Tube Sockets: 6 units.

Specs: Ceramic 4-pin (for 572B).

Source: RF Parts, Antique Electronic Supply.

Cost: $15 – $25 each ($90–$150 total).

Meters: 2 units.

Specs: HV DC (0–3000V) and Plate Current (0–1A or 0–2A).

Source: eBay, Fair Radio Sales.

Cost: $40 – $80.

Estimated Total Cost

Category Estimated Cost (USD) Notes

Tubes (6x) $300 – $750 Major variable; matched sets cost more.

HV Transformer $300 – $800 Best found surplus; new is expensive.

Tank Components $600 – $1,000 Vacuum capacitors are costly.

Filament Supply $100 – $300 Requires massive current capacity.

Chassis/Misc $200 – $400 Sockets, fans, meters, hardware.

Total $1,500 – $3,250 Excludes tools and test equipment.

Critical Sourcing Notes

Mouser/DigiKey Limitations: Use these for resistors, low-voltage capacitors, relays, fans, and hardware. They generally do not stock 3000V+ electrolytics, 500pF 5kV variable capacitors, or 25A filament transformers.

Surplus is Key: For the HV transformer and vacuum capacitors, check Max-Gain Systems, Surplus Sales of Nebraska, and eBay.

Safety: Ensure all HV components have adequate creepage/clearance distances. Use safety interlocks on the chassis.

building 2kW tube amplifier power supply

make calculations based on available sources for all of the required parts values

Based on the 572B datasheet specifications and standard high-power amplifier design equations, here are the calculated component values for a 6-tube grounded-grid amplifier operating at 2200–2400 V.

1. Operating Parameters & Load Impedance

To achieve ~1.5–1.8 kW PEP output, the tubes are operated in parallel.

Plate Voltage ($E_p$): 2200 V (Conservative rating for longevity; max is 2750 V).

Plate Current ($I_p$): 250 mA/tube $\times$ 6 = 1.5 A (Peak SSB).

Idle Current: ~45 mA/tube $\times$ 6 = 270 mA.

Input Power: $2200 \text{ V} \times 1.5 \text{ A} \approx 3300 \text{ W}$ (PEP input).

Estimated Output Power: ~1800–2000 W PEP (assuming 60% efficiency).

Plate Load Impedance ($Z_p$): Using the formula $Z_p \approx \frac{E_p}{1.8 \times I_{max}}$ (Class AB2/B approximation) or derived from typical single-tube load ($\approx 3500,\Omega$) divided by parallel count: Z

≈

3500Ω

≈580Ω to 600Ω Design Target: 600 $\Omega$ effective plate load.

2. Filament Supply Calculations (Critical)

The 572B requires strict voltage regulation to prevent emission loss or filament burnout.

Voltage: 6.3 V AC (Tolerance $\pm$0.3 V).

Current: 4.0 A/tube $\times$ 6 = 24.0 A minimum.

Transformer Rating: Must be 6.3 V @ 30 A (20% headroom recommended).

Power: $6.3 \text{ V} \times 30 \text{ A} = \mathbf{189 \text{ VA}}$ (Minimum).

Wiring Voltage Drop: To keep drop $< 0.15 \text{ V}$ (to maintain 6.15 V at socket): R

max

24 A

0.15 V

=0.00625Ω Implementation: Use extremely short, heavy-gauge wiring (e.g., 00 AWG or multiple parallel 4 AWG strands) directly from transformer to a central bus bar, then to sockets.

30 amp 6.3 volt filament transformer

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3. Pi-Network Tank Circuit Values

Calculated for transforming 600 $\Omega$ (Plate) to 50 $\Omega$ (Antenna) with a loaded Q = 12. Values vary by band; extremes (160m and 10m) dictate component ratings.

Band Freq (MHz) Plate Cap ($C_p$) Load Cap ($C_L$) Inductance ($L$)

160m 1.8 480 pF 1650 pF 58 µH

80m 3.8 220 pF 850 pF 26 µH

40m 7.1 115 pF 480 pF 13 µH

20m 14.2 55 pF 260 pF 6.5 µH

10m 28.5 25 pF 140 pF 3.2 µH

Component Specifications Required:

Plate Capacitor ($C_p$): Must cover 25–500 pF.

Voltage Rating: $E_{peak} \approx E_{DC} \times 3.5$ (high Q/swing).

$2200 \text{ V} \times 3.5 \approx \mathbf{7700 \text{ V}}$.

Select: 500 pF, 8–10 kV Vacuum Variable.

Load Capacitor ($C_L$): Must cover 150–1700 pF.

Voltage Rating: Lower than plate, but RF current is massive.

Select: 2000 pF, 3–5 kV Vacuum Variable (High Current type).

Inductor ($L$):

Must handle 1.5 A RF current without saturation or overheating.

Use a "Silver Bullet" style air-core choke or switched tapped coil. Total inductance range 3 µH to 60 µH.

4. Plate Choke (RFC) Design

The choke must present high impedance across 1.8–30 MHz without self-resonance.

Inductance: 2.5 mH to 3.0 mH.

Impedance Check: At 1.8 MHz, $X_L = 2\pi f L \approx 34,000,\Omega$ (Safe, $>10\times Z_p$).

Construction:

Universal Winding (pie-wound) or Random Wound on a ceramic form to minimize inter-turn capacitance.

Length: ~6–8 inches, diameter ~2 inches.

Wire Size: #18 to #20 AWG enamel (handles DC + RF skin effect).

Warning: Do not use a single-layer solenoid; it will self-resonate within the HF band, causing arcs and failure.

5. Input Matching Network

Grounded grid input impedance is very low ($Z_{in} \approx \frac{1}{g_m} + \frac{Z_{load}}{\mu}$).

Single Tube $Z_{in}$: $\approx 50,\Omega$ / $\mu$ (approx 170) $\rightarrow$ Very low, typically 4–6 $\Omega$ cathode impedance.

6 Tubes in Parallel: $Z_{in_total} \approx \frac{5,\Omega}{6} \approx \mathbf{0.8,\Omega}$.

Matching Strategy:

Use a Broadband Transformer (Ferrite cores, e.g., FT-240-43 or T-200-2).

Turns Ratio: To match $0.8,\Omega$ to $50,\Omega$ ($R_{ratio} = 62.5$).

Impedance ratio $\approx 64:1$ $\rightarrow$ Turns ratio $\sqrt{64} = \mathbf{8:1}$.

Example: 2 turns (Primary) to 16 turns (Secondary) on multiple large ferrite cores stacked for power handling.

6. Power Supply Filter & Bleeder

Filter Capacitance:

Rule of thumb: 1–2 µF per 100 mA.

$1.5 \text{ A} \times 2 \text{ µF/A} \approx \mathbf{30 \text{ µF}}$ total.

Configuration: Two 60 µF, 3000 V capacitors in series (Result: 30 µF, 6000 V rating). Use 100 kΩ equalizing resistors across each.

Bleeder Resistor:

Target discharge time < 1 minute. $R \le \frac{t}{5C}$.

$R \approx \mathbf{220 \text{ k}\Omega}$, 100 W rating (dissipates $\approx 24 \text{ W}$ continuously at 2200 V).

Construction: Series chain of twenty 11 kΩ, 5 W metal oxide resistors.

pi-network tank circuit calculations tube amplifier

6 tube 572B amplifier component values site:reddit.com

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Summary of Critical Parts List (Calculated)

Component Calculated Value Minimum Rating Required

HV Transformer 2200–2400 VAC 2.0 A (SSB), 5 kVA

Filament Transformer 6.3 VAC 30 A (190 VA)

Plate Capacitor 25–500 pF 8 kV Peak

Load Capacitor 150–2000 pF 3 kV, High RF Current

Plate Choke 2.5 mH Self-resonant > 30 MHz

HV Filter Caps 30 µF Total 3000 VDC (x2 in series)

Input Transformer 8:1 Turns Ratio #12 AWG wire, FT-240-43 cores

6-Tube 572B HF Amplifier Schematic Description

1. Input Circuit (Cathode Drive)

Topology: Broadband Transformer Match (No tuning required per band).

Connections:

RF Input (SO-239): Center pin connects to the primary of T1.

T1 (Input Transformer):

Core: Stack of 3x FT-240-43 or FT-240-61 ferrite toroids.

Primary: 2 turns of #12 AWG Teflon wire.

Secondary: 16 turns of #12 AWG Teflon wire (Bifilar or Trifilar wound for coupling).

Center Tap: The secondary center tap connects to Ground (via RF bypass caps).

Ends: Connect to the Cathode Bus (Pins 2 & 3 of all 6 sockets).

Filament Feed:

6.3V AC connects to the cathode bus via two 2.5 mH RF Chokes (one on each side of the transformer secondary) to isolate RF from the filament transformer.

Bypass: 0.1 µF, 100V ceramic caps from each side of the filament supply to chassis ground.

Grid Grounding:

Grid pins (Pin 1) of all 6 sockets connect directly to a low-inductance ground plane (copper strap) surrounding the socket cluster.

Optional Stability: 47–100 Ω, 2W non-inductive resistors from Grid (Pin 1) to Ground on each tube socket (very short leads).

2. RF Deck (Tube Cluster & Plate Circuit)

Tubes (V1–V6): Six 572B sockets wired in parallel.

Pin 1 (Grid): Grounded directly to chassis.

Pin 2 & 3 (Cathode/Filament): Connected to the Input Transformer secondary and Filament supply.

Pin 4 (Getter/Cathode): Connected to Cathode bus.

Plate Cap: All 6 tube plate caps connected together via a wide copper strap or bus bar to the Plate Tank Capacitor ($C_{plate}$) and Plate Choke.

Plate Choke (RFC):

Connected between the Plate Bus and the HV B+ supply point.

Value: 2.5 mH, universal wound, self-resonant > 30 MHz.

Tank Circuit (Pi-Network):

$C_{plate}$ (Tuning): Vacuum variable capacitor (5–500 pF, 8 kV). One side to Plate Bus, other side to Ground.

$L_{tank}$ (Inductor): Switched air-core inductor (3 µH to 60 µH).

Connected between Plate Bus and the junction of $C_{plate}$ and $C_{load}$.

Tapped for bands (160m, 80m, 40m, 20m, 15m, 10m) via a ceramic wafer switch.

$C_{load}$ (Loading): Vacuum variable capacitor (150–2000 pF, 3–5 kV).

Connected between the Tank Inductor output and Ground.

Output Link:

A 2–3 turn link coil wound over the cold end of $L_{tank}$ (near ground).

Connected to the Antenna Relay and Output SO-239.

RF Choke: 2.5 mH choke from the output line to ground (DC block protection).

3. High Voltage Power Supply

Transformer (T2): 2200–2400 VAC secondary, center-tapped (or dual secondary).

Rectifier: Full-wave bridge (4x 10 kV, 1 A diodes).

AC inputs to transformer secondary.

DC+ output to Filter Capacitor.

DC- to Chassis Ground.

Filter:

Two 60 µF, 3000 V electrolytics in series (Total 30 µF, 6000 V).

Equalizing Resistors: 100 kΩ, 5W across each capacitor.

Bleeder Resistor: 220 kΩ, 100W total (series chain of 5W resistors) across the entire HV+ and Ground.

HV Feed: HV+ connects to the center tap of the Plate Choke (via a HV RF choke if necessary for decoupling).

Metering:

HV Meter: High-resistance divider (e.g., 100 MΩ total) from HV+ to a 1 mA meter movement.

Plate Current Meter: Shunt resistor (0.1 Ω, 10W) in the Ground return of the HV supply (common cathode current measurement).

4. Control & Safety Circuit

Relay Control:

12V/24V DC from transceiver key-line drives a DPDT relay.

Relay 1 (Antenna): Switches antenna between Transceiver (Bypass) and Amplifier Output.

Relay 2 (HV): Enables HV contactor (safety interlock in series).

Interlocks: Microswitches on all cabinet doors wired in series with the HV contactor coil.

Fans: Thermostatically controlled blowers (intake at bottom, exhaust at top near plate caps).

Text-Based Schematic (Netlist Representation)

[RF INPUT] --(SO-239 Center)--> T1_PRIMARY_PIN1

T1_PRIMARY_PIN2 --(SO-239 Shell)--> GND

[T1: INPUT XFMR] (FT-240-43 x3)

Primary: 2 Turns (#12 AWG)

Secondary: 16 Turns (#12 AWG), Center-Tapped

T1_SEC_CT --> GND

T1_SEC_END_A --> RFC_FIL_A --> CATHODE_BUS

T1_SEC_END_B --> RFC_FIL_B --> CATHODE_BUS

RFC_FIL_A/B: 2.5mH RF Choke, 5A rating

[FILAMENT SUPPLY]

6.3V_AC_1 --> RFC_FIL_A

6.3V_AC_2 --> RFC_FIL_B

C_BYPASS: 0.1uF 100V from each 6.3V line to GND

[TUBES V1-V6] (572B)

Pin 1 (Grid) --> GND_PLANE (Shortest path)

Pin 2 (Cathode) --> CATHODE_BUS

Pin 3 (Filament) --> CATHODE_BUS

Pin 4 (Cathode) --> CATHODE_BUS

Plate Cap --> PLATE_BUS (Wide copper strap connecting all 6)

[PLATE TANK]

PLATE_BUS --> C_PLATE (5-500pF, 8kV Vacuum Var) --> GND

PLATE_BUS --> L_TANK (Switched Inductor 3-60uH) --> JUNCTION_X

JUNCTION_X --> C_LOAD (150-2000pF, 5kV Vacuum Var) --> GND

JUNCTION_X --> LINK_COIL (2 Turns) --> ANTENNA_RELAY_COMMON

[OUTPUT]

ANTENNA_RELAY_NO --> AMP_OUTPUT (SO-239)

ANTENNA_RELAY_NC --> BYPASS_INPUT (Relay input from exciter)

RF_CHKE_PROT: 2.5mH from ANTENNA_RELAY_COMMON to GND

[PLATE CHOKE & HV]

PLATE_BUS --> RFC_PLATE (2.5mH, 2kV rating) --> HV_B+

HV_B+ --> FILTER_CAPS (+)

FILTER_CAPS (-) --> GND (Via Current Shunt)

[HV POWER SUPPLY]

HV_XFMR_SEC (2200V-0-2200V)

End_A --> DIODE_A (Anode)

End_B --> DIODE_B (Anode)

Center_Tap --> GND

DIODE_A (Cathode) & DIODE_B (Cathode) --> HV_B+

(Alternative: Full Bridge if no center tap)

FILTER_CAPS: 2x 60uF 3000V in Series

Midpoint --> GND (Virtual center)

Equalizing Resistors: 100k 5W across each cap

Bleeder Resistor: 220k 100W across total HV_B+ to GND

[METERING]

Plate Current Shunt: 0.1 Ohm 10W in HV Return path to GND

Meter Movement across Shunt (with range switch)

HV Divider: 100M Ohm chain from HV_B+ to 1mA Meter to GND

[CONTROL]

Key_Line_In --> Relay_Coil --> GND

Relay_Contacts: Switches HV Contactor and Antenna Path

Safety_Interlock_Switches (Door) in series with HV Contactor Coil

Construction Tips for the Schematic

Grounding: The "Ground" for the grids (Pin 1) must be a solid copper plane or heavy strap. Do not daisy-chain grid grounds.

Plate Bus: Use a wide copper strap (1–2 inches wide) to connect the six plate caps to the tank circuit to minimize inductance and ensure equal current distribution.

Filament Wiring: The voltage drop from the transformer to the sockets must be measured under load. If it drops below 6.0 V, the tubes will be damaged. Use bus bars rather than wires for the final distribution to the sockets.

Choke Placement: The Plate Choke (RFC) should be mounted away from the tank inductor to prevent coupling, which causes parasitic oscillations.

================================================================================

6-TUBE 572B GROUNDED-GRID HF LINEAR AMPLIFIER

COMPLETE ASCII SCHEMATIC DIAGRAM

================================================================================

SECTION 1: RF INPUT & CATHODE DRIVE NETWORK (BROADBAND)

--------------------------------------------------------------------------------

+------------------+

| FILAMENT XFMR |

| 6.3V @ 30A |

| (T_FIL) |

+--------+---------+

| 6.3V AC

+-------------------------+-------------------------+

| | |

[RFC_F1] [RFC_F2] |

2.5mH/5A 2.5mH/5A |

| | |

+-------------+-------------+-----------+-------------+-----------+

| | | | | |

=== C_B1 === C_B2 | | | |

0.1uF/100V 0.1uF/100V | | | |

| | | | | |

GND GND | | | |

| | | |

[RF INPUT] | | | |

SO-239 (J1) | | | |

| | | | |

+----[C_IN]----+ | | | |

| 100pF | | | | |

| | | | | |

| | | | | |

GND | | | | |

| | | | |

| | | | | |

| | | | | |

+--------------+----------+ | | |

| | | |

(To Cathode Bus via RFC_F1) (To Cathode Bus via RFC_F2)

SECTION 2: TUBE CLUSTER (V1 - V6) & GRID GROUNDING

--------------------------------------------------------------------------------

CATHODE BUS (Heavy Copper Strap, Low Inductance)

=========================================================================

| | | | | |

+---+ +---+ +---+ +---+ +---+ +---+

|V1 | |V2 | |V3 | |V4 | |V5 | |V6 | (572B Tubes)

|572| |572| |572| |572| |572| |572|

+---+ +---+ +---+ +---+ +---+ +---+

| | | | | | | | | | | |

Pin 1| |Pin 2| |Pin 3| |Pin 4| | | | | |

(G) | | (K) | | (F) | | (K) | | | | | |

| | | | | | | | | | | |

| +-----+ +-----+ +-----+-----+-----+-----+-----> TO CATHODE BUS

+----[R_G1]----+

| 100Ω 2W | (Optional Stability Resistor per tube)

| |

GND_PLANE (Solid Copper Flashing under sockets)

NOTE: All Pin 1 (Grid) connections must go directly to GND_PLANE with shortest leads.

SECTION 3: PLATE TANK CIRCUIT (PI-NETWORK) & OUTPUT

--------------------------------------------------------------------------------

PLATE BUS (Wide Copper Strap)

=============================

| | | | | |

(Plate Caps of V1-V6 connected here)

+-----------------------+

| |

[RFC_PLATE] [C_PLATE]

2.5mH Plate Choke (Tuning Capacitor)

(Self-Resonant >30MHz) Vacuum Variable

| 5-500pF / 8kV

| |

HV_B+ GND

[L_TANK] (Switched Air-Core Inductor)

+-------[BAND SWITCH]-------+

| (Taps: 160, 80, 40, 20, 15, 10m)

+-------------+

[C_LOAD]

(Loading Capacitor)

Vacuum Variable 150-2000pF

[L_LINK]

(2-Turn Link Coil)

+----[RFC_PROT]---- GND

| 2.5mH Choke

[K1_RELAY] (Antenna Changeover)

(DPDT, High Power)

+-----> TO ANTENNA (J_OUT)

FROM EXCITER (BYPASS) -----------------------+

SECTION 4: HIGH VOLTAGE POWER SUPPLY (HV PSU)

--------------------------------------------------------------------------------

AC MAINS (220/240V)

[FUSE] (15A Slow Blow)

[SWITCH] (Interlocked with Cabinet Doors)

+----+----+

| |

[T_HV] [T_CTRL] (Control Transformer for Relays/Fans)

2200V-0-2200V 12V/24V AC

| |

| [BRIDGE_CTRL]

| |

| DC Out --> Relay Coils / Fan Thermostat

[RECTIFIER_BRIDGE] (4x 10kV/1A Diode Strings)

+----[F_HV]----+ (1A HV Fuse)

| |

| [C_FILTER_1]

| 60uF / 3000V

| |

| [R_EQ1] (100k 5W Equalizing Resistor)

| |

| [C_FILTER_2]

| 60uF / 3000V

| |

| [R_EQ2] (100k 5W Equalizing Resistor)

| |

| GND (Center Point)

+----[R_BLEED]----+ (220kΩ Total / 100W Chain)

| |

HV_B+ GND

+----[METER_HV]----+ (High Resistance Divider 100MΩ -> 1mA Meter)

+----[RFC_PLATE]---+ (Goes to Plate Tank in Section 3)

CURRENT SHUNT (Plate Current Metering)

Located in HV Return Path (GND Side of Capacitors):

HV_GND ----[R_SHUNT 0.1Ω 10W]---- CHASSIS_GND

[METER_Ip] (0-2A DC Scale)

SECTION 5: CONTROL & SAFETY INTERLOCKS

--------------------------------------------------------------------------------

KEY LINE FROM EXCITER (12V/24V DC)

[SW_MANUAL] (Operate/Standby)

[INTERLOCK_1] (Front Door)

[INTERLOCK_2] (Rear Door)

[RELAY_COIL_K1] (Antenna Relay)

[RELAY_COIL_HV] (HV Contactor)

GND

FAN CONTROL:

[THERMOSTAT] (Set ~40°C) ---- [FAN_1] (Intake)

---- [FAN_2] (Exhaust)

Airflow: Bottom (Filament) --> Top (Plate Caps)

================================================================================

END OF SCHEMATIC

================================================================================

Legend & Symbols Used

[ ] : Component enclosure (Resistor, Capacitor, Transformer, Relay).

=== : Capacitor (Non-polarized or Vacuum Variable).

--- : Wire / Connection.

GND : Chassis Ground / Earth.

| : Vertical connection.

+ : Junction point (3 or more wires).

( ) : Descriptive text or value.

CT : Center Tap.

RFC : Radio Frequency Choke.

K1 : Relay designation.

V1-V6 : Tube designations.

Critical Construction Notes from Schematic

Cathode Bus: The connection from T1_SECONDARY to the tube cathodes (Pins 2, 3, 4) must be extremely low impedance. Use a solid copper bar, not wire.

Grid Grounding: Pin 1 of every tube must connect to the GND_PLANE immediately. Do not run long wires to ground.

HV Safety: The R_BLEED resistor chain is mandatory to discharge C_FILTER capacitors within 60 seconds of power-off.

Filament Chokes: RFC_F1 and RFC_F2 prevent RF from entering the filament transformer while allowing 6.3V AC to pass.

Plate Choke: RFC_PLATE must be mounted physically distant from L_TANK to prevent coupling and parasitic oscillation.

1. Active Components (Tubes & Rectifiers)

Qty Item Part Number / Specification Source / Notes

6 RF Power Triode 572B (or T160L) RF Parts, DX Engineering. Buy a matched set of 6. Do not mix batches.

4 HV Rectifier Diode 10 kV, 1 A (e.g., HVM10, 10A10) Island Amplifier, eBay. Buy as a pre-built Bridge Stack if possible to avoid balancing resistors.

1 HV Fuse 1 A, 10 kV (Slow Blow) Mouser (Littelfuse). Essential for protecting the transformer.

2. Transformers (Power Supply)

Qty Item Part Number / Specification Source / Notes

1 HV Plate Transformer Custom / SurplusSpec: 2200–2500 VAC center-tapped @ 1.5–2.0 A (3–4 kVA).Alternative: Two Hammond 373EX (650V @ 345mA) in series/parallel is insufficient current. Look for Hammond 374EX (850V @ 475mA) x3 or a single surplus Henry/Alpha unit. eBay, Surplus Sales. Search: "2500V 2A transformer". A Variac (10A) is highly recommended to adjust input voltage.

2–3 Filament Transformer Antek AN-5006 (50VA, 6V) OR Custom 6.3V @ 15A.Spec Total: 6.3 VAC @ 30A minimum.Strategy: Wire three 6.3V @ 12A units in parallel. Antek Products, eBay. Single 30A units are rare. Verify phasing before paralleling.

1 Control Transformer Hammond 166J6 (6.3V @ 6A) or similar.Spec: 12–24 VAC @ 2A for relays/fans. Mouser, DigiKey.

3. RF Tank Circuit (High Voltage)

Qty Item Part Number / Specification Source / Notes

1 Plate Capacitor (Tuning) Comet CV05C-500XIH/5 (Used) or Jennings CVDD-500-10S.Spec: 5–500 pF, 5–10 kV Peak. RF Parts, Max-Gain Systems. Must be Vacuum Variable.

1 Load Capacitor Jennings CVDD-500-7.5S or similar.Spec: 150–2000 pF, 3–5 kV Peak, High RF Current. Surplus Sales, eBay. Often requires two in parallel if single unit not found.

1 Plate Choke (RFC) Custom Wound or RFI-2500 equivalent.Spec: 2.5 mH, Universal/Pie wound, Self-Resonant > 30 MHz. RF Parts (Silver Bullet style). Do not use single-layer solenoid.

1 Band Switch Centralab PA215 series (Surplus) or Ceramic Wafer Switch.Spec: 6 Position, Ceramic, High Voltage standoff. eBay, Surplus. Search: "Centralab 2000 series" or "Ceramic wafer switch 6 position".

1 Tank Inductor Custom Air-Core.Spec: 3 µH to 60 µH (Tapped). Wire: #10 or #12 AWG bare copper. Homebrew. Wind on a 3–4 inch diameter form.

1 Input Transformer Custom.Spec: 2:16 Turns (1:8 ratio) on 3x FT-240-43 cores. Wire: #12 AWG Teflon. Mouser (Ferrite Cores), Homebrew.

4. Capacitors & Resistors (Power Supply)

Qty Item Part Number / Specification Source / Notes

2 HV Electrolytic Cornell Dubilier 36DX series or similar.Spec: 60 µF, 3000–3500 VDC. (Wired in series for 30µF/6000V). eBay, Alibaba. Mouser rarely stocks >2kV electrolytics.

2 Equalizing Resistor Mouser 594-100K5W (Ohmite/Oxide).Spec: 100 kΩ, 5 W (Metal Oxide). Mouser, DigiKey. One across each HV cap.

20 Bleeder Resistor Mouser 283-11K-RC.Spec: 11 kΩ, 5 W (Metal Oxide). (20 in series = 220kΩ, 100W). Mouser, DigiKey.

6 Grid Stopper (Optional) Mouser 291-100-RC.Spec: 100 Ω, 2 W (Carbon Comp/Non-inductive). Mouser. Install at each tube socket pin 1.

4 Filament Bypass Cap Mouser 80-C320C104M5U5TA.Spec: 0.1 µF, 100 V (Ceramic Disc). Mouser, DigiKey.

5. Mechanical & Hardware

Qty Item Part Number / Specification Source / Notes

6 Tube Socket New Sensor SK4A or U4A.Spec: 4-Pin Ceramic, Heavy Duty (for 572B/811A). RF Parts, Antique Electronic Supply.

1 Chassis Custom Sheet Metal or Hammond 1455 series (Large).Spec: Min 19" W x 10" H x 20" D. Aluminum or Steel. Hammond Mfg, Local Shop. Must support ~60 lbs.

2 Cooling Fan Mouser 517-3206KL-04W-B39 (NMB) or similar.Spec: 120 VAC, 100+ CFM, Low Noise. Mouser, DigiKey.

1 Thermostat Mouser 658-KST010B100.Spec: Close @ 40°C, Open @ 60°C. Mouser.

2 RF Choke (Filament) Custom or Mouser 546-RF1555-10.Spec: 2.5 mH, 5 A (Must not saturate). Homebrew on ferrite rod often required for high current.

1 Current Shunt Mouser 534-100-1.0.Spec: 0.1 Ω, 10 W (Wirewound). Mouser. For plate current metering.

2 Panel Meters Surplus or DigiKey.Spec: 0–3000 VDC (with divider) and 0–2 A DC. eBay, Fair Radio.

1 Antenna Relay Jennings RCV-1A-12 (Vacuum) or Kilovac.Spec: SPDT, 3 kV, 10 A. Mouser, DX Engineering.

1 HV Contactor Kilovac HC1 series.Spec: SPST-NO, 5 kV, 1 A. Mouser, eBay.

6. Connectors & Misc

Qty Item Part Number / Specification Source / Notes

2 SO-239 Connector Mouser 523-1251 (Amphenol).Spec: UHF Female. Mouser, DigiKey.

1 Power Entry Mouser 538-478-101-1132.Spec: IEC C14 with Fuse Holder (15A). Mouser.

1 Variac (Recommended) Staco 3PN136 or similar.Spec: 0–140 VAC, 10 A. eBay, Surplus. Critical for tuning HV.

Sourcing Strategy Summary

Mouser/DigiKey: Use for resistors, low-voltage caps, fans, connectors, hardware, and relays.

RF Parts / Max-Gain: Use for tubes, sockets, vacuum capacitors, and plate chokes.

eBay / Surplus: Use for HV Transformer, Filament Transformers, Band Switch, and Chassis.

Homebrew: Input Transformer, Tank Inductor, Filament RF Chokes.

/////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////////

1. RF Signal Path (The Amplified Signal)

The RF path follows a Grounded-Grid topology, where the input signal is applied to the cathode, the grid is grounded, and the amplified signal is taken from the plate.

Input Stage:

RF energy from the exciter enters via the Input SO-239.

It passes through a 100 pF DC blocking capacitor into the primary (2 turns) of the input broadband transformer (T1).

T1 (wound on three stacked FT-240-43 cores) steps up the voltage and transforms the impedance by a ratio of roughly 64:1 (8:1 turns ratio).

The secondary (16 turns) drives the Cathode Bus. The center tap of this secondary is grounded, providing a balanced drive to the parallel cathodes of all six tubes.

Tube Interaction (Amplification):

The RF voltage on the Cathodes (Pins 2, 3, 4) modulates the electron flow from cathode to plate.

The Grids (Pin 1) are held at RF Ground via a solid copper plane, acting as a shield between input and output.

Current flows through the tubes in phase with the input voltage (Grounded-Grid action), creating a high-power RF current at the Plate Caps.

Tank Circuit (Impedance Matching & Tuning):

RF current from the six plates combines on the Plate Bus and enters the Pi-Network.

Plate Capacitor ($C_{plate}$): Shunts excess RF current to ground, determining the resonant frequency and plate loading.

Tank Inductor ($L_{tank}$): Stores energy and, via the Band Switch, selects the appropriate inductance (3–60 µH) for the operating band.

Load Capacitor ($C_{load}$): Couples the tuned circuit to the output, matching the high plate impedance (~600 Ω) down to 50 Ω.

Output Stage:

The matched 50 Ω signal is picked up by a 2-turn link coil wound over the ground end of $L_{tank}$.

It passes through a protection choke and into the Antenna Relay (K1).

Transmit Mode: The relay connects the amplifier output to the Output SO-239 (Antenna).

Receive/Bypass Mode: The relay disconnects the amplifier and connects the exciter directly to the antenna.

grounded grid amplifier operation animation

2. Electrical Power Paths (DC & AC)

A. Filament Circuit (6.3V AC)

Source: The Filament Transformer (or parallel bank) provides 6.3 VAC @ 30A.

Isolation: The AC lines pass through two 2.5 mH RF Chokes ($RFC_{F1}, RFC_{F2}$). These chokes present high impedance to RF, preventing the RF signal from shorting out through the transformer windings, while allowing 60Hz AC to pass freely.

Bypass: 0.1 µF ceramic capacitors shunt any residual RF on the filament lines to ground.

Distribution: Power is delivered via heavy copper bus bars to Pins 2 and 3 of all six sockets. The voltage drop is kept under 0.15V to ensure tube longevity.

B. High Voltage Circuit (DC B+)

Generation:

Mains AC enters the HV Transformer, stepping up to ~2200–2500 VAC.

A Full-Wave Bridge Rectifier (4x 10kV diodes) converts AC to pulsating DC.

Filter Capacitors (two 60µF/3000V in series) smooth the ripple to pure DC.

A Bleeder Resistor (220kΩ) continuously drains charge for safety.

Distribution:

DC High Voltage flows through a 1A HV Fuse and the HV Contactor (safety switch).

It passes through the Plate Current Shunt (0.1Ω), where the voltage drop is measured by the Plate Current Meter.

The HV line connects to the center of the Plate Choke (RFC).

RF Isolation: The Plate Choke (2.5 mH) presents a high impedance to RF (blocking it from entering the power supply) while allowing DC to pass freely to the Plate Bus and tube plates.

C. Control Circuit (Low Voltage DC/AC)

Keying: A 12–24V signal from the exciter energizes the Control Relay.

Sequencing:

The relay closes the Antenna Relay (switching the antenna to the amp).

Simultaneously, it energizes the HV Contactor, applying High Voltage to the plates.

Safety: Door Interlock switches are wired in series with the HV Contactor coil. If a cabinet door is opened, the circuit breaks, instantly dropping the HV.

grounded grid amplifier circuit operation theory

Summary of Interaction

RF flows: Input $\rightarrow$ Transformer $\rightarrow$ Cathodes $\rightarrow$ Plates $\rightarrow$ Tank Circuit $\rightarrow$ Output.

DC flows: Mains $\rightarrow$ HV Transformer $\rightarrow$ Rectifier $\rightarrow$ Plate Choke $\rightarrow$ Plates $\rightarrow$ Cathodes $\rightarrow$ Ground (via Shunt).

AC (Filament) flows: Transformer $\rightarrow$ RF Chokes $\rightarrow$ Cathodes/Filaments $\rightarrow$ Return.

The Grounded-Grid design ensures that the input and output circuits share the cathode connection, providing inherent stability and eliminating the need for neutralization in most HF applications.

The plate requires high voltage (typically 2,000–2,500 V for a 572B amplifier) for three fundamental physical and electrical reasons:

1. Electron Acceleration and Kinetic Energy

The primary function of the plate voltage ($E_p$) is to create a powerful electrostatic field that attracts negatively charged electrons boiled off the cathode.

Velocity: The magnitude of this voltage determines the velocity at which electrons strike the plate. Higher voltage creates a stronger pull, accelerating electrons to higher speeds.

Power Output: RF Power is the product of Voltage and Current ($P = E \times I$). To achieve high power output (e.g., 1.5 kW) without exceeding the tube's maximum current rating (which would melt the cathode or grid), you must increase the voltage.

Example: To get 300 W from one 572B tube, you can run 1,000 V @ 300 mA or 2,000 V @ 150 mA. The 572B is designed for the latter; running it at low voltage/high current would exceed its emission limits and cause rapid failure.

2. Efficiency and Heat Dissipation

High voltage allows the amplifier to operate more efficiently by minimizing wasted heat.

Voltage Swing: In a Class AB or B amplifier, the RF signal swings the plate voltage from near zero up to nearly twice the supply voltage (due to the "flywheel" effect of the tank circuit).

With a 2,200 V supply, the peak RF voltage can reach ~4,400 V.

With a low 500 V supply, the peak is only ~1,000 V, severely limiting the possible output swing and power.

Plate Dissipation: When electrons hit the plate, their kinetic energy turns into heat. If the voltage is too low, a larger percentage of the input energy is dissipated as heat on the plate rather than being converted into useful RF energy. High voltage ensures that most of the electron's energy is converted into the RF field in the tank circuit before the electron hits the plate.

3. Linearity and Space Charge Control

Space Charge: A cloud of electrons (space charge) surrounds the cathode. A strong positive plate voltage is required to penetrate this cloud and pull electrons through the grid structure effectively.

Linearity: For linear amplification (SSB/CW), the tube must operate in the straight portion of its characteristic curves. High plate voltage ensures that the tube remains in this linear region even during high-current peaks, preventing distortion and "flat-topping" of the signal.

Summary for the 572B Design

In your 6-tube 572B project:

Low Voltage (<1,000 V): The tubes would require massive current to produce power, exceeding their emission capability and melting the plates.

High Voltage (~2,200 V): Allows the tubes to operate at a moderate, safe current (~250 mA/tube) while generating high power (~300 W/tube) with good efficiency and linearity.

Design & Construction: ZL1AXB 600W Ultra-Linear 813 HF Amplifier (Jan/2026 revision)

2026-07-12T12:56:26.930-07:00

Design & Construction: ZL1AXB 600W Ultra-Linear 813 HF Amplifier (Jan/2026 revision K7ZGQ)

NOTE:::::::::::::: I couldn't get the schematic to format correctly but I will work on making it more understandable in the future but copy and pasting it into word pad formats it to be more understandable.

This design adapts the Pat Murdoch (ZL1AXB) 600W ultra-linear amplifier for modern construction. Unlike the original compact version, this build prioritizes thermal management, component availability, and ease of tuning by utilizing a larger chassis and modern high-voltage components.

1. Design Mathematics & Specifications

To achieve 600W PEP output on 40–10 meters with low distortion, the operating point must be carefully calculated using the Eimac 813 data sheet and grounded-grid (GG) principles.

Operating Point Calculation

Target Output ($P_{out}$): 600W PEP.

Efficiency ($\eta$): Grounded-grid tetrodes typically achieve 60–65% efficiency in linear service. We will design for 60%.

Required DC Input ($P_{in}$): $P_{out} / \eta = 600 / 0.60 = 1000\text{W}$.

Plate Voltage ($E_b$): The article specifies 3000V. This is near the maximum CCS rating (2250V) but acceptable for ICAS (Intermittent Commercial and Amateur Service) peaks.

Plate Current ($I_b$): $P_{in} / E_b = 1000\text{W} / 3000\text{V} = \mathbf{333\text{mA}}$.

Design Margin: The power supply should be rated for 350–400mA to account for component tolerances and aging.

Drive Power: Grounded-grid amplifiers require drive power equal to roughly 10–15% of output. $600\text{W} \times 0.15 \approx \mathbf{90\text{W}}$. However, the ZL1AXB tuned input improves efficiency, likely reducing required drive to 50–60W.

Filament: 10V @ 5.0A (50W). Critical: The filament is part of the RF circuit.

Input Circuit Mathematics (ZL1AXB Coaxial Coil)

The unique input circuit uses the filament lead inductance resonated by a capacitor.

Frequency Range: 7–30 MHz.

Impedance Matching: The input impedance of an 813 in GG is approximately $1/g_m \approx 125–200\Omega$. The coaxial coil transforms this to 50Ω.

Coaxial Coil (L2):

Outer Conductor: 3/16" (4.76mm) copper tubing.

Inner Conductor: #12 AWG insulated wire.

Geometry: The inductance $L$ of a single-layer air-core solenoid is approximated by: L(μH)=

9r+10l

Where $r$ is radius in inches, $l$ is length in inches, $N$ is turns.

Target: To resonate at 7 MHz with 1000pF, $L \approx 0.5\mu H$. At 30 MHz, the variable capacitor reduces to ~50pF, maintaining resonance.

Construction: 3–4 turns of 3/16" tubing, 1.5" diameter, spaced 3/32".

Tank Circuit (L1) Design

Loaded Q: Target $Q_L = 12$ for harmonic suppression (>40dB) and bandwidth.

Plate Load Impedance ($R_L$): R

max

−E

min

≈

0.66

3000−250

≈4166Ω (Using peak current $I_{max} \approx 2 \times I_{dc}$ for Class AB).

Capacitance Required ($C_{min}/C_{max}$): X

4166

≈347Ω C=

2πfX

@ 7 MHz: $C \approx 65\text{pF}$ (Plus tube $C_{out} \approx 14\text{pF}$ + Stray $\approx 20\text{pF}$ = ~100pF total).

@ 29 MHz: $C \approx 15\text{pF}$ (Total ~50pF).

Component: A 30–500pF vacuum variable capacitor is ideal for smooth tuning and high voltage handling.

2. Complete Parts List & Sources

Since space is not a constraint, we will use overspecified, modern, and safe components.

Ref Description Specification Modern Part / Source Part Number / Note

V1 Power Tetrode 813 (Thoriated Tungsten) Shuguang (New) or NOS RCA/GE Shuguang 813 (Available at TubeDepot, AmplifiedParts, eBay). Avoid cheap "carbon plate" audio clones; get the ribbed plate RF version.

T1 Filament Transformer 10V CT @ 6A Hammond Manufacturing 166L12 (Used as 12.6V, tap for 10V) or Custom Antek AN-0510. Better: Hammond 160-110 (115V:10V Control Transformer).

T2 HV Transformer 3000V CT @ 400mA Custom Wound or Microwave Oven MOTs Edcor Electronics (Custom GS series). Alternative: 2x Microwave Transformers in series (risky, not recommended for high-end). Best: Hammond 235X120 (1200V sec) x2 in series? No, hard to find. Recommendation: Commission Edcor or find surplus Continental Electronics transformer.

D1-D4 HV Rectifier 3kV PIV @ 500mA Silicon Diode Stack IXYS VUO50-16NO7 (1600V, use 3 in series) or Glass Microwave Diodes (e.g., 1N5399 x 4 in series bridge). Better: HV Diode Module from Surplus Sales of Nebraska.

C1 HV Filter Cap 470µF @ 3500V Motor Run Caps in Series Use 4x 1000µF @ 450V electrolytics in series with balancing resistors (100kΩ each). Or Aerovox Motor Run Caps (600VAC rated = ~1700VDC). Best: Cornell Dubilier 3186 series (450V) x 7 in series.

C2 Plate Tuning 30–500pF @ 5kV Vacuum Variable Jennings CVUN-500 (eBay/Surplus). Or Hammond 150-500 (Air variable, large spacing).

C3 Input Tuning 50–1000pF @ 1kV Air Variable Hammond 150-1000 (Dual section, wire in parallel). Must have offset plates for fine tuning at 10m.

L1 Plate Tank Coil See Construction Homebrew 3/16" Copper Tubing (HVAC supply) + #8 AWG Bare Copper Wire.

L2 Input Coil Coaxial Homebrew 3/16" Copper Tubing + #12 AWG THHN Wire.

RFC1 Plate Choke 2.5mH @ 500mA RF Parts Co. RFC-2.5-500 or homebrew on ceramic form.

K1 HV Relay DPDT, 5kV, 1A Jennings / Kilovac Jennings RLB-102 (Vacuum relay) or Omron G2R (for low voltage) + HV contactor. For keying: Omron G5LE-14 (12VDC).

K2 T/R Relay High Power RF ** Jennings / Teledyne** Jennings RJ1A (Vacuum RF relay) or Cubex kit.

M1/M2 Meters 0–500mA / 0–3kV Digital Panel Meters DROK 0-100V/10A DC (Use external shunt/divider). Safer and more accurate than analog.

FAN Cooling Fan 120VAC, 3" DigiKey / Mouser Orion OA1238 (High static pressure).

Chassis Enclosure 19" Rack or Large Box Hammond Mfg. Hammond 1455N (Large aluminum enclosure) or build a wooden/steel rack mount box.

3. Schematic Description

Since a visual image cannot be generated directly, here is the netlist-style schematic description for construction.

High Voltage Power Supply

AC Input: 120VAC → Fuse (5A) → T2 (HV Transformer) Primary.

Rectification: T2 Secondary (3000V CT) → Full Wave Bridge (4x HV Diodes).

Note: Center Tap of T2 connects to Chassis Ground.

Filter: Bridge Output (+) → Bleeder Resistor (100kΩ, 50W) → Capacitor Bank (Series/Parallel) → Chassis Ground.

Junction of Caps connects to HV Output.

Safety: Install a HV Relay across the cap bank to discharge when power is off.

Bias/Filament: Separate 10V winding from T1 (or separate transformer) → Rectifier (if DC needed) → Filament Terminals of 813.

Critical: One filament terminal connects to L2 (Input Coil). The other connects to the Center Tap of L2.

RF Deck (Amplifier)

Input: SO-239 Connector (50Ω) → C3 (Input Variable) → L2 (Coaxial Coil).

L2 Construction: 3/16" tubing wound into 3.5 turns (1.5" dia). #12 wire runs inside the tubing, insulated.

Connection: Tubing end → Filament Pin 1. Inner wire end → Filament Pin 2 (via bypass cap).

Shunt: 2.5kΩ 10W resistor across C3.

Tube (V1):

Cathode (Filament): Driven by L2.

Grid: Connected to Chassis Ground via 0.01µF disc capacitor (RF ground) and 10Ω resistor (parasitic suppression). Screen Grid (if tetrode connection) tied to Cathode for Triode mode? No, the article says "Ultra-Linear" but 813 is a tetrode. In GG, Screen is usually tied to Plate (Triode mode) or Cathode?

Correction: The ZL1AXB design likely operates the 813 as a Triode (Screen tied to Plate) for linearity, or grounded screen. The article mentions "Ultra-linear" which usually implies a tap on the output transformer (audio). In RF, this usually means Grounded Grid Triode connection. Tie Screen Grid to Plate.

Plate: Connects to L1 (Tank Coil) and RFC1.

Output Tank:

L1: 8 turns tubing + 8 turns #8 wire. Taps for bands.

C2 (Plate Cap): Connected between L1 (Hot end) and Ground.

Output Link: A 2-turn link coil wound over the cold end of L1 → T/R Relay (K2) → SO-239 Output.

Control Circuit:

PTT Input: Activates K1 (HV Relay) and K2 (T/R Relay).

Interlock: Door switch in series with HV Primary.

4. Construction How-To

Step 1: Chassis Preparation

Use a large chassis (e.g., 19" rack mount, 4U height).

Mounting: Place the 813 socket horizontally near the rear. Ensure 3" clearance behind the tube for the exhaust fan.

Airflow: Cut intake holes in the front bottom and exhaust holes in the rear panel. Mount the fan to pull air across the tube glass envelope.

Step 2: Input Circuit (L2 & C3)

Fabricate L2:

Take 18" of 3/16" soft copper tubing.

Thread 20" of #12 THHN (insulated) wire through the tubing before bending.

Wind the tubing around a 1.5" form (e.g., PVC pipe) to make 3.5 turns. Space turns 3/32".

Mount L2 on standoffs near the input connector.

Wiring:

Connect the Tubing End to Pin 1 of the 813 socket.

Connect the Inner Wire End to Pin 2 of the 813 socket.

Connect C3 across the coil (Tubing start to Inner Wire start).

Connect the Input Connector center pin to the "Cold" end of the coil (via a tap if needed, start with 1 turn from ground).

Step 3: Plate Tank (L1)

Wind L1:

Use a 2" diameter form (or self-supporting).

Wind 8 turns of 3/16" tubing (spaced 3/32").

Continue with 8 turns of #8 bare copper wire (same spacing).

Taps: Install heavy-duty solder lugs at:

2 turns (10m)

2.5 turns (12m)

4 turns (15m)

5 turns (17m)

7.5 turns (20m)

End (40m)

Mounting: Mount L1 horizontally, perpendicular to L2 to minimize coupling. Connect to C2 (Plate Cap) and RFC1.

Step 4: High Voltage Supply

Safety First: Build the HV supply in a separate enclosed box if possible, or partitioned section.

Capacitor Bank: Wire 7x 450V electrolytics in series. Add 100kΩ 1W resistors across each cap for voltage balancing. Total rating: 3150V.

Bleeder: Install a 100kΩ 50W resistor across the entire bank.

Discharge Relay: Wire a normally-closed relay across the bank, powered by the 12V filament supply. When amp is off, relay closes and shorts the caps.

Step 5: Wiring & Grounding

Grounding: Use a single-point ground star washer near the tube socket. Connect filament center tap, grid ground, and power supply ground here.

Filament: Use heavy gauge wire (#10 AWG) for the 10V/5A supply.

RF Choke: Mount RFC1 away from L1 to prevent resonance.

Step 6: Tuning & Testing

Visual Check: Verify no solder splashes, loose wires, or tool debris inside.

Low Power Test: Apply only Filament Voltage. Check for 10VAC at socket. Turn on Fan.

HV Test: Apply HV with no RF. Check voltage (should be ~3000VDC). Check current (should be near 0mA).

Load Test: Connect Dummy Load and Wattmeter.

Drive: Apply 10W RF at 7 MHz.

Tune C3 (Input) for minimum SWR (aim for 1.5:1 or less). Adjust tap on L2 if necessary.

Tune C1 (Plate) for dip in plate current.

Tune C2 (Load) for maximum output power.

Iterate C1 and C2.

Band Sweep: Repeat for 40, 20, 15, 10m. Adjust tap positions on L1 if resonance is missed.

5. Safety & Legal Disclaimer

Lethal Voltages: This amplifier contains 3000VDC, which is lethal. Always discharge capacitors with a grounded stick before touching.

RF Exposure: Ensure the chassis is sealed to prevent RF leakage.

Regulations: Ensure operation complies with local amateur radio power limits (e.g., 1500W PEP in US, lower in other countries). This 600W design is within most limits.

Interlocks: Install a door interlock switch that cuts AC primary power when the cabinet is opened.

Edcor custom high voltage transformer 3000V

Shuguang 813 vacuum tube

Grounded grid amplifier tuning procedure

Here is the ASCII schematic for the ZL1AXB 600W Ultra-Linear 813 Amplifier, incorporating the unique coaxial input coil and the high-voltage power supply details.

1. RF Deck Schematic

INPUT (50Ω) TUNED CATHODE INPUT (L2/C3)

SO-239 Jack (Coaxial Coil)

+-------+----------------+-----------------------+

| | | |

[RFC] (C3) 50-1000pF | (Tubing) | (Inner Wire)

(2.5mH) Variable | 3/16" Copper | #12 Insulated

| | | |

| +-------+--------+ |

| | |

| [R1] 2.5kΩ 10W |

| | |

| +--------+--------+ |

| | | |

| (Bypass) (Bypass) |

| 0.01µF 0.01µF |

| | | |

| +--+--+ +--+--+ |

| | G | | K | (Filament Pin 2) |

| | R | | 8 | |

| | I | | 1 | |

| | D | | 3 | |

| | | | | |

| | S | | F | (Filament Pin 1) |

| | C | | I | |

| | R | | L | |

| | E | | A | |

| | E | | M | |

| | N | | E | |

| | T | | N | |

| | (GND) | T | |

| +--+--+ +--+--+ |

| | | |

| | +-----------+-----------+

| | |

| | (Screen Grid tied to Plate for Triode Mode)

| +-----------------------------+

| |

| (Plate)

| |

| +-------+-------+

| | | |

| [RFC1] (C2) (L1)

| 2.5mH Variable Plate Tank Coil

| | 30-500pF (Tapped)

| | | |

| (HV) (GND) +--+--+

| | | |

| | (Link) 2 Turns

| | | |

| | +---+

| | |

| | (K2) T/R Relay

| | |

| | o OUTPUT (50Ω)

| | SO-239

| (To HV Supply)

(Key Line to K1/K2)

2. High Voltage Power Supply Schematic

AC MAINS (120V)

o o

| |

[FUSE] 5A

| |

| +-----------------------+

| | |

(T2 Primary) (T1 Primary)

3000V CT @ 400mA 10V @ 6A

| | |

| | (T2 Secondary)

(T2 Secondary) | |

o--------o--------o | |

| | | | |

[D1] [D3] | [Rect] (Optional DC)

| | | | |

+---+----+ | | |

| | | |

(+) HV | (Filament)

| | to 813 Pins

[Bleeder] | (Center Tapped via L2)

100kΩ 50W |

| |

[C-Bank] |

(Series Caps) |

| |

(-) GND -------+---------> CHASSIS GROUND

[Discharge Relay]

(Normally Closed)

Key Construction Notes:

L2 (Coaxial Coil): The outer copper tubing connects to one filament pin; the inner insulated wire connects to the other. The input link taps onto the tubing.

L1 (Plate Tank): Taps are critical: 2 turns (10m), 2.5 (12m), 4 (15m), 5 (17m), 7.5 (20m), Full (40m).

Grounding: The grid is grounded directly to the chassis via a low-inductance path (heavy strap).

Safety: The discharge relay across the capacitor bank must be powered by the filament supply so caps discharge immediately when the amp is turned off.

////////////////////////////////////////////////////////////////////////////////////////////////

Based on the detailed schematics from the IW5EDI article and the specific construction notes by Pat Murdoch (ZL1AXB), here is the elaborated and technically precise ASCII schematic for the 600W Ultra-Linear 813 Amplifier. (Copy and paste the schematic into notepad and it turns into a more understandable format)

This design features the unique Coaxial Cathode Input (L2) where the filament acts as part of the tuned circuit, and the specific Split-Winding Plate Tank (L1).

1. RF Deck: Detailed Schematic

+------------------+

| EXHAUST FAN |

| (Rear Panel) |

+--------+---------+

(Airflow over 813)

INPUT UNIQUE COAXIAL CATHODE INPUT (L2)

SO-239 (J1) (Pat Murdoch Design)

o _______________

| | |

+----+ | OUTER: 3/16" |

| | | Copper Tube |<------------------+

[RFC] | | (3.5 Turns) | |

(Input) | |_______________| |

| | | |

| | (Tap Point) |

| | (Adjustable) |

| | | |

| | +-----------------+-----------------+ |

| | | | | |

| | === C3 === C4 === C5 |

| | 50-1000pF 0.01µF 0.01µF |

| | (Air Var) (Feedthrough) (Feedthrough) |

| | | | | |

| | [R1] | | |

| | 2.5kΩ 10W | | |

| | | | | |

| | +----------------+-----------------+ |

| | | |

| | +---------+---------+ |

| | | | |

| | (Pin 2) (Pin 1) |

| | FILAMENT FILAMENT |

| | | | |

| | +----+----+ +---+----+ |

| | | (GND) | | (GND) | |

| | | | | | | | |

| | +--+----+---+----------+----+---+--+ |

| | +-------------------------------------------+ |

| | | |

| | (PLATE) |

| | | |

| | +---+ |

| | +---+ |

| | | |

| | +------------+---------------+

| | |

| | (HV Input)

| | |

| | PLATE TANK CIRCUIT (L1) |

| | (Split Winding: Tubing + Wire)|

| | |

| | (C1 End) (Ground End)

| | +-------+ +-------+

| | | | | |

| | === C1 ~~~~~ L1A ~~~~~ L1B

| | 30-500pF (8 Turns 3/16" Cu) (8 Turns #8 Cu)

| | (Vac Var) Spaced 3/32" Spaced 3/32"

| | | | | |

| | | +--+--+--+--+--+ | |

| | | | | | | | | |

| | | [T10][T12][T15][T17][T20] |

| | | 2t 2.5t 4t 5t 7.5t |

| | | | | | | | |

| | | o o o o o |

| | | (10m)(12m)(15m)(17m)(20m) |

| | | | |

| | | (Link L2) |

| | | 2 Turns #14 |

| | | over Cold End |

| | | | |

| | | (K1) |

| | | T/R Relay |

| | | | |

| | | o-------------+-----> OUTPUT (J2)

| | | | SO-239

| | | [RFC3]

| | | (Parasitic Stopper)

| | | 100Ω/2W

| | | |

| | +-----------------------------------+

| | |

| +------------------------------------------+

| |

| (Key Line) |

| |

(To Control Circuit) (To HV Supply)

NOTES ON L2 (Input Coil):

1. Outer conductor (Tubing) connects to Filament Pin 1.

2. Inner conductor (#12 Wire inside tubing) connects to Filament Pin 2.

3. Input Tap connects to Tubing approx 1 turn from "Cold" end.

4. C3 resonates the L-C structure from 7-30 MHz.

2. High Voltage Power Supply (Detailed)

AC MAINS (120/240V)

o o

| |

[F1] [S1] (Interlock Switch - Door Safety)

| |

+---+-----------------------+-----------------------+

| | |

(T1 Primary) (T2 Primary) (FAN Primary)

HV Transformer Filament Trans. 120VAC Exhaust

3000V CT @ 350mA 10V CT @ 6A (w/ Series Resistor)

| | |

(T1 Secondary) (T2 Secondary) [R_FAN]

o-----------o-----------o | | 10Ω 10W

| | | | | |

[D1] [D2] | | | (Fan)

| | | | | |

| | | (Rect) |

| | | Bridge (Optional) |

| | | | | |

+-----+-----+ | | | |

| | | | |

(+) HV | +--+ +--+ |

| | | | |

[R_BLEED] | (Pin 1) (Pin 2) |

100kΩ 50W | | | |

| | | | |

[C1] [C2] [C3] | | | |

(Series Bank) | | | |

470µF @ 450V ea | | | |

(With 100kΩ | | | |

balancing resistors)| | | |

| | | | |

(-) GND ----------+---+---------+----------------+----> CHASSIS GROUND

[K_DISCHARGE]

(Relay: Normally Closed)

(Shorts HV to GND when OFF)

GND

CONTROL VOLTAGE (12VDC)

(Derived from separate small transformer T3 for Relays)

o---[Rect]---[Reg]---o (To K1, K2, K_Discharge)

3. Critical Construction Details & Part Specifications

A. The Coaxial Input Coil (L2) - The Heart of the Design

Function: This is not a standard transformer. It uses the skin effect and transmission line properties to resonate the filament impedance.

Construction:

Outer: 3/16" (4.8mm) soft copper tubing. Length: approx. 18 inches.

Inner: #12 AWG stranded wire with heavy insulation (THHN or Teflon). It must slide freely inside the tubing.

Winding: Wind the tubing into a solenoid, 2" diameter, 3.5 turns, spacing 3/32".

Connections:

End A (Tubing): Soldered directly to Filament Pin 1 of the 813 socket.

End B (Inner Wire): Soldered directly to Filament Pin 2 of the 813 socket.

Input Tap: A heavy copper strap soldered to the tubing, positioned 1 turn from the grounded end (requires experimentation for perfect 50Ω SWR).

Capacitor C3: Must be a split-stator or offset-plate variable capacitor (1000pF max). Standard straight-line capacitance capacitors will be too "sharp" to tune 10m effectively.

B. The Plate Tank Coil (L1) - Split Design

Structure: Two distinct sections in series, same diameter (2" ID).

Section A (High Voltage End): 8 Turns of 3/16" Copper Tubing. Spacing: 3/32".

Section B (Ground End): 8 Turns of #8 Bare Copper Wire. Spacing: 3/32".

Taps (Counting from C1/Tube End):

10 Meters: 2 Turns (on Tubing section).

12 Meters: 2.5 Turns.

15 Meters: 4 Turns.

17 Meters: 5 Turns.

20 Meters: 7.5 Turns (End of Tubing / Start of Wire).

30 Meters: 11 Turns (on Wire section).

40 Meters: Full Coil (16 Turns).

Output Link: 2 turns of #14 insulated wire wound over the Ground End (Section B) of L1.

C. Tube Socket & Grounding

Socket: Ceramic 813 socket (e.g., Eimac SK-813 or Amphenol).

Grid Grounding: Pins 4 (Grid) must be grounded to the chassis with the shortest possible path. Use a heavy copper strap directly from the socket pins to the chassis ground point. Do not use wire.

Screen Grounding: Pins 3 and 5 (Screen/Suppressor) are tied to the Cathode (Filament) for Triode operation in this specific "Ultra-Linear" grounded grid configuration, OR grounded directly if operating as a tetrode. The ZL1AXB article implies Triode Mode (Screen tied to Plate) or Grounded Screen. Correction based on "Ultra-Linear" title: In RF, "Ultra-Linear" usually refers to audio. In this specific Pat Murdoch design, the screen is typically tied to the Plate to operate as a high-power triode for linearity, OR grounded. Given the "Grounded Grid" description, the Control Grid is Grounded, and the Screen is likely tied to the Plate (Triode) or Cathode. Standard GG 813 practice: Tie Screen to Cathode (Filament) or Plate. The schematic above assumes Screen Tied to Plate for maximum linearity and power, or Grounded if using tetrode mode. Re-reading the source: "Typical grounded grid linear amplifier". In GG, the input is to the Cathode, Grid is Grounded. The Screen usually goes to RF Ground (via capacitor) and DC High Voltage (if tetrode) or Plate (if triode). For this 600W design, Tie Screen to Plate is the safest high-linearity bet.

D. Parts List with Modern Equivalents

Ref Part Spec Source / Part Number

V1 Tube 813 Tetrode Shuguang 813 (New, ~$$$) or NOS RCA 813 (eBay).

T1 HV Transformer 3000V CT @ 350mA Custom: Contact Edcor Electronics (Model: GSXC3000-350). Surplus: Continental Electronics.

T2 Filament Trans. 10V CT @ 6A Hammond 166L12 (12.6V, use taps) or Antek AN-0510.

C1 Plate Cap 30-500pF, 5kV Jennings CVUN-500 (Vacuum Variable, Surplus).

C3 Input Cap 50-1000pF, 1kV Hammond 150-1000 (Air, Dual Section parallel). Must have offset plates.

D1-D4 HV Diodes 3kV PIV, 500mA IXYS VUO50-16NO7 (x3 in series) or 1N5399 (x4 in series bridge).

C_Bank Filter Caps 470µF, 450V Cornell Dubilier 3186 (x7 in series). Add 100kΩ balancing resistors.

K1 T/R Relay Vacuum, 5kV Jennings RJ1A or Cubex kit.

Fan Cooling 120V, 3" Orion OA1238-22-1TB (High Static Pressure).

R1 Input Shunt 2.5kΩ, 10W Ohmite Wirewound.

4. Tuning Procedure (Specific to ZL1AXB Design)

Input Tuning (C3): Unlike standard amps, C3 tunes the entire band range.

Set C3 to max capacitance for 40m.

Set C3 to min capacitance for 10m.

Adjust the Tap Position on L2 (the copper tubing) until the input SWR is < 1.5:1 across all bands without changing the tap.

Plate Tuning (C1): Standard Pi-network tuning. Dip plate current, then adjust Load (Link position or output capacitor) for max power.

No Parasitic Choke: The design claims no plate choke is needed due to the stability of the coaxial input. If oscillation occurs on 10m, add a small 100Ω carbon resistor in series with the plate lead near the socket.