In Figure 1, here’s a purchased PC board module based on the Diodes Inc. PAM8403 stereo class-D power amplifier, 2 watts per channel. I was working on a project that used several of these prefabricated ‘8403 boards, and I needed a way to check their outputs with a ‘scope to make sure they were not “clipping” (in analog-speak) or running out of headroom. I planned on building a monitoring device to easily and quickly run tests on multiple devices. But first, I decided to set up one board all by itself so I could figure out how to run my test procedure. To make working with this one board a bit easier, I mounted it on a larger piece of perfboard. I added a few 0.1″ pin connectors to make it easier to connect power, signal inputs, and speaker outputs.

Since the output from each amplifier is a series of pulse-width modulated rectangular waves (see Figure 2 and Figure 3), you can’t simply put a ‘scope probe on an output pin and see if it’s clipping.

Instead of trying to evaluate the amplifier’s performance by scoping its outputs, you need to perform the same action that the speaker does – you need to integrate the applied waveform (i.e., low-pass filter it) and look at that with the scope.

This brings us to the actual test jig that I built. Recall that I would be testing multiple devices; I needed a way to quickly evaluate the ‘8403 modules that were piggy-backed onto my PC board. The class-D amplifier output is a type of differential output. On one amplifier output, there is a stream of rectangular waves of varying duty cycle from approximately zero to 100% (50% at idle). On the other amplifier output, there is a matching synchronized stream varying from approximately 100% to zero.

The load (the speaker) is tied across the two outputs rather than from output to circuit common or ground. This is commonly referred to as a bridge-tied load or BTL. I needed a way to monitor the amplifier outputs in differential mode and then produce the analog equivalent through integration. In Figure 4, here’s what I built.

There are two boards here. One is a PC board, and the other is a perfboard. The perfboard contains a bit of circuitry: a quad op amp and a handful of resistors and capacitors. I mounted both boards on a piece of scrap plywood using standoffs and 4-40 hardware. I also added a copper foil ground plane beneath the perfboard. The use of scrap lumber to build test jigs is a favorite method of mine – it’s inexpensive and can easily accommodate various amounts of circuitry. It’s not perfectly shielded, but that’s not always necessary.

The PC board is an Instrumentation Amplifier Eval board I acquired during my time working for Analog Devices. I populated it with an AD8421 and powered it from ±15 VDC. I equipped the (true) differential input with a connector to match the pins on the ‘8403 amplifier boards I was building. I also added some small ferrite toroids to reduce the high-frequency content. The perf-board is a four-pole active low-pass filter with a corner frequency of 10 kHz and a pass-band gain of unity (1 V/V). Figure 5 shows the set up in use. Figure 6 and Figure 7 show the ‘scope output with a 2 kHz signal input, below and above the level where clipping and distortion start.

This setup made it easy to evaluate my devices. I could ensure they were all built correctly with no defective components.

I built another piece of audio equipment to make it easy to test microphones that needed phantom power. A phantom power supply provides (usually) +48 VDC to microphones that have built-in preamplifiers. The term phantom and the use of 48 VDC derive from the early days of telephony and tube preamplifiers. A low-power tube amplifier’s plate circuitry could be operated from 48 V effectively. Phantom circuitry allows a second circuit (the power, in this case) to be overlaid on the primary circuit (the audio).

With a balanced audio line – the kind that typically uses XLR connectors on the cable – there are two wires carrying the audio inside the grounded shield. The signal on one wire is 180° out of phase with the signal on the other. Stated another way, the audio is sent differentially through the cable. The 48 VDC is coupled onto both these wires through two almost identical resistors (1% tolerance) sized at a few thousand ohms. This means the DC is sent via common mode (with respect to ground) on the balanced audio line. The resistors don’t appreciably load the audio from the microphone, which typically has a source impedance of perhaps 50Ω to 150Ω. The circuit I built to provide the phantom power is shown in the schematic in Figure 8. Power comes from a 48 VDC wall-wart plugged into J1. I added a network of one more resistor and three capacitors to minimize the power supply ripple.

The apparatus is shown in Figure 9, again, built on a scrap of plywood with some copper foil added as a low impedance ground plane. The left cable goes to the female XLR connector (J2); the right cable goes to the male XLR connector (J3). The power cable comes in at the top and plugs into J1.

Another useful jig to have for audio work is a microphone matching/isolating transformer, again built on plywood and using parts I had available to control costs. See Figure 10. The microphone plugs into the cable that comes in from the top of the picture (via J1 as seen in Figure 12). There is a lo-Z to hi-Z transformer mounted upright on the board, followed by a ¼” audio connector. This connector is a two-circuit connector, sometimes referred to as a stereo audio jack or a T-R-S connector. T-R-S refers to tip-ring-sleeve, the three electrically isolated sections of the jack and corresponding plug. As before, the terminology derives from telephony technology.

This jig is set up as balanced in, balanced out. Occasionally, I used this jig simply as an adapter to convert an XLR connector to a ¼”audio connector with no need for an impedance transformation. To completely bypass the transformer, I needed a 4PDT switch. I had an abundance of DPDT slide switches, so I used two and ganged them together as seen in more detail in Figure 11.

Since I sometimes plugged the hi-Z side directly into an amplifier that was configured as single-ended, I added a switch (SW2) to ground the ring circuit. Refer to Figure 12 to see how switches SW1 and SW2 function.

As a variation of the previous transformer jig, and to address my concerns regarding the pickup of stray electrical fields (RF and audio frequency), I built the test jig seen in Figure 13. Electrically, much like the version in Figures 10 to 12, but without the switches, I built this to minimize extraneous pickup. It is configured as Lo-Z balanced in and Hi-Z unbalanced out. To provide the needed shielding and to minimize cost, I wrapped sections of cardboard (from the back of a pad of paper) with adhesive-backed copper tape and soldered these panels together. Not surprisingly, it’s fabricated on a scrap of plywood.

In Figure 14, note the added RCA pin jack. I installed that to provide an easy ‘scope test point to monitor the signal going to my amplifier while still maintaining my shielding.

Speaking of amplifiers, if you are testing high-power units and don’t want to blast your eardrums into oblivion, you need to operate the amp into a dummy load. Typical loads are made of 4Ω, 8Ω, or 16Ω resistors. I built a simple one that I can use as either a stereo 8Ω load or, with suitable jumpers, a mono 4Ω or 16Ω load. I happened to have four 15Ω, 1%, 50W power resistors. With two in parallel, I had a 7.5 Ω, 100W resistor – close, but no cigar. I also had a couple of 0.5Ω, 10%, 5W resistors on hand. With one of those in series with each 7.5Ω, I had my dual 8Ω dummy load resistors. An additional advantage of the 7.5Ω+0.5Ω arrangement: It provides a convenient voltage divider point that can drive headphones without damage – it’s functioning as a 1:16 voltage divider.

Referring to Figures 15 and 16, the input is applied via the left two screw terminals on the barrier block. The amplifier’s output can be monitored via a ‘scope at the far-left RCA jack or via headphones at the second from the left RCA jack. The second or right-hand channel is a mirror image of the first with respect to the connections.

While the power resistors have a rating of 100W, realistically, that presumes an infinitely large heat sink. Since I don’t have one of those available in my lab, I used a substantially smaller than infinity aluminum plate. Heat sink size and thermal conductivity between the resistors and the heat sink; and the heat sink and the ambient air surrounding the heat sink; and the temperature of the ambient air affect how hot the resistors will get versus the power being dissipated in them. So that smaller size heat sink means the resistors’ power rating must be derated from 100W, but the rating is still big enough for most of my testing, especially with short duty cycles of applied power. The schematic of the dummy load is shown in Figure 17.

Also on the topic of amplifiers, sometimes you need to amplify low-level signals (e.g., from a guitar) enough to be useful (i.e., be able to hear them). I built a small headphone amplifier using an LM386. It actually can drive a speaker to a quite usable level. One minor detail that makes it slightly problematic: The typical LM386 circuit has a fairly low input impedance of 50kΩ. To work around this, I added a simple JFET source follower circuit to buffer the input of the LM386. The JFET input is bootstrapped to make the input impedance super high. I built this all up on a perfboard and made provisions for gain switching between 20 V/V and 200 V/V. I also provided a direct output to drive a speaker and a resistively padded output to drive stereo headphones. The completed amplifier is shown in Figures 18, 19, and 20.

The schematic is shown in Figure 21.

Please post suggestions of test jigs and apparatus you’ve constructed for your own specific applications.

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IMS 2026 took place the week of June 7-12 at the Menino Convention Center in Boston’s Seaport District. Over 400 large and small exhibitors displayed their components and equipment in the exhibition hall. EEWorld was there. Through videos and photos, here’s a roundup of the exhibits:

Aaronia

Aaronia’s large booth featured spectrum analyzers, antennas, and software. The booth featured large screens showing real-time spectral analysis and, in my opinion, stole the show for the most color. The photo shows Spectran Ultra baseboard, which produces 16 phase-coherent outputs, each at 40 Gsamples/sec with up to 20 GHz bandwidth per channel. It connects to networks and other equipment through eight Ethernet and eight USB ports.

Analog Devices

The photo shows a demonstration of the AD9084 Apollo MxFE Quad, 16-Bit, 28 Gsample/sec RF DAC and Quad, 12-Bit, 20 Gsample/sec RF ADC. In this application, the oscilloscope shows the boot sequence ending once the RF output starts.

Anritsu

Anritsu showed the latest in its Tensor series of vector-network analyzers, announced on June 9 at IMS. In the video, you’ll see the latest hardware improvements to this four-port VNA: four signal sources for four ports. Watch the video to the end for a preview of the instrument’s special new features.

Bird

Known for its handheld testers for RF installation and maintenance, Bird displayed the SiteHawk SK-6000 Cable and Antenna Analyzer, which operates at frequencies up to 6 GHz. The company also offers models for frequencies to 4.5 GHz and 9 GHz. The instruments use frequency-domain reflectometry to detect faults such as breaks in cables.

Copper Mountain

The company known for its USB-connected vector-network analyzers (VNAs) came to Boston with several instruments, including the BFx-02 Frequency Extension Base for S-Parameters Measurement. Using this instrument and frequency extenders, engineers can make S-parameter measurements on components, systems, and materials at frequencies beyond those usually accessible for traditional VNAs.

Emerson NI

The company formerly known as National Instruments produces modular RF instruments such as its PXI Vector Signal Generators, Analyzers, and Transceivers. In this exhibit, the company used its vector-signal transceiver in combination with a software-defined radio to develop an RF test bench.

Keysight

Long known for its RF test equipment, Keysight Technologies demonstrated its 32 GHz XA5 Signal Analyzer, among other equipment. The swept-spectrum analyzer features 2 GHz analysis bandwidth. As the video shows, the instrument’s measurement speed provides a significant improvement over previous swept-spectrum analyzers.

Mathworks

Working with Analog Devices and Leonardo, Mathworks demonstrated an RF digital twin. The model simulates a radar and satellite RF system using components from Analog Devices. With the model, engineers can alter parameters and see how different settings affect system performance.

Microchip

Havily invested in wireless and wired network system timing, Microchip exhibited oscillators for GNSS timing systems.

Pico Technology

Known for its USB-connected oscilloscopes and network analyzers, Pico technology exhibited several instruments including the PicoScope 500 Series featuring what the company calls it’s FlexRes signal resolution, which produces from 8 bits to 16 bites with the tradeoff being sample rate (62.5 Msamples/s at 16 bits). The model shown features four analog inputs with a connector for adding logic-level signal capture.

Pickering Interfaces

Specializing in PXI modular switching instruments, Pickering Interfaces came to IMS with (l-r) a PXI/PXIe Triple SP6T, 18 GHz, 50 Ω, SMA, Failsafe Microwave Multiplexer, a Quad 4-Channel, 4GHz, 50Ω, SMB MEMS RF Multiplexer, and a Single SP6T, 67 GHz, 50 Ω, SMA-1.85, multiplexer. This model uses a cable to connect a remote multiplexer, which lets you install a single-slot module in the chassis.

Qorvo

Fixed wireless access (FWA) has emerged as one of 5G’s successes and one of the few use cases for mmWave signals. Qorvo makes parts that you can use to up convert and down convert carrier frequencies. The demonstration uses the AWMF-0224, a 24 GHz to 30 GHz Dual-channel IF transceiver that performs the frequency conversion. The demonstration also uses the AWMF-0221 Dual Polarization Quad 4×2 Beamformer. The combination can achieve data rates of 3 Gbits/s at distances up to 3 km.

Samtec

The company known for connectors and cables demonstrated its Bulls Eye BE130A, a 130-GHz test-point system. The video provides insight into the system, which can accommodate up to 32 channels.

Rigol

While RF engineers likely gravitated to Rigol’s DNA6264 four-port VNA, what caught my eye was the MHO984 oscilloscope. This 800-MHz, 12-bit model on the MHO900 series features four channels. Lower-cost models start at 350 MHz. What the photo doesn’t show is the oscilloscope’s footprint, or lack thereof. It’s just 77 mm (about 3 in.) deep.

Rohde & Schwarz

IMS is the company’s biggest event, and the venerable company was out in force. As always, the company exhibited RF test equipment that included spectrum analyzers, vector-network analyzers, signal generators, and oscilloscopes. The video shows a demonstration using a signal generator and spectrum analyzer to characterize a 3-GHz RF amplifier.

Sage Instruments

Known for field installation and maintenance measurement equipment, Sage Instruments exhibited its WSA-408 Wireless Signal Analyzer. This handheld instrument includes an 8-kHz-to-8-GHz spectrum analyzer, an interference analyzer, and a 5G signal analyzer.

Siglent

Siglent exhibited its line of equipment, including spectrum analyzers, oscilloscopes, and RF signal generators.  

SignalHound

The RF test equipment company’s USB-connected equipment includes VNAs, signal generators, phase-coherent receivers, and phase-noise testers. The photo shows its VNA400, a two-port, 40 GHz instrument connected to a bandpass filter.

Poster session: DC-DC converter switches at 200 MHz

The exhibit hall featured more than just companies showing off their products.

Tabor Electronics

This maker of portable, benchtop, and modular RF test equipment demonstrated equipment designed for software-defined radio (SDR) testing. On the left in the photo is the LSX2091D, a 20 GHz microwave signal generator.

Uni-Trend

Known for basic test instrumentation, including RF equipment, Uni-T exhibited one of its oscilloscopes, as well as (top-to-bottom) a 60 MHz function generator, a 5-1/2 digit multimeter, and a power supply.

IMS 2027 will take place from May 23rd to 28th in San Antonio, TX. It returns to Boston in 2031.

EEWorld Online related articles

IMS 2024 roundup: test equipment
Spectrum measurements shed light on sign malfunctions
Antennas to bits: Modeling real-world behavior in RF and wireless systems
Tryout: Uni-T UPO1202 200 MHz oscilloscope
Tryout: Uni-T UTG962E Function/Arbitrary Waveform Generator

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B&K Precision has introduced the DML Series DC electronic load system, a modular platform with a 4U mainframe that supports up to four channels and 800 W. The system uses swappable single- or dual-channel load modules with front-panel inputs providing up to 80 V and 80 A, or up to 160 A in parallel operation, and supports constant current, constant voltage, constant resistance and constant power modes for testing DC power supplies, batteries, fuel cells and photovoltaic arrays. The front panel allows independent control of each module and setup of parameters including voltage, current, slew rate and pulse width, with settings saved to internal memory for recall. Remote control options include LAN, USB, RS232 and GPIB with SCPI support, and the system also offers fast transient operation up to 25 kHz in CC mode, sweep modes and built-in protection features.

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The instrument captures heat flow, evolved gas, and electrochemical data (voltage monitoring plus charge/discharge) simultaneously, in a single run on a single intact cell. That’s a meaningful change from running separate tests on separate samples and trying to correlate results afterward. The temperature range spans -80°C to 600°C, which the company positions as best-in-class for this category, wide enough to cover both low-temperature performance work and high-temperature thermal runaway events in one sweep.

TA Instruments is positioning this as filling the gap between materials-level thermal analysis and full-cell DSC testing. Compared to accelerating rate calorimetry, which needs large-format cells, dedicated facilities, and long test cycles, this approach lets researchers work with intact coin cells earlier in development. For labs running early-stage characterization, that’s a faster, lower-overhead path to the same kind of insight.

It can also be coupled with mass spec, FTIR, or GC-MS for more detailed gas analysis when needed.

Available for order now, with shipments starting August 2026.

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Tektronix® has introduced wideband shunts for IsoVu isolated current probes and the THDP0400 high voltage differential probe for power electronics validation in EVs, industrial systems, AI data centers and power conversion applications. The shunts support bandwidth up to 250 MHz for low-current transient measurements and include built-in communication, temperature compensation and fuse protection, while the THDP0400 provides 400 MHz bandwidth and ±2000 V measurement capability for floating high-voltage measurements, wide-bandgap semiconductor development and double-pulse testing. Together, the two accessories help engineers capture fast transients and low-current behavior with less setup complexity by working within a single oscilloscope-based measurement workflow.

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Engineers are problem solvers. It’s what we do. When working on a project at your bench, you probably have the usual collection of hand tools, test equipment, and a soldering station or two. Sometimes you need more than these tools. Sometimes, you have to solve a practical problem before solving the problem at hand. That’s where adding and building jigs, holders, compartments, and other things can help.

Some common add-ons are the alligator clip “helping hands” (Figure 1). The clips let you hold PCBs, wires, connectors, or components, which certainly helps when soldering. A magnifying glass enhances the experience, especially when working with small parts.

Figure 2 shows a PCB vice, another device that can hold a PCB during soldering. Several such tools are on the market. See EEWorld’s review of the PCBite kit from Sensepeak.

You may also need special tools, stands, or jigs to assist in construction and testing. One of my projects involves adding on-board electronics to electric guitars. I do enough of this work that I don’t want to prop up the guitar body on whatever cardboard or plastic project boxes I have in my lab and start soldering. Instead, I built a couple of simple jigs. One allows me to hold the guitar upside down with strings facing the bench but not touching anything (so the strings aren’t muted). See Figure 3 and Figure 4. This makes it easy to tinker with the circuit boards I’m installing on the back of the body. A second jig allows me to clamp the guitar body sideways (without marring the body) when that’s needed. See Figure 5.

Another handy device is a compartmentalized component and hardware holder: an egg carton. There’s one visible in Figures 3 through 5. These help keep mounting hardware from vanishing under the various bench equipment or, worse yet, into the vacuum cleaner.

Preventing ESD damage

Let’s now consider assembling a PC board populated with surface-mount components that might be damaged by electrostatic discharge (ESD). Let’s assume you know that ESD should be mitigated with a conductive mat, upon which you place your PCB. You could buy a conductive mat, but an inexpensive mat can be equipped with a piece of black conductive IC foam, shown in Figure 6. Note the clip lead that attaches and grounds the mat. I added a small piece of brass shim stock to make it easier to clip onto the mat without chewing through it.

On my bench, I added the grounding connectors (banana jacks), also visible in Figure 6. The plate is connected to the same earth ground connection that my circuit breaker panel uses. It’s a combination of a ground stake and a connection to the incoming copper water supply line. I’m assuming there will never be any high magnitude ground fault currents flowing while I’m working at my bench, so my ground connection should really be at a (true) earth potential.

I also realized I would scrape my knees on the tips of the banana connectors on the underside of my ground block, so I added a piece of ¼-in. rubber tubing (slit the entire length and attached with hot melt adhesive), shown in Figure 7.

Besides these ESD features, you will likely want a magnifier to help place tiny surface-mount components. A magnifying glass can provide some help, but I prefer a video microscope. These are available in various magnification levels and price points. I’m using an inexpensive device from Amazon with a mag range of 40 X to 1000 X. I mounted it on a used bench light boom I got from Goodwill. See Figure 8. For the video monitor, I’m using a laptop computer that originally ran Windows 10. Since that’s no longer supported, I wiped the hard drive and converted the laptop to a Chromebook.

Sometimes, you need a camera to document your work or to see things on a large screen. You surely need something to hold the camera, such as a boom or tripod. Figures 9 and 10 show the details of how I attached a camera to the end of the boom.

Hold components in place while soldering

For around $30, I had a useful setup that mostly made soldering tiny components with my hot air soldering station pretty easy. Occasionally, when attaching parts using a soldering iron and 0.020″ solder, I found I would sometimes bump the component out of position. For those occasions, I built a tool to hold the component while soldering the first few pins. See Figure 11. I built this using a tuna fish can (without the tuna), some pea gravel, epoxy glue (acting as a sort of potting compound), a piece of threaded stock plus nuts and washers, a piece of 14 AWG copper wire, and an old ballpoint pen to act as a spring-loaded pin. I changed the ink cartridge and spring locations around to get the desired spring action. This applies just enough pressure to keep things in place.

For soldering with hot air, solder paste is the preferred method. I was using some paste that had much higher viscosity than I would have liked, but I didn’t want to waste it, so I built a small heater box to keep it warm, which lowered its viscosity. See Figure 12. It’s simply a spare electrical junction box, a [grounded] power cord, a 15 W candelabra light bulb, and a scrap of stainless-steel screen. Once again, a low-cost solution.

Returning to my engineering work on guitars and guitar electronics, I needed a way to evaluate different styles of pickups and methods of inducing sympathetic vibration into the strings. Referring to Figure 13, I built a jig with some used strings, a piece of MDF (medium-density fiberboard), a piece of pine trim board, miscellaneous bits of hardware, small blocks of (scrap) oak (acting as bridges), and a piezo (PZ) disc salvaged from a PZ beeper.

In use, I placed my electromagnetic drivers beneath the strings at different points between the left and right bridges and noted their effectiveness as drivers.

When testing line-powered equipment, you may find that the equipment suffers from internal short circuits. This problem commonly arises from power transformers with shorted turns within windings, shorted rectifier diodes, shorted filter capacitors, or other supply rail shorts in the equipment under test (EUT). We need a good way to detect and analyze these problems without repeatedly blowing the EUT’s built-in fuses or tripping its circuit breakers.

The clever way is by using a Variac to power the EUT and inserting a medium to high wattage incandescent light bulb in series with the power being supplied from the Variac. For line-powered equipment, we don’t want to use a suicide cord and a bunch of Radio Shack clip leads. A more elegant solution is shown in Figure 14.

With the appropriate wattage bulb, normal current draw causes little heating of the bulb’s filament. Based on the positive temperature coefficient of tungsten, the barely warm filament’s voltage drop is very low, and the EUT receives close to the full line voltage. You can crank up the line voltage slowly, and if excessive current flows, you’ll know because the light bulb (but not the fuses) will glow brightly.

The schematic for this device (Figure 15), like many of my preceding test jigs, is simple. I recommend having several different bulb wattages on hand, from 40 W to 250 W. You should procure several now before they become obsolete.

Lastly, consider those times when you’re trying to spray contact cleaner into a volume control whose opening is obscured due to its mounting position. Trying to bend the small-gauge tubing that comes with the spray can rarely work — it will probably kink. I worked around this by fabricating an elbow from some scraps of larger-gauge plastic tubing. I was able to bend it by carefully heating it with my hot-air soldering station. The larger gauge made it far less likely to pinch shut at the bend. See Figure 16.

In part 2, we’ll take a close look at some of my custom-built test jigs to simplify some of the audio testing that I do.

What test jigs or time-saving tools have you devised? Tell us in the comments.

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Ranatec AB has introduced standard 8-channel and 12-channel attenuator box configurations for RF and microwave test systems. The off-the-shelf units are intended for telecom production, electronics manufacturing and automated test environments where users need faster deployment and channel counts matched to specific setups without custom development. The added configurations give test engineers more flexibility when scaling RF measurement systems while helping reduce sourcing time and system integration effort. They are aimed at applications that require repeatable signal conditioning in production test, verification and troubleshooting workflows.

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Pico Technology has introduced the PicoScope 5000E Series USB-C oscilloscopes for low-amplitude analog, digital and mixed-signal measurements. The series provides 16-bit resolution, bandwidths up to 200 MHz, sample rates up to 2.5 GS/s and capture memory up to 1 GS, while Plus models add switchable 8-bit operation with bandwidth up to 500 MHz, sample rates up to 5 GS/s and memory up to 2 GS for faster edge and serial-bus analysis. The scopes are designed for applications such as power integrity, sensor measurement, analog front ends, medical electronics and mixed-signal debug, with a noise floor below 22 µV RMS and better than -73 dB THD for low-level signal analysis. Additional features include 4 analog channels with optional 16 digital channels on MSO models, more than 40 serial protocol decoders, advanced triggering, a 200 MS/s 14-bit AWG, FFT analysis and PicoSDK® support for C, C#, C++, Python, MATLAB and LabVIEW.

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In this series on strain gauges, we’ve looked at quarter- and half-bridge configurations. In this final part, we will look at a full-bridge implementation with four active elements.

Q: Where do these active elements get placed on our test specimen?
A: Figure 1 shows one possibility. With stress applied in the direction of the arrow, the red elements on top, each of length l in the unstressed state, will expand by Dl, and the blue ones of the same initial unstrained length will contract by Dl.

Q: How do we connect these elements?
A: Figure 2 shows these elements connected in the Wheatstone-bridge circuit, where VEX is the excitation voltage, and V_O is the output voltage proportional to strain e.

Q: How do we derive strain from VO?
A: We basically have two voltage dividers with the voltages at the positive and negative terminals of our voltmeter as follows:

Then, V_O is V₊–V_–:

Given a gauge factor (GF), we want to find V_O as a function of strain. From part 3, we know we can substitute R(GF)e  for DR:

For GF = 2, the Figure 3 traces show V_O as a function of e  for full-, half-, and quarter-bridge circuits along with the corresponding equations. Note that for the quarter-bridge case, the equation represents a linear approximation (black trace) of the actual nonlinear response (dashed black trace) that we calculated in part 2. Note that as we double the number of active elements, V_O doubles, thereby increasing our measurement sensitivity.

Figure 4. Lead resistance RL can drift with ambient temperature (a), but an additional sense lead can provide compensation (b). (Image: Rick Nelson)

Q: Couldn’t we also increase sensitivity by increasing the excitation voltage?
A: Yes, increasing the excitation voltage would increase the sensitivity and signal-to-noise ratio. However, this approach has a downside: it increases power dissipation in each strain-gauge element, with power increasing with the square of the excitation voltage. As power increases, the gauge becomes susceptible to self-heating, which can cause the gauge to expand and contract relative to the test specimen [1], thereby introducing a thermal error that is difficult to compensate for. Consequently, the excitation voltage should remain as low as possible while maintaining an adequate signal-to-noise ratio.

Q: What about lead-length resistance while measuring strain on large structures such as wide-body airframes?
A: As illustrated in Figure 4 in a quarter-bridge configuration, our strain gauge R_X attaches to our data-acquisition system through two leads, Lead 1 (red) and Lead 2 (blue), of resistance R_L each, so our meter will respond as if R_X=R_X+2R_L. If we know R_L, we can compensate, but R_L will vary with ambient temperature, making it difficult to know its value exactly.

The gold-standard approach for compensating for lead resistance is to use a four-wire Kelvin measurement, but that incurs additional cost. With the Wheatstone bridge, however, we only need to add one additional sense lead and make a minor wiring change, shown in orange in Figure 4b. Note that with this approach, we’ve moved the Lead 1 resistance from the bottom right of the bridge to the top right, while the Lead 2 resistance remains in the bottom right. Consequently, the bridge will cancel out any resistance changes due to temperature, as described in an earlier article. Of course, the sense lead has some resistance, but it’s negligible compared to the voltmeter’s impedance.

Reference

[1] Strain Gauges: Selecting an Excitation Voltage, Siemens

Related EEWorld Online content

Defining and measuring strain: part 1
The basics of Kelvin connections
Making sense of test circuits with Kirchhoff’s laws: part 4
What’s the difference between 2-, 3-, & 4-wire RDT sensing?
Stress & Strain, Part 1: fundamental principles
How to avoid errors in low-voltage measurements

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In this series, we’ve been looking at the strain gauge, a type of sensor that can measure how a test specimen deforms as a function of applied stress. In part 2 of this series and in an earlier article on the Wheatstone bridge, we looked at how a single active strain-gauge element coupled with a passive element in a half-bridge configuration can measure tensile and compressive axial strain while providing temperature compensation. In this article, we’ll look at the use of two active strain-gauge elements.

Q: Could we first review the basic Wheatstone bridge topology?
A: Yes, and in fact, this is a good time to update one of our labels. Figure 1 shows the basic Wheatstone-bridge circuit. So far, our unknown resistance of interest has been R_X on the lower right. With multiple active strain-gauge elements, however, we can have multiple unknown resistances in our bridge, so I’ll rename R_X as R₄ to emphasize that it’s not the only potential unknown. Note also that V_EX is the excitation voltage, and V_O is the output voltage, with V_O varying with strain e.

Q: So how do we make use of multiple strain-gauge elements?
A: We concluded part 2 of this series with a figure similar to Figure 2a, with strain gauges mounted on the top and bottom of a test specimen, and with the long sense conductors aligned in parallel with the direction of axial strain. Both will increase in resistance when in tension and decrease in compression. This, in and of itself, can be useful, but first let’s look at a slightly different configuration. In Figure 2b, our test specimen becomes a cantilever subjected to a bending stress, as shown by the red arrow. Here, the top strain-gauge element of length l increases in length by Dl, and the bottom element decreases in length by the same amount.

Q: How do we connect this to our bridge?
A: Figure 3 shows one possibility. Let’s say the fixed resistors and the unstrained resistance values of our strain gauges are all equal to R. We can see immediately that the voltage at the negative terminal of our V_O meter is V_EX/2, and the stressed resistances of the R₂ and R₄ strain gauges are R–DR and R+DR, respectively. Note that this configuration retains the temperature-compensation characteristic of our previous implementation with one active and one dummy element.

We can then calculate V_O:

From part 1, we know that

And therefore:

We can now substitute R(GF)e  for DR in our equation for V_O:

The Figure 4 trace in blue plots V_O vs e  for the Figure 3 circuit. The red trace shows the relationship for our circuit from part 2 with the single active element. Note that for a given e, the voltage for the two-active-element circuit approximately doubles, boosting measurement sensitivity.

Q: Why say “approximately doubles”?
A: The equation above for V_O for two active elements is clearly linear, and the equation for the single-element version is nearly linear over the ranges of strain found in many applications, especially ones with metallic test specimens. However, if we look at a wider range of strain, as shown in Figure 5, we see that the single-active-element response is actually nonlinear—the dashed trace is a straight line connecting the end points of the red trace. Because of the nonlinearity of one of our traces, it wouldn’t be accurate to say the blue trace voltage is exactly double the red trace voltage for any value of strain.

Q: What else should we know about strain gauges?
A: We will conclude this series with a look at full-bridge implementations as well as considerations regarding excitation voltage levels.

Related EEWorld Online content

Sensors expert talks data acquisition, IoT, wearables, and AI
Stress & Strain, Part 2: Implications for electronics
Wheatstone bridge, Part 2: Additional considerations
Defining and measuring strain: part 1
Making sense of test circuits with Kirchhoff’s laws: part 1
Quantifying and measuring non-electrical phenomena: Heat

The post Defining and measuring strain: part 3 appeared first on Test & Measurement Tips.

Welcome to the April edition of Engineering Deep Dive — a curated selection of the most engaging technical threads from the Electro-Tech-Online community’s Electronic Projects Design/Ideas/Reviews category. Questions are selected based on view counts, reply depth, and educational value. Each entry below has been expanded with context, key concepts, and suggestions for further exploration.

Questions at a glance

Q1  Why Does My Window Comparator Output Stay ON at 0V Input Despite Correct Threshold Voltages?
Q2 Does Cable Velocity Factor Affect Pulse Propagation Delay or Only Phase Shift?
Q3  How Does Pulse Ignition and Flame Rectification Work in Gas Water Heater Ignition Systems?
Q4  How Do You Identify Whether a PCB Failure Is Caused by Design Errors or Manufacturing Defects?
Q5  Why Is Reflection Cancellation Still Considered Resonance in Time-Domain Analysis?
Q6  How Can You Measure and Isolate PCB Trace S-Parameters Without RF Connectors?
Q7  What Is the Most Convenient Way to Implement Real-Time Audio FFT Analysis?

Q1  Why does my window comparator output stay on at 0V input despite correct threshold voltages?

A window comparator circuit based on the LM339 is behaving unexpectedly — the output LED stays ON even when the input voltage is 0V, despite threshold voltages appearing correct. This thread walks through threshold calculation, LM339 open-collector output behavior, and systematic troubleshooting of component failures that cause latched outputs.

Key technical topics covered

• Window comparator design with LM339 (open-collector output stage)
• Upper and lower threshold voltage calculation
• Diagnosing latched or stuck outputs caused by faulty components
• Pull-up resistor selection and LED drive circuit

Why It Matters | Window comparators appear in battery monitors, motor-speed controllers, temperature alarms, and ADC over-range detectors. Misunderstanding open-collector outputs is one of the most common LM339 pitfalls for beginners.

Topic tags

Circuit Design | LM339 | Comparator | Troubleshooting

Supporting data: Circuit schematic image

Community thread: Thread link

Q2  Does Cable velocity factor affect pulse propagation delay or only phase shift?

When feeding a digital pulse into a coaxial or transmission-line cable, does the cable’s velocity factor (VF) introduce a propagation delay, or does it only shift the phase of a sinusoidal signal? This thread distinguishes group velocity (relevant to pulse delay) from phase velocity (relevant to sinusoidal phase shift) and explains why both are numerically identical in non-dispersive media.

Key technical topics covered

• Velocity factor and its physical origin (permittivity of the dielectric)
• Group velocity vs. phase velocity — when they differ
• Propagation delay calculation: t_d = length / (VF × c)
• Practical impact on digital timing in long cable runs

Why It Matters | Signal integrity engineers designing high-speed serial links, RF engineers building phased arrays, and hobbyists working with long cable runs all need to understand how VF affects their signals.

Topic tags

Signal Integrity | Transmission Line | RF | Digital Timing

Supporting data: Oscilloscope waveform image

Community thread: Thread link

Q3  How Does Pulse Ignition and Flame Rectification Work in Gas Water Heater Ignition Systems?

This thread goes beyond simple ignition spark generation to explore how a flame rectification sensor confirms combustion. The discussion covers the ionization current produced by a gas flame, how it is used as a half-wave rectifier in the safety circuit, and the risks of DIY modifications to gas appliance electronics.

Key technical topics covered

• High-voltage spark generation via a pulse ignition module
• Flame rectification: DC bias through an ionized gas column
• Safety interlocks and why repeated ignition failures must not be bypassed
• Troubleshooting the sensor electrode (fouling, misalignment, cracked ceramic)

Why It Matters | Gas appliance faults can be dangerous. Understanding the intended safety logic helps technicians and advanced hobbyists diagnose faults responsibly, without disabling protective interlocks.

Topic tags

Power Electronics | Safety Systems | Sensors | Gas Ignition

Supporting data: Module photograph

Community thread: Thread link

Q4  How do you identify whether a PCB failure is caused by design errors or manufacturing defects?

When a PCB batch fails, the root cause might be in the Gerber/drill files or in the fabrication process itself. This thread provides a structured methodology: cross-referencing design files with fab specifications, identifying tell-tale defect signatures (trace opens, plating voids, layer misregistration), and communicating findings to the PCB house.

Key technical topics covered

• DFM (Design for Manufacturability) review checklist before ordering
• Common fab defects: plating voids, trace opens, drill inaccuracies, solder-mask misalignment
• Layer misregistration detection via cross-section or X-ray inspection
• How to document and report defects to get boards replaced or credited

Why It Matters | PCB fabrication failures are costly in both money and schedule. A systematic approach reduces finger-pointing between design and manufacturing teams and speeds up root-cause resolution.

Topic tags

PCB Design | DFM | Manufacturing | Quality Assurance

Supporting data: PCB microscopy image

Community thread: Thread Link

Q5  Why Is Reflection Cancellation Still Considered Resonance in Time-Domain Analysis?

Resonance is traditionally taught in the frequency domain as a sharp peak at a natural frequency. This thread unpacks the conceptual bridge to the time domain: how delayed reflections, constructive/destructive interference, and oscillating energy exchange between inductance and capacitance all manifest as what we still call ‘resonance’, regardless of domain.

Key technical topics covered

• Time-domain view of resonance: energy oscillating between L and C
• Reflections on transmission lines and how they create standing waves
• Fourier duality: why time-domain oscillation maps to a frequency-domain peak
• Practical examples: stub resonance, via resonance in PCBs

Why It Matters | Signal integrity engineers and RF designers often switch between domains. Understanding why the same physical phenomenon appears as both a time-domain ringing and a frequency-domain peak prevents analysis errors.

Topic tags

Signal Integrity | RF Theory | Frequency Domain | Time Domain

Supporting data: Simulation waveform image

Community thread: Thread Link

Q6  How can you measure and isolate PCB trace S-parameters without RF connectors?

Characterizing a PCB interconnect with a VNA is straightforward when SMA connectors are available — but what if there are none? This thread covers probe-based measurement, the de-embedding process to remove fixture and pad parasitics, and the 2x-Thru method for extracting a single trace’s S-parameters from a back-to-back structure.

Key technical topics covered

• Probe landing and transition de-embedding concepts
• 2x-Thru and Short-Open-Load-Through (SOLT) calibration strategies
• Reducing fixture discontinuities with careful pad geometry
• Software tools: IDEM, OpenDEKit, or VNA manufacturer utilities

Why It Matters | As PCB speeds push into multi-GHz territory, accurate S-parameter extraction without connectors is essential for channel simulation, equalizer design, and compliance testing.

Topic tags

RF Measurement | S-Parameters | PCB | Signal Integrity

Supporting data: VNA measurement image

Community thread: Thread Link

Q7  What is the most convenient way to implement real-time audio FFT analysis?

Real-time FFT analysis of audio turns a time-domain waveform into a live frequency spectrum. This thread compares hardware (dedicated FFT modules, FPGA), microcontroller (ARM CMSIS-DSP, ESP32 FFT), and PC-based (Python, MATLAB) approaches, weighing latency, cost, and complexity.

Key technical topics covered

• FFT fundamentals: window functions, bin resolution, sample rate requirements
• Microcontroller options: ARM CMSIS-DSP library, ESP32 FFT example
• Dedicated modules: MSGEQ7 7-band analyzer IC, OpenMusicLabs FHT
• PC/software approaches: Python (numpy.fft), MATLAB, Audacity spectrum view

Why It Matters | Audio FFT is used in music visualizers, hearing aid design, acoustic testing, and voice-command pre-processing. Choosing the right platform depends on the required resolution, update rate, and available hardware.

Topic tags: Audio DSP | FFT | Embedded Systems | Signal Processing

Supporting data: N/A — community discussion

Community thread: Thread Link

Join the conversation

If any of these questions sparked an idea or you have hands-on experience with a related problem, jump into the thread — the community benefits most when engineers at all levels contribute. You can also start your own question in the Electronic Projects Design/Ideas/Reviews category on Electro-Tech-Online.

Browse all categories: electro-tech-online.com

The post Engineering deep dive-monthly forum highlights April edition appeared first on Test & Measurement Tips.

The Arbitrary Waveform Generators from Spectrum Instrumentation now include a new Sequence Restart Mode for the 65xx and 66xx series, allowing the full sequence of looped and linked waveforms to restart automatically on a trigger event with fixed trigger-to-output timing. The mode is intended for automated test environments where repeatable sequence control can help reduce test time and improve measurement efficiency. The feature is available at no additional charge through the latest driver installation and is supported on Windows and Linux, with programming examples for Python, MATLAB, C++, and LabVIEW as well as a high-level Python API.

The post New AWG mode restarts sequences on trigger appeared first on Test & Measurement Tips.

The update expands NI Nigel AI with prompt-based code generation in the LabVIEW+ Suite and extends AI-assisted capabilities across FlexLogger, InstrumentStudio, TestStand and SystemLink. The tools are designed to support engineers throughout the test lifecycle, offering context-aware suggestions for development, debugging, validation and system reuse.

According to Emerson, the AI features are built specifically for test engineering environments, where traceability, repeatability and system visibility are required. Engineers retain control over generated code and workflows while using AI to reduce manual effort.

The NI platform combines modular instrumentation hardware with open software and a shared data framework, enabling teams to manage diverse signal types, scale test systems and reuse data across projects and locations.

In internal use, Emerson reports reductions in some test development and troubleshooting tasks from hours or days to minutes. Availability of the new capabilities is expected later in 2026.

The post Emerson adds AI-assisted code generation and workflow support to NI test platform appeared first on Test & Measurement Tips.

Emerson announced updates to its NI test software portfolio, adding AI-assisted features designed to improve test development efficiency and system integration. The NI Nigel AI technology will expand to include prompt-based code generation in LabVIEW+ and broader support across tools such as FlexLogger, InstrumentStudio, TestStand, and SystemLink.

The updates are intended to help engineers develop, validate, and deploy tests more quickly while maintaining visibility into system behavior. Nigel AI is designed for test environments, providing context-aware suggestions across the workflow, from development and code reuse to validation and deployment.

The NI platform combines modular hardware, open software, and a shared data framework to support complex test requirements. Engineers can configure systems to handle a range of signals, manage large data sets, and integrate evolving computing technologies over time.

According to internal testing, AI-assisted workflows reduced some development and troubleshooting tasks from hours or days to minutes. The platform is used in industries including aerospace, semiconductor, and transportation, where reliability, traceability, and performance are critical.

The post Emerson introduces AI capabilities for test automation at NI Connect 2026 appeared first on Test & Measurement Tips.

In Part 1 of this series, I discussed how to use near-field probes to help identify high-frequency harmonic energy sources on PC boards. Characterizing each energy source, such as processors, memory, and power conversion, helps identify the source of coupled EMI being radiated from your product. This is “Step 1” of my three-step process for troubleshooting radiated emissions issues and has been pretty well documented in other articles and industry application notes.

What has not been as well covered is my “Step 2”, how to use RF current probes to further characterize attached cable harmonic currents and reveal “red flags” in your designs. The ability to identify potential coupling mechanisms from observing harmonic energy on attached cables is an important part of the troubleshooting process. In Part 2 of this series, I’ll show how I use one of my most important tools for troubleshooting radiated emissions issues.

RF current probes

I suspect most product designers are familiar with the smaller current probes designed for oscilloscopes or digital multimeters (DMMs). These typically have smaller apertures that fit a wire or small cable and generally extend from DC to 100 MHz, at best. There are also current probes for electrical measurements with larger apertures that range up to only a few MHz and are really designed for mains frequencies.

RF current probes, on the other hand, are designed to measure microamps of RF current from kHz to hundreds of MHz flowing on I/O and power cables. They usually have a hinged aperture that can accept everything from a single wire to large-diameter cables (Figure 1). When their 50Ω port is connected to a spectrum analyzer, you’ll observe an RF spectrum similar to that when using a near-field probe. Many manufacturers make these probes, but for this article, we’ll use the affordable Tekbox Model TBCP2-750 ($879). See [2].

Cable radiation is usually the dominant reason for radiated emission failures. The problem is that the various harmonic energy sources measured on your circuit boards or system can couple to any attached cables, resulting in high-frequency harmonic currents that now create an antenna effect, causing radiated emissions.

We’ll use the RF current probe to characterize and reduce these coupled RF currents by clamping it around each I/O and power cable in turn and recording the spectral characteristics for each. The typical RF current probe is sensitive enough to measure µA of RF current, and only 6 to 8 µA of harmonic current can fail the FCC class B limit.

If cables are the dominant cause of emission failure, then I often leave the probe connected and fixed in place while I’m trying various mitigations back in the PC board or system. Observing changes in real time is quite efficient!

RF current probe measurements

The RF current probe is merely a current transformer that measures RF currents in the primary (wire or cable to be measured) and couples that to the secondary, which is loaded by the 50Ω input impedance of the spectrum analyzer (Figure 2). This produces a voltage across 50Ω that is usually in terms of dBµV. Switching the MXO38 spectrum display from the default dBm to dBµV allows us to calculate and directly compare to the emissions limits, as you’ll see further on.

I usually insert a bit of “bubble wrap” within the probe aperture to keep the wire or cable centered and away from the metal probe case in order to minimize measurement errors.

Because of resonances on cables, it’s best to slide the RF current probe back and forth on the cable or wire in order to maximize the dominant harmonic or harmonics. Once the larger harmonics are maximized, I tape the probe down to the table to minimize variables while trying different mitigations to reduce cable coupling from the board.

Mitigations could include rerouting internal cables, improved bonding of cable shields to chassis or digital return plane, adding or improving common mode filtering at the I/O or power connectors, shielding energy sources using local shields, etc.

Caution

RF current probes are so sensitive that when clamped around a wire or cable, that cable tends to act as a receiving antenna. Ambient transmissions, like AM/FM broadcast stations, two-way communications, digital TV, and cellular phones, can confuse measurements from the equipment under test (EUT). It is important to be aware of certain bands in the RF spectrum where these services reside, so they can be differentiated from product emissions.

To ease this confusion, I usually clamp the current probe around the cable under test with the EUT turned off. You’ll like to see at least the FM broadcast stations in the region 88 to 108 MHz. Placing these captured signals in Max Hold mode provides an initial plot of current ambient signals. On the MXO3, this can be saved as a “reference” trace, as you’ll see in the next section.

Making a measurement

We’ll be measuring the common-mode harmonic currents on the power input cable of a 1 MHz GaN buck converter. This converter has a strong 230 MHz ring frequency that I use to demonstrate how ring frequency can manifest as a broad peak in the frequency spectrum. We’ll use a Fair-Rite 43 material snap-on choke to demonstrate a reduction in harmonic currents in the cable. Figure 3 shows the test setup.

As mentioned above, when using the RF current probe to test common mode cable emissions outside of a shielded chamber, I first perform an ambient measurement with the EUT off. This will record and save the specific RF environment so you can better differentiate EUT emissions from ambient signals.

First, connect the current probe to channel 1 (C1) and set the Termination to 50Ω. Leave all other settings to their defaults. Turn on the EUT in order to set up the oscilloscope for an acceptable time domain signal. I usually just press Auto-set at first.

Then turn on the Spectrum mode (Menu > Spectrum), turn On the Display, set the Source to C1, touch Start/Stop and set the frequency limits to Start = zero, Stop = 1 GHz, Normally, the EMC standards have you set the resolution bandwidth to either 100 kHz (military) or 120 kHz (commercial), but I find 1 MHz may have a cleaner averaged signal for troubleshooting purposes. Turn off Auto BW and set RBW for 1 MHz. Press Max Hold to save the ambient plot.

Now, switch the Menu to Apps > Reference and select R1 (reference waveform 1), turn Show = On, and select Source = S1MaxH. Select Create/Update and Save As “Ambient”. This will save the ambient Max Hold plot as reference 1 (R1).

Note that every time you reopen the Reference screen, the Source keeps defaulting to C1, so you’ll need to keep selecting S1MaxH, or it will save a time domain reference for C1. This messed me up several times when trying to add reference waveforms to the spectrum display.

Next, turn on the EUT and select Max Hold, then save this as R2 [2]. This will become the saved “before” spectrum case. You can change the default color by going to Settings > Appearance, Category = Reference, and Color Source = R2. Go ahead and set the color as desired. For this article, I set it to light blue. In a similar fashion, save this as reference 2 (R2) and name it “EUT_On”.

You should end up with a screen capture as in Figure 4 with the ambient in default grey, EUT-On (“before case”) in the color of your choice, and the real-time waveform in the default yellow. From here, you can try various mitigations and compare the resulting spectrum with the saved reference 2 (R2) waveform. The ambient spectrum (R1) may be used to differentiate the ambient RF signals from the “live” signals.

The blue trace is the measurement with the EUT turned on and shows the broad peaks at the ring frequency of 233 MHz and board resonance at 558 MHz. Now, with the saved reference plots, you can commence troubleshooting and applying various mitigations while observing immediate results!

Estimating pass/fail

One important use for the RF current probe is to provide an estimate of passing or failing specific emission test limits. By knowing the dominant currents in an I/O or power cable, we can calculate the E-field at the test distance per the standard used (usually 3m or 10m for commercial products). While this won’t necessarily be precise, it still gives us a “ballpark” estimate to which we can compare with the test limit at that frequency.

Commercial RF current probes come with a calibration chart of transfer impedance versus frequency (Figure 5). Using Ohm’s Law, we can use this chart to calculate the measured common mode current in the wire with respect to the voltage measured at the probe output port, assuming a 50Ω system. This is based on work by Dr. Clayton Paul [3] and further refined by Henry Ott [4]. There are example calculations in [5] and [6].

Let’s assume we measure one of the dominant harmonics in a cable as 28 dBµV at 120 MHz at the spectrum analyzer. We can also read of a transfer impedance of 20 dBΩ at 120 MHz from the calibration chart in Figure 5.

Using Ohm’s Law, we can calculate the common mode current (Icm) in the cable:

Icm (A) = E (V) / R (Ω), or, in converting to terms using log identities,

Icm (dBµA) = Vprobe (dBµV) – 20 dBΩ = 28 – 20 = 8 dBµA

Now using the E-field equation from Paul and Ott:

where

Ec is the calculated E-field in V/m due to common-mode current flowing on the cable,

Ic is the current through the wire or cable (A),

f is the harmonic frequency being measured (Hz),

L is the length of the cable in meters and

d is the measured distance during the compliance testing (usually 3 or 10m).

Converting the measured values to basic units and plugging into the E-field equation, we get 1.26E-4 (V/m). Converting this back to log units, we get 42.03 dBµV/m. Comparing this with the FCC class B limit at 120 MHz (43.5 dBµV/m) indicates we may be just under the limit by only 1.47 dB. Typically, we’d want at least a 6 dB margin, so this is probably not good enough.

To streamline all these calculations, I developed a simple Excel spreadsheet, which may be downloaded from my website [7]. Figure 6 shows an example calculation. By entering the specific probe transfer impedance, the frequency of concern, the cable length, and test distance (typically 3 or 10m), the E-field in dBµV/m is calculated and may be compared to the appropriate test limit.

Summary

The RF current probe is not only a useful tool for general troubleshooting, but may be used to determine potential passing or failing due to harmonic currents on a radiating cable. While they may be a bit pricy, I find the RF current probe is one of my most-used tools for troubleshooting emissions.

Most importantly, if you know that cables are the dominant radiating source, all the troubleshooting can be done in-house with an RF current probe. No need to run back and forth between your facility and the 3rd-party test lab to perform this troubleshooting, saving cost and time!

I also have a short video showing how to use these RF current probes in [8].

With its fast acquisition update rate, the MXO38 spectrum feature works extremely well for general bench-top EMC troubleshooting and debugging, and it’s become one of my favorite bench-top tools. The ease of setting up the analyzer is a step ahead of other leading manufacturers. Once saved, the reference waves stay in memory along with the instrument setup, so if you have to move locations, everything is preserved.

References

[1] Rohde & Schwarz
[2] Tekbox current probes
[3] Paul, Introduction to Electromagnetic Compatibility (2nd Edition), Wiley Interscience, 2006, pages 518-532.
[4] Ott, Electromagnetic Compatibility Engineering, Wiley, 2009, pages 690-693.
[5] Wyatt, Workbench Troubleshooting EMC Emissions (Volume 2), Amazon.
[6] Wyatt, The RF Current Probe: Theory and Application, Interference Technology
[7] Wyatt, E-Field Calculator (scroll down to find it)
[8] Wyatt, Current Probe Demo, Current Probe Demo

The post R&S MXO3 Oscilloscope for EMC measurements: part 2 appeared first on Test & Measurement Tips.

Rohde & Schwarz has released a Pulsar signal simulation option for the R&S SMBV100B and R&S SMW200A vector signal generators, adding support for the LEO satellite navigation service from Xona for receiver development and production testing. The software allows engineers to simulate next-generation positioning, navigation and timing signals alongside existing GNSS services such as GPS, helping validate compatibility as Pulsar deployment scales. The option is intended for manufacturers developing navigation receivers and other PNT-enabled devices that need test coverage for stronger signals with improved resilience to threats and interference.

The post New LEO signal simulation speeds receiver validation appeared first on Test & Measurement Tips.

Q: In Figure 1, it looks like you’ve changed some of the labels.
A: Right. I had been using V_DMM to emphasize that we are using a modern digital multimeter (DMM) instead of a 19th-century galvanometer, which would have been used in early Wheatstone bridges. 19th-century galvanometers were quite useful in detecting zero current and voltage conditions, with applications extending to the decoding of transmitted signals in early trans-Atlantic telegraphy. However, such galvanometers were not good at quantifying non-zero levels, which modern instruments can do easily and accurately. In Figure 1, I’m using labels commonly found in strain-gauge literature. For example, I’ve changed V_DMM to V_O, for output voltage. In addition, I changed V_IN to V_EX, for excitation voltage. Finally, in the setup from our earlier series, V_DMM varied inversely with R_X. In Figure 1, I have reversed the meter polarity, so V_O varies directly with R_X and hence strain e.

To review, in Figure 1, R_X is an active strain gauge, and R₂ is a dummy strain gauge used for temperature compensation. Note that the long, thin wires of R₂ are mounted perpendicular to the direction of tension, so their resistance is relatively unaffected by the strain.

Q: So, how do we calculate strain given V_O?
A: From our previous series, we demonstrated how to calculate R_X based on the voltage reading, with the DMM or output voltage equaling V_V – V_X. For the configuration shown in Figure 3 with the polarity switch,

We can rearrange this equation as follows:

Then we can calculate R_X as a function of V_O:

Q: And how does strain relate to R_X?
A: From part 1 of this series, we know that given a gauge factor GF, strain is

We also see that DR is R_X – R₂, where R₂ equals the unstrained resistance of R_X, or 120 W. So, given V_O and a GF value of 2, we can directly calculate strain:

Figure 2 plots this equation for values of V_O from -10 mV to +10 mV.

Q: Can we design a bridge configuration where both strain-gauge elements are active, and if so, what would be the advantages?
A: Figure 3 shows an alternative arrangement to Figure 1, with one strain gauge depicted in red placed at the top of a test specimen, while another depicted in blue is placed at the bottom, with the sense wires in both aligned with the direction of stress, so both are active.

Next time, we’ll take a look at how to apply this configuration and describe its benefits.

Related EEWorld Online content

Making sense of test circuits with Kirchhoff’s laws: part 4
How to use remote sensing for DC programmable power supplies
Things to know about multimeters
The first undersea transatlantic cable: An audacious project that (eventually) succeeded, Part 2
Wheatstone bridge, Part 1: Principles and basic applications
Stress & Strain, Part 1: fundamental principles

The post Defining and measuring strain: part 2 appeared first on Test & Measurement Tips.

The CP1000 supports up to 1000 A rms continuous and peak currents up to 1400 A. But the specification that generated the most conversation was bandwidth. Most competitive products in this current range top out around 10 kHz. The CP1000 goes from DC to 1.5 MHz, which matters more than it might seem at first glance. Additional specs worth noting: rise time is 235 ns typical, AC noise at 20 MHz bandwidth limit is 10 mA, and the probe offers two output voltage settings at 0.005 V/A and 0.05 V/A with a minimum sensitivity of 100 mA/div. The 6-meter cable gives engineers some practical flexibility when working around large power setups.

The Teledyne LeCroy team explained it this way: when engineers calculate switching losses, they multiply the current and voltage waveforms. If the current probe’s bandwidth is too narrow, it artificially slows the rise time, making losses appear larger than they actually are. As one engineer at the booth put it, “By having a higher bandwidth probe, you can get a more accurate representation of how fast you’re switching, and you can make better efficiency measurements and loss measurements.”

That point connects to a broader issue the team raised about oscilloscope measurements generally: “A lot of people who are doing oscilloscope measurements have this assumption that what I’m seeing on the oscilloscope is what’s happening on my system. We spend a lot of time explaining to people that’s not what you’re seeing in your system. That’s what you’re seeing after it goes through the probe, after it goes through what’s happening on the front end of the oscilloscope. It’s a total system measurement. Engineers don’t care about that. They want to know what’s happening.”

The CP1000 uses Hall effect technology, so there’s no need to break the line, unlike current transformers or Rogowski coils, which remain the workaround most engineers are using today. The maximum conductor diameter the probe can accommodate is 33 mm. It connects via Teledyne LeCroy’s ProBus interface for automatic scaling, degauss, and autozero functions on compatible oscilloscopes, and it’s available now at a list price just under $9,000.

The post Teledyne LeCroy at APEC 2026: the case for higher bandwidth current measurement appeared first on Test & Measurement Tips.

Today’s designs are more complex than ever. Signal frequencies are climbing, package sizes are shrinking, and the range of bus types and standards that engineers need to support continues to expand. A scope that checks the bandwidth box but falls short on noise performance, probing capability, or analysis software can slow you down when it matters most.

The right approach starts with the measurement problem, not the spec sheet. Power integrity work demands low-noise front ends and high vertical resolution. Serial data conformance testing adds requirements around jitter analysis and de-embedding. Embedded system troubleshooting calls for strong trigger systems and protocol decode capability. RF and wireless applications require high-performance FFTs and spectrogram display support.

On April 16, EE World Online is hosting a webinar that tackles exactly this. Industry veteran Chris Loberg will walk through a practical framework for oscilloscope selection across all four application areas, with plenty of time for audience questions.

Do you have questions now you would like us to address during the webinar? Email those to akalnoskas@wtwhmedia.com.

Beyond Bandwidth: How Engineers Should Really Choose an Oscilloscope takes place Thursday, April 16, at 2:00 PM EDT. Register here.

The post Editorial Webinar April 16: Beyond bandwidth: how engineers should really choose an oscilloscope appeared first on Test & Measurement Tips.

PicoRaptor combines high bandwidth with high insertion count, using a single elastomer for consistent contact force while allowing a slight wiping action. Its Advanced Contact Finish (ACF) technology stabilizes contact resistance (Cres), limits solder migration, and extends load board life. PowerRaptor includes a contact wipe for Cres stability and an optional Airtherm integration for rapid, accurate pin temperature conditioning. Both variants are engineered to increase Mean Time Between Assist (MTBA) across demanding test environments.

The post ATE contactors combine kelvin configuration with thermal pin conditioning appeared first on Test & Measurement Tips.

In a previous series, we investigated the Wheatstone-bridge circuit topology and described how strain-gauge elements could be used in the bridge legs.

Q: At that point, I asked the question, what is strain, and what are its units?
A: Right, so we’ll take up that question in this new series. Figure 1 at the top shows a specimen under test of length l. In the center image, we apply an axial stress in the form of tension to the specimen, and it lengthens by an amount Dl. The strain, indicated by a lower-case epsilon, is

In addition, as shown at the bottom, if you apply axial compression to a specimen, it shrinks in length, and you’ll have a negative strain.

Like the radian, strain is a ratio of lengths and is therefore dimensionless. It can be helpful, however, to think of it in units such as meters per meter. You’ll also see strain expressed in microstrain, abbreviated µe, which is 1 millionth of e. If you have a specimen 1 meter long and you apply tension that expands its length by 1 micron, you’ll have a strain of 1 µe.

Q: OK, given that we’ve defined strain, what’s an effective way to measure it?

A: Just as we can use a thermocouple to measure temperature or an accelerometer to measure vibration, we can use a metallic strain gauge to measure strain. The metallic strain gauge consists of a conductive foil pattern on a flexible insulating backing, such as the one shown in Figure 2, with the conductive foil shown in black and the insulating backing, called a carrier, shown in blue. The image on the left shows the strain gauge in an unstrained state, for which it has a resistance R. When a test specimen on which the gauge is mounted undergoes tension, as shown in the center, the vertical conductive elements of the pattern elongate and become thinner, and their resistance increases by an amount DR. Conversely, under compression, as shown on the right, the resistance decreases by DR.

Q: How do we relate strain-gauge resistance changes to test-specimen length changes?
A: Your strain gauge will have a parameter called gauge factor, abbreviated GF, which you will find on the data sheet. GF, usually about 2 for a metallic strain gauge,^[1] relates to gauge resistance and specimen length as follows:

Now we can solve for strain as a function of the resistances and gauge factor:

Q: So we just glue the strain gauge to the specimen, and we are all set.
A: Right, but you’ll need to use a special adhesive, such as cyanoacrylate, methacrylate, or epoxy resin, that can accurately transfer your specimen’s deformation to the strain gauge. Factors that influence which adhesive you use include the strain and temperature ranges.^[2]

Q: How do we measure strain using the Wheatstone bridge?
A: Figure 3 is an alternate view of Figure 2 from part 4 of our previous series. Here, R_X is an active strain gauge, and R₂ is a dummy strain gauge used for temperature compensation. Note that the long, thin wires of R₂ are mounted perpendicular to the direction of tension, so their resistance is unaffected by the strain, and you will not need a special adhesive. However, if you use two different adhesives, you should ensure that their thermal properties are similar.

Q: So how do we calculate strain?
A: We’ll look at the details of the calculation next time, after which we’ll discuss some additional strain-gauge considerations. For example, Figure 3 shows a half-bridge application with one active strain-gauge element, but other configurations are possible. We’ll also look at optimizing the excitation voltage (V_IN in Figure 3). Finally, strain-gauge applications often involve large structures, such as wide-body airframes, antenna towers, stadiums, buildings, or bridges, so we’ll look at lead-resistance and signal conditioning considerations.

References

[1] Measuring Strain with Strain Gages, Emerson
[2] How to Select the Right Adhesive for your Strain Gauge Installation, HBK

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The DDS option adds frequency and amplitude slopes, flexible command sequencing, and real-time signal adaptation without large data transfers or complex waveform calculations. These capabilities address applications including network stimulation, filter and amplifier testing, vibration shaker control for mechanical resonance testing, and simulation of power fault conditions for circuit fault detection. Software support covers Windows and Linux, with programming interfaces for Python, MATLAB, C++, and LabVIEW, plus a no-code DDS CONTROL GUI for direct signal generation. All products include lifetime technical support from Spectrum Instrumentation’s engineering team and free software and firmware updates.

Link to video

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Moku:Delta is a meaningful hardware upgrade over its predecessor, featuring eight channels instead of four, a larger FPGA, improved analog input noise performance, and QSFP ports that support high-speed data streaming up to 80 Gbps. The device runs eight instruments simultaneously at 2 GHz, consolidating significant test-and-measurement capability into a single unit.

Generative Instrumentation is an AI-powered tool that writes HDL/Verilog code on behalf of the engineer, removing the steep learning curve traditionally associated with custom FPGA instrument development. Engineers describe what they need, the tool generates and compiles the code, and the entire process remains fully auditable in real time. Auto-generated plain-language documentation accompanies each build. Public release is slated for later this spring.

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At first glimpse, you may not realize the common thread in the three sketches in Figure 1. The left (a) shows a transmission-line symbol. The middle (b) shows a fraction of a schematic, potentially depicting two components on a power-distribution network (PDN). The right (c) shows a satellite communications example. Basic physics tells us that you can view all three in the context of characteristic impedance and propagation delay.

Signal Integrity Perspective

SI engineers are familiar with the fundamental equations that describe the Z₀ characteristic impedance and t_pd propagation delay of uniform interconnects. If we ignore losses, these equations became a function of per-unit-length inductance (L) and per-unit-length capacitance (C) of the transmission line.

The center schematic snippet may represent the simplified impedance of a DC source (R₁ and L) and a bulk capacitor (C and R₂) in the frequency range where the DC source becomes inductive, and the bulk capacitor’s impedance flattens out. PI engineers would know that if we want a smooth transition between the two impedances, we need to ensure the following:

We also know that the f_c resonance frequency between the capacitor and inductor is:

Though this is just a one-port lumped circuit, we can notice that (3) and (4) have essentially the same form of expressions as (1) and (2).

We may also wonder what is the corresponding item in Figure 1a to R₁ and R₂ in Figure 1b? Figures 2 and 3 explain and illustrate the connection. In short, when R₁ and R₂ in (3) are the same, they represent what we may call the lumped characteristic impedance of this circuit, making the impedance of the circuit frequency independent, just as terminating a lossless transmission line with its characteristic impedance makes its input impedance frequency independent.

Figure 2 shows the input impedance magnitude of a lossless 50-Ω transmission line with 2.5 ns propagation delay with different values of load resistance (R_load). At low frequencies, where the transmission line is electrically very short, the input impedance magnitude equals the load resistance. At higher frequencies, we notice the familiar periodic fluctuation of the input impedance. Notice the logarithmic frequency scale that compresses the sinusoidal variation as a function of frequency. As the load resistance approaches the characteristic impedance, the impedance peaks and valleys come closer and eventually, when the load resistance equals the characteristic impedance, the curve becomes a straight line. This shows what SI engineers know well: terminating the transmission line with its characteristic impedance eliminates reflections over a wide range of frequencies.

Power Integrity Perspective

Figure 3 illustrates how this relates to PI. We use the schematics from Figure 1 to look at the combined impedance of the parallel-connected R₁–L and C–R₂ network. The component values on the left illustrate potentially a medium-power DC source (this is the R₁–L leg) interacting with the bulk capacitor, represented by the C–R₂ leg. We vary only one component value, R₂, which in this example represents the equivalent series resistance (ESR) of the capacitor.

On the impedance plot in Figure 3, the black line represents the R1–L leg, the red lines show the impedance of the C–R₂ leg for the three C–R₂ values (short dashed line: R₂_max, solid line R₂_nom, long-dashed line: R₂_min) and the blue lines represent the combined impedance magnitude for the three R₂ values. We can see that when R₁ equals R₂ (and it also equals the

L/C = 10 mΩ value), the impedance magnitude plot becomes frequency independent, similar to the input impedance of a matched-terminated lossless transmission line. Another similarity is that pushing R₂ lower than the optimum value required for flatness actually creates an impedance peak at the LC resonance frequency.

To make the connection between the SI and PI illustrations in terms of cutoff frequency and propagation delay as well. Figure 4 and Figure 5 compare the frequency-domain and time-domain behavior of two circuits: a transmission line with Z₀ = 10 mΩ and t_pd = 1 µs propagation delay. For this, take the SI illustration circuit from Figure 2 and rerun that simulation with the characteristic impedance, LC values, and relative load impedance steps around 10 mΩ nominal value. Though a transmission line with 10 mΩ characteristic impedance is not practical for our typical signaling tasks, it matches the component values in our PI example in Figure 3.

Figure 5 shows the result of using extreme termination and looking at the response with fast step excitations. We take the transmission lines from Figure 2 and the equivalent transmission line from Figure 4 and apply a fast voltage source with a 0 V to 1 V swing.

Both waveforms show a damped periodic square-wave ringing, where the period of the ringing equals four times the propagation delay: 10 ns for the 50-Ω transmission line and 4 µs for the transmission line approximating the power circuit. (The 4x multiplier comes from the well-known quarter-wave resonator structure, since we have low impedance at one end and high impedance termination at the other end.)

Another way to view these circuits is to stay with the lumped circuit equivalent of the power circuit. The 10 nH inductance and 100 µF capacitor could be considered as the inductance and capacitance of a single-lump LC approximation of a transmission line. This leads us to the schematics shown on the left of Figure 6. We use the same source and load conditions we used on Figure 5: 1 mΩ source resistance and 100 mΩ load resistance.

From these figures, we can summarize that the lowest resonance frequency in a distributed transmission line occurs at the quarter-wave resonance:

The resonance frequency of the lumped LC circuit from Figure 6 was given in (4). Though the constant in the formula is slightly different, both expressions rely on the square root of the LC product.

EMC Perspective

Recall that Figure 1 includes an EMC case, where electromagnetic waves travel through a dielectric medium, most often through free space. In free space, we cannot speak about the medium’s capacitance and inductance. Instead, we can look at the permittivity and permeability material constants, which are proportional to capacitance and inductance when conductors form terminals. The permittivity of free space is ε₀ = 8.85 pF/m, and the permeability of free space is µ₀ = 4π × 10^-7 H/m. If we substitute L and C with these material constants and units, we get the very familiar results: the 120π= 377 Ω impedance of free space (in the far field) and the inverse of the speed of light: c = 3 × 10⁸ m/s.

Summary

From these examples, I’ve shown how the three disciplines — SI, PI, and EMC — are related. Even though they were introduced at different times and were motivated by seemingly different practical concerns, they share the same roots. PI engineers tend not to think about reflections during the design of lumped power distribution circuits. SI people tend to think of interconnects as conductor-bound distributed transmission lines, even though their behavior can also be described using lumped expressions. EMC people consider electromagnetic waves bouncing around in space. You can see how propagating waves connect transmission lines and lumped circuits through basic formulas. Once you understand their common roots, you can appreciate that distance along the signal propagation comes with finite delay and can be associated with inductance, no matter which discipline you look at. You can also see that the lumped equivalent circuit of a power distribution network can relate to reflections, commonly used in the context of transmission lines. Knowing these common roots helps you make more effective and better designs.

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R&S sent me a model MXO38, the eight-channel version with built-in 50 MHz single-channel AWG and two optional 8-bit digital channels. At just 15-in. wide x 9-in. tall x 6-in. deep, the MXO3-series is scaled down to fit most lab benches [1]. I’ve been using their larger MXO4 for a few months, so I was immediately familiar with the user interface, which is laid out intuitively and just the same. The MXO4-series is an inch wider and an inch taller with the same 6″ depth. The smaller MXO3 just seems more at home on my test bench. The base price for the four-channel model is $5,890, while the base price for the eight-channel model is $13,800.

As Figure 1 shows, R&S reduced the size when designing the MXO-series compared to the MXO4, yet kept most of the basic specs. The 1 mV low-noise vertical sensitivity, 12-bit resolution, 500 Msamples memory depth, and 21 ns trigger re-arm allow a terrific FFT spectrum display. The waveform capture is 4.5 million waveforms per second, providing real-time capture of up to 99%. From an EMC perspective, this provides a nearly real-time spectrum capture.

The 11.6-in. full HD touch screen provides plenty of room for multiple waveforms. For this EMC engineer, a big advantage is the ability to display up to four independent spectral displays simultaneously.

The MXO3 menu system is straightforward, and the keyboard is well laid out. Top to bottom, the front panel includes triggering, horizontal, vertical, and various modes: Autoset, Preset, cursor control, and screen capture buttons (Figure 2).

In use, I found the default backlight was too bright for my office lab. This can be adjusted via Menu > Settings > Display > then touch the Backlight button and use the universal control knob (lower panel) to set the level.

Ports located on the rear include the usual communications connections, plus an HDMI port for connecting an external monitor (Figure 3). You’ll also find Trigger In and Trigger Out connectors, as well as the 50 MHz arbitrary waveform generator (AWG) RF output. The AWG is controlled from the front and can be set to several normal waveforms, including sample ECG and EEG.

Figure 4 shows the two 8-channel digital input ports. Being an EMC engineer, I did not test the digital ports. Attaching the cables could potentially pull on the connectors, causing reliability issues.

Setting up the MXO38 for EMC measurements

When I’m looking for clock harmonics on a PCB or cable, I’ll set the vertical sensitivity to the most sensitive: 1 mV/division. For larger signals such as power conversion circuits, I’ll use the Autoset button to get me in the general range, then adjust trigger level, horizontal, and vertical adjustments to obtain the best viewable display.

Next, I turn on the spectrum analyzer by either selecting Menu > Spectrum or simply pressing the “Spec” button in the vertical section of the front panel. You’ll obtain the menu selections shown in Figure 5.

I usually prefer to set the Start and Stop frequencies manually, so press Start/Stop, then double-tap the Start and manually enter the desired start frequency. I usually leave this at the default “zero”. Repeat this by double-tapping the Stop button and manually setting the frequency. Sometimes, I will leave this at the default 1 GHz if I wish to see the whole spectrum.

Next, I like to turn off Auto resolution bandwidth (RBW), so I can manually set it to 100 kHz or 120 kHz (MIL or commercial, respectively), depending on what the appropriate standard requires for frequencies below 1 GHz.

I set the Window Type to the default Blackman-Harris setting. Note that there are four capture modes: Normal, Min Hold, Max Hold, and Average. For general probing, you should leave this set to Normal. For intermittent or broadband signals, I often use Max Hold to capture the signal envelopes for a few seconds.

The last steps include pressing Scale and choosing dBµV from the default dBm. Usually, I didn’t have to change the default scale factors, but you can change them as needed. This setup was far easier than I’ve seen on other manufacturers’ oscilloscopes.

The spectrum analyzer menu includes a feature that automatically adds peak markers (Peak List; Figure 6) to spectral peaks. These labels include frequency and amplitude for each peak, as you’ll see later. You can add annotations (text, arrows, boxes, etc.) to the display. The oscilloscope stores screen captures in its internal memory or on a USB thumb drive, if installed. You can save and recall setups as well.

EMC measurements

Let’s try it out on some basic EMC characterization measurements on an Arduino Due Version R3 embedded processor, as there’s plenty to see. Let’s dive in!

This older model Arduino Due R3 is ideal for testing because it includes a conventional DC-DC converter with a switch inductor easily accessible (Figure 7). It is based on an Atmel ATSAM3X8E ARM Cortex 32-bit processor clocking at 84 MHz (12 MHz clock oscillator). The DC-DC converter runs at about 500 kHz and USB at 16 MHz. We’ll see all those signals clearly. I had to order mine from eBay, as the newest versions of the board do not include the same type of power conversion circuit.

Let’s make some real measurements. A proper characterization of a product for radiated emissions requires three steps. I discuss this in detail in volume 2 of my EMC Troubleshooting Trilogy books [2]. We’ll demonstrate the first step in this article.

1. Near-field probing to characterize the spectral response for all high-energy components
2. Characterization of high-frequency cable harmonic currents for all attached cables
3. Measurements a short distance away (~1 m) to confirm the harmonics that actually radiate

I used an R&S H-field probe (1 cm loop) for all the measurement points (Figure 8). Near-field probing of a PC board or cable should always be the first step. We want to identify each high-energy component and document its spectral characteristics.

A loop probe couples nicely to the switch inductor of the DC-DC buck converter, which allows us to observe many important waveforms and spectral displays (Figure 9).

Using cursors and measuring the distance between switched waveforms confirms the converter is switching at a little over 500 kHz. There is ringing due to operating in discontinuous conduction mode (DCM). This ringing will manifest as a broad peak at the ring frequency, in this case, 4.2 MHz. You can observe this peak in Figure 10. I describe this concept in more detail in [3]. We can observe that each frequency spike occurs at 500 kHz intervals.

Now, let’s look at the 12-MHz crystal oscillator, shown in Figure 11. Probing near the oscillator confirms the 12-MHz harmonics. Here, we’ve increased the upper frequency limit to 500 MHz.

Note the use of Peak List to identify major harmonics. The adjustments, Max Results, Threshold, and Peak Excursion can be varied to automatically label just the most important peaks. Note that the delta difference in adjacent peaks in Figure 12 is exactly 12 MHz and that some harmonics extend out to 500 MHz, or more.

Next, let’s measure the 16-MHz USB clock.

Again, I coupled the probe to the USB IC and its 16-MHz clock oscillator in Figure 13. Note that the delta difference in adjacent peaks is exactly 12 MHz and that some harmonics extend out to 400 MHz, shown in Figure 14.

Summary

I found the MXO3 oscilloscope is very useful for general EMC debugging and troubleshooting. The ability to observe both the time domain and frequency domain with its fast capture is a real advantage. Up to eight waveforms may be stored for comparison to the measured waveform. This will greatly help in comparing “before and after” measurements as various mitigations are tried.

Part 2 of this series will discuss how to measure high-frequency currents on cables. Because cables are the most frequent contributor to radiated emissions, characterizing these harmonic currents is really one of the most important steps in mitigating emissions. We’ll also demonstrate how to connect a simple antenna to the scope to measure the relative strength of the actual emissions.

References

[1] Rohde & Schwarz, https://www.rohde-schwarz.com/us/home_48230.html
[2] Wyatt, EMC Troubleshooting Trilogy, https://www.amazon.com/stores/Kenneth-Wyatt/author/B00SNQ1LJ2
[3] Sensepeek, https://sensepeek.com
[4] Wyatt, Review: PCBite circuit board holder and probe kit, https://www.testandmeasurementtips.com/review-pcbite-circuit-board-holder-and-probe-kit/
[5] Wyatt, Characterize EMI from DC-DC converter ringing, https://www.eeworldonline.com/characterize-emi-from-dc-dc-converter-ringing/

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