At first glimpse, you may not realize the common thread in the three sketches in Figure 1. The left (a) shows a transmission-line symbol. The middle (b) shows a fraction of a schematic, potentially depicting two components on a power-distribution network (PDN). The right (c) shows a satellite communications example. Basic physics tells us that you can view all three in the context of characteristic impedance and propagation delay.

Signal Integrity Perspective

SI engineers are familiar with the fundamental equations that describe the Z₀ characteristic impedance and t_pd propagation delay of uniform interconnects. If we ignore losses, these equations became a function of per-unit-length inductance (L) and per-unit-length capacitance (C) of the transmission line.

The center schematic snippet may represent the simplified impedance of a DC source (R₁ and L) and a bulk capacitor (C and R₂) in the frequency range where the DC source becomes inductive, and the bulk capacitor’s impedance flattens out. PI engineers would know that if we want a smooth transition between the two impedances, we need to ensure the following:

We also know that the f_c resonance frequency between the capacitor and inductor is:

Though this is just a one-port lumped circuit, we can notice that (3) and (4) have essentially the same form of expressions as (1) and (2).

We may also wonder what is the corresponding item in Figure 1a to R₁ and R₂ in Figure 1b? Figures 2 and 3 explain and illustrate the connection. In short, when R₁ and R₂ in (3) are the same, they represent what we may call the lumped characteristic impedance of this circuit, making the impedance of the circuit frequency independent, just as terminating a lossless transmission line with its characteristic impedance makes its input impedance frequency independent.

Figure 2 shows the input impedance magnitude of a lossless 50-Ω transmission line with 2.5 ns propagation delay with different values of load resistance (R_load). At low frequencies, where the transmission line is electrically very short, the input impedance magnitude equals the load resistance. At higher frequencies, we notice the familiar periodic fluctuation of the input impedance. Notice the logarithmic frequency scale that compresses the sinusoidal variation as a function of frequency. As the load resistance approaches the characteristic impedance, the impedance peaks and valleys come closer and eventually, when the load resistance equals the characteristic impedance, the curve becomes a straight line. This shows what SI engineers know well: terminating the transmission line with its characteristic impedance eliminates reflections over a wide range of frequencies.

Power Integrity Perspective

Figure 3 illustrates how this relates to PI. We use the schematics from Figure 1 to look at the combined impedance of the parallel-connected R₁–L and C–R₂ network. The component values on the left illustrate potentially a medium-power DC source (this is the R₁–L leg) interacting with the bulk capacitor, represented by the C–R₂ leg. We vary only one component value, R₂, which in this example represents the equivalent series resistance (ESR) of the capacitor.

On the impedance plot in Figure 3, the black line represents the R1–L leg, the red lines show the impedance of the C–R₂ leg for the three C–R₂ values (short dashed line: R₂_max, solid line R₂_nom, long-dashed line: R₂_min) and the blue lines represent the combined impedance magnitude for the three R₂ values. We can see that when R₁ equals R₂ (and it also equals the

L/C = 10 mΩ value), the impedance magnitude plot becomes frequency independent, similar to the input impedance of a matched-terminated lossless transmission line. Another similarity is that pushing R₂ lower than the optimum value required for flatness actually creates an impedance peak at the LC resonance frequency.

To make the connection between the SI and PI illustrations in terms of cutoff frequency and propagation delay as well. Figure 4 and Figure 5 compare the frequency-domain and time-domain behavior of two circuits: a transmission line with Z₀ = 10 mΩ and t_pd = 1 µs propagation delay. For this, take the SI illustration circuit from Figure 2 and rerun that simulation with the characteristic impedance, LC values, and relative load impedance steps around 10 mΩ nominal value. Though a transmission line with 10 mΩ characteristic impedance is not practical for our typical signaling tasks, it matches the component values in our PI example in Figure 3.

Figure 5 shows the result of using extreme termination and looking at the response with fast step excitations. We take the transmission lines from Figure 2 and the equivalent transmission line from Figure 4 and apply a fast voltage source with a 0 V to 1 V swing.

Both waveforms show a damped periodic square-wave ringing, where the period of the ringing equals four times the propagation delay: 10 ns for the 50-Ω transmission line and 4 µs for the transmission line approximating the power circuit. (The 4x multiplier comes from the well-known quarter-wave resonator structure, since we have low impedance at one end and high impedance termination at the other end.)

Another way to view these circuits is to stay with the lumped circuit equivalent of the power circuit. The 10 nH inductance and 100 µF capacitor could be considered as the inductance and capacitance of a single-lump LC approximation of a transmission line. This leads us to the schematics shown on the left of Figure 6. We use the same source and load conditions we used on Figure 5: 1 mΩ source resistance and 100 mΩ load resistance.

From these figures, we can summarize that the lowest resonance frequency in a distributed transmission line occurs at the quarter-wave resonance:

The resonance resonant? frequency of the lumped LC circuit from Figure 6 was given in (4). Though the constant in the formula is slightly different, both expressions rely on the square root of the LC product.

EMC Perspective

Recall that Figure 1 includes an EMC case, where electromagnetic waves travel through a dielectric medium, most often through free space. In free space, we cannot speak about the medium’s capacitance and inductance. Instead, we can look at the permittivity and permeability material constants, which are proportional to capacitance and inductance when conductors form terminals. The permittivity of free space is ε₀ = 8.85 pF/m, and the permeability of free space is µ₀ = 4π × 10^-7 H/m. If we substitute L and C with these material constants and units, we get the very familiar results: the 120π= 377 Ω impedance of free space (in the far field) and the inverse of the speed of light: c = 3 × 10⁸ m/s.

Summary

From these examples, I’ve shown how the three disciplines — SI, PI, and EMC — are related. Even though they were introduced at different times and were motivated by seemingly different practical concerns, they share the same roots. PI engineers tend not to think about reflections during the design of lumped power distribution circuits. SI people tend to think of interconnects as conductor-bound distributed transmission lines, even though their behavior can also be described using lumped expressions. EMC people consider electromagnetic waves bouncing around in space. You can see how propagating waves connect transmission lines and lumped circuits through basic formulas. Once you understand their common roots, you can appreciate that distance along the signal propagation comes with finite delay and can be associated with inductance, no matter which discipline you look at. You can also see that the lumped equivalent circuit of a power distribution network can relate to reflections, commonly used in the context of transmission lines. Knowing these common roots helps you make more effective and better designs.

The post How physics relates signal integrity, power integrity, and EMC appeared first on Test & Measurement Tips.

R&S sent me a model MXO38, the eight-channel version with built-in 50 MHz single-channel AWG and two optional 8-bit digital channels. At just 15-in. wide x 9-in. tall x 6-in. deep, the MXO3-series is scaled down to fit most lab benches [1]. I’ve been using their larger MXO4 for a few months, so I was immediately familiar with the user interface, which is laid out intuitively and just the same. The MXO4-series is an inch wider and an inch taller with the same 6″ depth. The smaller MXO3 just seems more at home on my test bench. The base price for the four-channel model is $5,890, while the base price for the eight-channel model is $13,800.

As Figure 1 shows, R&S reduced the size when designing the MXO-series compared to the MXO4, yet kept most of the basic specs. The 1 mV low-noise vertical sensitivity, 12-bit resolution, 500 Msamples memory depth, and 21 ns trigger re-arm allow a terrific FFT spectrum display. The waveform capture is 4.5 million waveforms per second, providing real-time capture of up to 99%. From an EMC perspective, this provides a nearly real-time spectrum capture.

The 11.6-in. full HD touch screen provides plenty of room for multiple waveforms. For this EMC engineer, a big advantage is the ability to display up to four independent spectral displays simultaneously.

The MXO3 menu system is straightforward, and the keyboard is well laid out. Top to bottom, the front panel includes triggering, horizontal, vertical, and various modes: Autoset, Preset, cursor control, and screen capture buttons (Figure 2).

In use, I found the default backlight was too bright for my office lab. This can be adjusted via Menu > Settings > Display > then touch the Backlight button and use the universal control knob (lower panel) to set the level.

Ports located on the rear include the usual communications connections, plus an HDMI port for connecting an external monitor (Figure 3). You’ll also find Trigger In and Trigger Out connectors, as well as the 50 MHz arbitrary waveform generator (AWG) RF output. The AWG is controlled from the front and can be set to several normal waveforms, including sample ECG and EEG.

Figure 4 shows the two 8-channel digital input ports. Being an EMC engineer, I did not test the digital ports. Attaching the cables could potentially pull on the connectors, causing reliability issues.

Setting up the MXO38 for EMC measurements

When I’m looking for clock harmonics on a PCB or cable, I’ll set the vertical sensitivity to the most sensitive: 1 mV/division. For larger signals such as power conversion circuits, I’ll use the Autoset button to get me in the general range, then adjust trigger level, horizontal, and vertical adjustments to obtain the best viewable display.

Next, I turn on the spectrum analyzer by either selecting Menu > Spectrum or simply pressing the “Spec” button in the vertical section of the front panel. You’ll obtain the menu selections shown in Figure 5.

I usually prefer to set the Start and Stop frequencies manually, so press Start/Stop, then double-tap the Start and manually enter the desired start frequency. I usually leave this at the default “zero”. Repeat this by double-tapping the Stop button and manually setting the frequency. Sometimes, I will leave this at the default 1 GHz if I wish to see the whole spectrum.

Next, I like to turn off Auto resolution bandwidth (RBW), so I can manually set it to 100 kHz or 120 kHz (MIL or commercial, respectively), depending on what the appropriate standard requires for frequencies below 1 GHz.

I set the Window Type to the default Blackman-Harris setting. Note that there are four capture modes: Normal, Min Hold, Max Hold, and Average. For general probing, you should leave this set to Normal. For intermittent or broadband signals, I often use Max Hold to capture the signal envelopes for a few seconds.

The last steps include pressing Scale and choosing dBµV from the default dBm. Usually, I didn’t have to change the default scale factors, but you can change them as needed. This setup was far easier than I’ve seen on other manufacturers’ oscilloscopes.

The spectrum analyzer menu includes a feature that automatically adds peak markers (Peak List; Figure 6) to spectral peaks. These labels include frequency and amplitude for each peak, as you’ll see later. You can add annotations (text, arrows, boxes, etc.) to the display. The oscilloscope stores screen captures in its internal memory or on a USB thumb drive, if installed. You can save and recall setups as well.

EMC measurements

Let’s try it out on some basic EMC characterization measurements on an Arduino Due Version R3 embedded processor, as there’s plenty to see. Let’s dive in!

This older model Arduino Due R3 is ideal for testing because it includes a conventional DC-DC converter with a switch inductor easily accessible (Figure 7). It is based on an Atmel ATSAM3X8E ARM Cortex 32-bit processor clocking at 84 MHz (12 MHz clock oscillator). The DC-DC converter runs at about 500 kHz and USB at 16 MHz. We’ll see all those signals clearly. I had to order mine from eBay, as the newest versions of the board do not include the same type of power conversion circuit.

Let’s make some real measurements. A proper characterization of a product for radiated emissions requires three steps. I discuss this in detail in volume 2 of my EMC Troubleshooting Trilogy books [2]. We’ll demonstrate the first step in this article.

1. Near-field probing to characterize the spectral response for all high-energy components
2. Characterization of high-frequency cable harmonic currents for all attached cables
3. Measurements a short distance away (~1 m) to confirm the harmonics that actually radiate

I used an R&S H-field probe (1 cm loop) for all the measurement points (Figure 8). Near-field probing of a PC board or cable should always be the first step. We want to identify each high-energy component and document its spectral characteristics.

A loop probe couples nicely to the switch inductor of the DC-DC buck converter, which allows us to observe many important waveforms and spectral displays (Figure 9).

Using cursors and measuring the distance between switched waveforms confirms the converter is switching at a little over 500 kHz. There is ringing due to operating in discontinuous conduction mode (DCM). This ringing will manifest as a broad peak at the ring frequency, in this case, 4.2 MHz. You can observe this peak in Figure 10. I describe this concept in more detail in [3]. We can observe that each frequency spike occurs at 500 kHz intervals.

Now, let’s look at the 12-MHz crystal oscillator, shown in Figure 11. Probing near the oscillator confirms the 12-MHz harmonics. Here, we’ve increased the upper frequency limit to 500 MHz.

Note the use of Peak List to identify major harmonics. The adjustments, Max Results, Threshold, and Peak Excursion can be varied to automatically label just the most important peaks. Note that the delta difference in adjacent peaks in Figure 12 is exactly 12 MHz and that some harmonics extend out to 500 MHz, or more.

Next, let’s measure the 16-MHz USB clock.

Again, I coupled the probe to the USB IC and its 16-MHz clock oscillator in Figure 13. Note that the delta difference in adjacent peaks is exactly 12 MHz and that some harmonics extend out to 400 MHz, shown in Figure 14.

Summary

I found the MXO3 oscilloscope is very useful for general EMC debugging and troubleshooting. The ability to observe both the time domain and frequency domain with its fast capture is a real advantage. Up to eight waveforms may be stored for comparison to the measured waveform. This will greatly help in comparing “before and after” measurements as various mitigations are tried.

Part 2 of this series will discuss how to measure high-frequency currents on cables. Because cables are the most frequent contributor to radiated emissions, characterizing these harmonic currents is really one of the most important steps in mitigating emissions. We’ll also demonstrate how to connect a simple antenna to the scope to measure the relative strength of the actual emissions.

References

[1] Rohde & Schwarz, https://www.rohde-schwarz.com/us/home_48230.html
[2] Wyatt, EMC Troubleshooting Trilogy, https://www.amazon.com/stores/Kenneth-Wyatt/author/B00SNQ1LJ2
[3] Sensepeek, https://sensepeek.com
[4] Wyatt, Review: PCBite circuit board holder and probe kit, https://www.testandmeasurementtips.com/review-pcbite-circuit-board-holder-and-probe-kit/
[5] Wyatt, Characterize EMI from DC-DC converter ringing, https://www.eeworldonline.com/characterize-emi-from-dc-dc-converter-ringing/

The post R&S MXO3 Oscilloscope for EMC measurements: part 1 appeared first on Test & Measurement Tips.

In part 3 of this series, we used Kirchhoff’s voltage law to derive the branch currents and node voltages for an unbalanced Wheatstone bridge with five known, fixed resistors (Figure 1). Now, we propose to replace R5 with a digital multimeter (DMM) to directly measure V_V – V_X and use that measurement to determine the value of an unknown resistor in the position of R₄.

Q: Could we backtrack first? Last time, we solved three loop equations. I was trying to use Kirchhoff’s current law to write six-node equations and got lost. Could you elaborate?

How far did you get?
I_IN = I₁ + I₂, I₁ = I₃ + I₅, I2 = I₄ + I₅, I₃ = I₁ – I₅…

Stop there. Your first three are good, but the last one is just a restatement of the second one. We need an independent equation for I₃.

We know I₃ = V_V/R₃, but we don’t know V_V.
Right, but we can express V_V in terms of one of our original six unknowns, namely I₁. V_V is V_IN minus the voltage drop across R₁, which is I₁ times R₁. Substituting our component values, we can write:

Similarly, we can derive V_X as a function of I₂ and solve for I₄:

And finally, I₅ is V_V minus V_X divided by R₅:

Table 1 shows all six node equations in the format required for the online solver^[1] I’m using, and the solutions are shown in red. With this approach, the solver generates the branch currents directly; we don’t have to derive them from the loop currents. I’ve also included our results from part 3 based on the loop method, with the loop values in blue, and the results agree within a third of a percent.

Table 1. Node equations and solutions.

Now, let’s look at the situation where we have an unknown resistor but a known voltage.
Right. Keep in mind that in the 19th century, galvanometers were good at indicating zero current, but not good at quantifying nonzero levels. Consequently, accurate measurements were made under zero-current conditions. We no longer face that limitation, and we can use a measurement setup in Figure 2, which shows a way to make strain-gauge measurements. Here, a DMM, data logger, or data-acquisition instrument replaces the galvanometer from part 2, and R₅ from Figure 1 becomes essentially infinite. R_X, which takes the place of R₄, is a strain-gauge element that has a resistance of 120 Ω when not subjected to strain. To optimize measurement resolution, we assign resistors R₁, R₂, and R₃ the same 120-Ω value.

So, we basically have two voltage dividers.
Right:

and

We can now write an equation for what our DMM would read for a resistance value R_X:

Now we can solve for R_X:

Figure 3 plots this transfer function. Note that for strain-gauge measurements, resistance changes are often less than 1%, and sometimes much less.

Figure 4 shows a simulation result.

Q: Why use the bridge—why not just measure the resistance of R_X directly?
A: Temperature compensation is one reason. In Figure 2, R₂ is often a strain-gauge element identical to R_X maintained at the ambient temperature of R_X, but that never undergoes strain. If both R₂ and R_X increase by 0.1% because of a rise in ambient temperature, but R_X continues to be unstrained, the DMM will continue to read zero.

Q: What is strain anyway, and what are its units?
A: Good question. That’s beyond the scope of this series on Kirchhoff’s laws, but we can consider strain measurement in depth in a future series.

Reference

[1] Simultaneous Linear Equations Solver for Six Variables, handymath.com.

Related EEWorld Online content

Sorting out PC-based instrumentation
Capabilities of modern data recorders
DAQ Series: Methodology associated with data acquisition
Wheatstone bridge, Part 2: Additional considerations
Things to know about multimeters
What’s a half digit and are they all the same?

The post Making sense of test circuits with Kirchhoff’s laws: part 4 appeared first on Test & Measurement Tips.

Connecting electronic devices to test equipment can be daunting, especially when building an automated test system and connecting several test instruments to devices under test. You probably start with a spreadsheet that lists connector types, cables, connector pins, signal names, and termination points. Things can get hairy very quickly. Is there a better way?

The people at Pickering Interfaces think so. Pickering specializes in PXI switching modules, so they’ve seen firsthand how messy designing cables can be. To help you design test cables, Pickering has developed Test System Architect, a free online design tool. Sign up and give it a try.

“All too often, engineers designing automated test systems don’t consider the signal path,” Pickering’s Kyle Voosen told me when I met him at Pickering’s Boston office. “Errors in spreadsheets only become apparent once you build the cable.” Voosen gave me a demonstration where he specified a cable to connect four devices to a multimeter through a switch matrix.

With Test System Architect, you specify the connector type, cable length (with or without accounting for connector size), and then you add the connections. You can also pull in specifications for Pickering’s PXI switching modules to complete the design. Test System Architect will show you the cable design and its schematic. When designing a cable, you can select from PXI switch modules that include matrix, mux, RF mux, and general-purpose switches. You can also add PXI, PXIe hybrid, and LXI/USB chassis into your overall design.

Through your online account, Test System Architect will let you save your design documentation. You can then export it as a data file, SVG file, PNG graphic, datasheet, or BOM CSV file.

The post Design test cables with this free online tool appeared first on Test & Measurement Tips.

We concluded part 2 of this series by starting to write node and loop equations for a five-resistor Wheatstone bridge circuit, to calculate the currents through each resistor. Figure 1 repeats Figure 3 from part 2, but I’ve assigned component values so we can calculate a numerical result. Note that I’ve retained the V_V and V_X labels, which denote the nodes connecting to the variable and unknown resistors in our version of the bridge with a galvanometer.

Q: How do we know which directions of the currents, for example, through R₅?
A: We don’t. The current direction through R₅ will depend on the resistor values. If our arrow direction is “wrong,” we will get a negative value of I₅.

Q: So, how do we solve this circuit?
A: We’ll need six equations to compute the six unknown branch currents (red arrows) or three equations to compute the three unknown loop currents. The latter approach looks easier, so we’ll start there, beginning with the loop on the left. Starting at the top and moving clockwise, we note that the current through R₁ is I_A–I_B, so the drop across R1 is (I_A–I_B)R₁. Next, the current through R₃ is I_A–I_C, so R₃ contributes a drop of (I_A–I_C)R₃. Finally, continuing clockwise, we start with our source, which, from our perspective, adds a negative voltage drop. So, we can write our first loop equation summing the voltage drops:

We continue this process for the top right and bottom right loops to get the additional loop equations:

We will probably want to use a program like MATLAB or an online tool to solve these equations, so we’ll put them in the following form, where each variable is preceded by a constant and the right-hand term is a constant:

With some algebra, we come up with these versions of the three equations:

Next, we insert the numerical values:

I used an online solver^[1] to get the results shown in Table 1. Since I entered values in kilohms, the results are in milliamps.

Table 1. Kirchhoff Loop equations and solutions:

Q: How do we use these results to get our branch currents and node voltages?
A: We quickly notice that I_IN equals I_A, or 0.393 mA, and we can calculate the equivalent impedance of our resistor network as 1.5 V divided by 0.393 mA, or about 3.82 kΩ. We also see that I₄ equals I_C, which equals 0.139 mA, so V_X equals 0.139 mA times 6.8 kΩ, or 947 mV. Similarly, I₂ equals I_B, which equals 0.251 mA, and the drop across R₂ equals 0.251 mA times 2.2 kΩ, or 553 mV.

For the other resistor currents, we’ll need to subtract loop currents. The R₅ current is I_C minus I_B, or -0.112 mA, so the actual current flow is in the opposite direction from the I₅ arrow, and the voltage across R₅ is -112 mV. We can continue with R₁ (for which I₁=I_A–I_B) and R₃ (I₃=I_A–I_C), with the complete results shown in Figure 2.

Q: Now we’ll have to build this circuit to see if the calculations are correct.
A: My thought was to simulate it in LTspice^[2] as shown in Figure 3. The traces on the left for V_V and V_X are a bit difficult to see, so I have added some notation. The simulation confirms our calculated result.

And then I breadboarded the circuit with 1% resistors and confirmed the equivalent resistance, as shown in Figure 4, and the node voltages.

Q: Why so many resistors?
A: I didn’t seem to have a 6.8-kΩ version in my collection, so, alluding back to the rat’s nest in part 1, I had to build one out of parts I did have.

Q: What do we look at next?
A: We previously commented on the downside of using a balanced bridge and galvanometer to determine the value of an unknown resistor (in the R₄ position). In part 4, we’ll look at using an unbalanced bridge and a high-impedance voltmeter to make that measurement, a technique useful in strain-gauge and other sensor applications.

References

[1] Simultaneous Linear Equations Solver for Three Variables, handymath.com.
[2] LTspice, Analog Devices

Related EEWorld Online content

Wheatstone bridge, Part 2: Additional considerations
Using online design tools
The why and how of matched resistors: part 1
How to choose analog-signal-chain components: part 2
When you can use Spice and its variants, when you can’t
Video: Prototyping electronic designs on a breadboard

The post Making sense of test circuits with Kirchhoff’s laws: part 3 appeared first on Test & Measurement Tips.

As powering circuits and systems get more complex, you often need to source, sink, and measure DC power for more than just DC levels. After all, DC isn’t really DC. Loads vary in real-time, which adds AC components to power rails. Furthermore, today’s boards and systems often have multiple power rails, and you need to see how they interact. The IT2705 mainframe and modules from ITECH provide eight channels of sources, loads, and source-measure units (SMUs).

DC power modules support:

• 20 V, 3 A, 20 W
• 30 V, 15 A, 200 W and 30 A, 500 W
• 60 V, 10 A, 200 W and 20 A, 500 W
• 150 V, 5 A, 200 W and 10 A, 500 W

The system can sample up to 200 ksamples/s. You can operate the system in oscilloscope or datalogger modes through the graphical display, front-panel controls, or with software running on a PC. That’s in addition to having digital voltage and current displayed on the screen. Communications links include USB and LAN.

You can operate the sources and loads in either a list mode to get precise steps or in arb mode to simulate transients such as surges and drops. A sequence mode lets you combine output waveforms to consistently simulate usage conditions. The system lets you use loads to simulate batteries for testing chargers under consistent conditions. Electronic loads let you test your power sources under constant voltage, constant current, and constant resistance conditions. A dynamic mode switches between two preset test conditions.

The post Modular DC analyzer sources, sinks, and measures appeared first on Test & Measurement Tips.

In part 1 of this series, we looked at ways to simplify resistor networks by identifying series and parallel combinations of resistors. We closed with a look at a version of the Wheatstone bridge, such as the one in Figure 1. Although it has only five resistors, not one of them is in series or parallel with another, so the simple equations we used in part 1 won’t work here. There is a workaround, called a delta-to-wye conversion.^[1] But that just gives you three more equations to memorize, and unless you do the conversion often (if you work with three-phase power, for example), those equations aren’t likely to become second nature, the way the series and parallel resistor equations are for most electrical engineers.

Q: Wait, what about the version of the Wheatstone bridge with a galvanometer taking the place of R₅?
A: That version, shown in Figure 2, is easier to analyze. Here, R₃ becomes a variable resistor R_V, and R₄ becomes an unknown resistor R_X, whose value we are trying to find.

In the classic Wheatstone bridge application, we manually adjust R_V until the galvanometer or multimeter voltage V_G is 0, at which point we have what we call a balanced bridge. It consists of two simple voltage dividers, and R5 from Figure 1 is essentially infinite. R₁ and R_V make up the first divider, and the voltage across R_V is:

R₂ and R_X make up the second divider, and the voltage across R_X is:

Since the galvanometer reads zero, we can determine R_X as follows:

Q: What’s the drawback to this balanced bridge?
A: If we are trying to measure a parameter such as strain in real time, it’s not practical to manually adjust R_v until V_G is zero, then record the value of R_V, and calculate R_X. We could probably kludge together something based on a voltage-controlled or digitally controlled resistor to replace the manual adjustment, but Kirchhoff’s and Ohm’s laws offer a simpler way for us to derive R_X from the voltages and currents in a fixed-resistor bridge.

Q: How does that work?
A: Let’s go back to our original Figure 1 bridge with five fixed resistors. In Figure 3, I’ve rearranged Figure 1 to make more room for notations showing branch (red arrows) and loop (blue arrows) currents. Kirchhoff gives us two laws that we can apply to solving such a problem. The current law states that the net sum of currents into any node is 0. In Figure 3, for example, that law implies the following:

The second law, Kirchhoff’s voltage law, states that the voltage drops around any closed loop must equal zero. For the loop on the left of Figure 3, this law results in the following equation:

For a given circuit, each law will give us multiple equations on multiple unknowns. Which law or combination of the two laws is easiest for us to use will depend on exactly what we are trying to determine. In our case, we have two goals. First, following up on our work from part 1, what is the equivalent impedance of the bridge if all resistors are fixed and we know their individual values? In other words, what is V_IN/I_IN? And second, of particular interest from a test-and measurement-perspective, is this question: If R₁, R₂, R₃, and R₅ are fixed and known and R₄ is unknown (it might be a strain-gauge element or temperature sensor, for instance), can we calculate R₄ based on the voltage V_V – V_X across R₅? We’ll take a closer look in part 3.

References

[1] Delta-to-Wye Equivalent Circuits, Utah State University.

Related EEWorld content

Basic instrumentation for the electronics workbench
William Sturgeon and his galvanometer
How does a thermocouple work, and do I really need an ice bath? part 1
Wheatstone bridge, Part 1: principles and basic applications
An intuitive view of Maxwell’s equations
FAQ on three-phase AC power: part 1

The post Making sense of test circuits with Kirchhoff’s laws: part 2 appeared first on Test & Measurement Tips.

Low-cost spectrum analyzers often lack tracking generators, preventing you from having a signal source that tracks the analyzer’s frequency sweep. Not having a tracking generator can make some measurements difficult. You can, however, use a low-cost noise generator to characterize filters, amplifiers, and coax cable loss. In effect, they serve as a scalar network analyzer (no phase information).

In this article, I’ll describe how to use a noise source as a substitute for a tracking generator.

Figure 1 shows a typical broadband noise source. Available from Amazon^[1], the broadband noise source will produce a wide range of RF signals from near DC to 2 GHz. This board is designed to run on 12 V, but I found it also operated well at 5 V. I did notice the amplifiers get quite hot.

The noise is generated by biasing a Zener or Schottky diode, which creates low-level broadband noise. The board then runs this noise voltage through three broadband amplifiers and terminates it into a 50 Ω matching network. In Figure 1, the noise diode is in the board’s lower-left corner, and the three amplifier stages lie between that and the RF output port on the right.

Noise voltage is difficult to capture on an oscilloscope because of randomness. I was, however, able to trigger on one cycle that appeared to have about 500 ps of rise time. I was using a R&S MXO38 (8-channel, 12-bit, 1 GHz bandwidth) oscilloscope and the spectrum plot showed usable broadband emission all the way past 1 GHz (Figure 2).

Noise generator applications

I switched over to using my Siglent SSA3032X spectrum analyzer so I could capture multiple spectral plots. We’ll measure cavity resonance and a couple of filter responses.

Cavity Resonance: We’ll use the “cookie tin” cavity resonance demo (Figure 3) we used to demonstrate ferrite absorber performance earlier^[2] to demonstrate how the noise source can identify structural resonances. Recall the cookie tin lid had two BNC connectors attached with short (1 cm) stubs soldered to the center conductors. The position of these is not important. We’ll drive one with the noise source and connect the other to the analyzer input port.

The resonance equation was covered in^[2], and for a circular cavity, the fundamental resonant frequency was 1.275 GHz. Figure 4 shows the resulting screen capture with the marker at the peak of 1.23 GHz. The yellow trace was the baseline measurement with the noise source off.

Scanner filter: The radio scanner filter is designed to allow VHF Low Band (30 to 50 MHz), VHF High Band (140 MHz to 174 MHz and UHF Band (420 MHz to 512 MHz) to be heard and to block other strong ambient signals such as AM/FM broadcast, TV, and other high-powered signals. Figure 5 shows the general test setup using a noise generator to drive the filter, which is connected to the analyzer input port.

Figure 6 shows the two bands being blocked with markers at the band edges of the VHF Low Band and VHF High Band. The notches are about 20 dB down. The yellow trace indicates the noise floor of the measurement.

FM Broadcast Notch Filter: This will be a similar measurement to that above, except we’ll use a Mini-Circuits ZX75BS-88108-S+ FM Broadcast Band Stop Filter^[3]. This would be a good filter to use when operating a spectrum analyzer near high-powered FM stations to prevent overload of the front end. Figure 7 shows the test setup.

In Figure 8, you can clearly observe the band-stop filter edges at 81 MHz and 108 MHz. The yellow trace is the measurement noise floor.

Summary

For spectrum analyzers lacking a tracking generator, this broadband RF noise source is affordable and will help measure a host of RF applications, including filter and amplifier responses. While missing the dynamic range of a vector network analyzer, it can serve in a pinch to help identify or confirm RF characteristics of many devices.

References

1. Noise Source (Amazon, $17)
2. Wyatt, Kenneth, How to compare EMI absorption materials with a cookie tin
3. Mini-Circuits FM Band Stop Filter

The post Noise generator substitutes for tracking generator appeared first on Test & Measurement Tips.

The post Transcat to distribute, calibrate Ametek Programmable Power supplies in Americas appeared first on Test & Measurement Tips.

When you buy test instruments, you hope they’ll have the flexibility to provide the necessary stimulus to the device under test (DUT) and acquire and process the response. Occasionally, however, you’ll need to design an external network to get exactly what you want, or perhaps worse, you’ll have to deal with a custom interface network that somebody else built.

Q: You mean like an external network of resistors?
A: Yes, something such as the rats’ nest in Figure 1. The result is similar to what you might get if you gave a handful of random components to an engineer and asked for an interface between a test system and DUT without using a circuit board.

Q: So basically, is it just a combination of resistors in parallel and series to get the necessary values?
A: Right. Figure 2 shows a specific schematic, but with fewer resistors than Figure 1.

Q: How can we analyze this?
A: If you have the actual circuit and a multimeter, you can disconnect the voltage source and measure the equivalent impedance. If you have a schematic such as Figure 2 or can derive one, you can use Ohm’s law and Kirchhoff’s current or voltage law and write node or loop equations, respectively, allowing you to calculate currents through each resistor and the voltages at each node. But we can avoid solving simultaneous equations in multiple unknowns if we can identify series and parallel combinations of resistors, as shown in Figure 3.

We know that for two resistors in series, we simply add the resistances, and for two resistors R₁ and R₂ in parallel, the resistance is the product over the sum:

For N resistors in parallel, you can take them two at a time, computing the successive products over sums, or you might find it more convenient to compute the inverse of the sum of the reciprocals:

For the three parallel resistors in the light blue box on the right, we can determine that 820 Ω in parallel with 360 Ω equals 250 Ω, which in turn is in parallel with 1 kΩ and equals 200 Ω. Or, we can simply calculate the inverse of 1/1,000 Ω plus 1/820 Ω plus 1/360 Ω, which also equals 200 Ω. That 200 Ω is in series with 1.8 kΩ, so in the dark blue block on the right, we have a total of 2 kΩ.

You can perform similar calculations on resistors in the other blocks shown in Figure 3, and get the voltage-divider network shown in Figure 4. This divider could be useful for a DUT that requires a supply voltage V, and that also has two high-impedance inputs requiring voltages of 50% and 20% of V. The resistor approach costs less than buying two additional programmable power supplies to generate the 50% and 20% levels. It can be faster than a multiplexer-based approach, where you would use a single supply to sequentially apply the required 100%, 50%, and 20% levels. And if your DUT requires that all three voltages be present simultaneously, the multiplexer won’t work at all. Just remember that the voltages must go to high-impedance loads to prevent loading down the circuit.

Q: Can all resistor networks be simplified into series and parallel combinations?
A: No. Consider the circuit on the left of Figure 5. Although it’s simple, consisting of just five resistors, none are in series or parallel with any of the others (unless R₅ is either zero or infinite). This particular circuit is very important in test and measurement. If I redraw it as shown on the right, you might recognize it.

Q: It looks like the Wheatstone bridge, but isn’t R₂ variable, and isn’t a galvanometer used in place of R₅?
A: Yes, it’s the Wheatstone bridge, and in the original Wheatstone bridges, you’re right about R₂ and the galvanometer. In addition, R₄ was the unknown value that the bridge was designed to measure. Next time, we’ll comment on why the variable resistor and galvanometer are not often used in modern applications. In part 2, we’ll see how we can analyze voltages and currents in this circuit using Ohm’s law and Kirchhoff’s current law, and we’ll see why this configuration is useful in strain gauges and other sensitive measurement devices.

Related EE World content

Does Kirchhoff’s Voltage Law really fail?
The why and how of matched resistors: part 1
Engineers use AI/ML to improve test
That Maxwell book that was returned to a library 115 years late: what’s it about? (Part 1)
Things to know about multimeters
Resistivity, conductivity, and Kirchhoff’s laws
Teardown: 1950s Heathkit vacuum tube oscilloscope

The post Making sense of test circuits with Kirchhoff’s laws: part 1 appeared first on Test & Measurement Tips.

NI’s line of PXI-based test instrumentation has received a significant boost. Nearing 30 years old, PXI has become the standard for modular instrumentation in automated test for it boasts a thriving multi-vendor ecosystem. PXI originator NI has added an 18-slot chassis, two embedded controllers, an two multifunction data-acquisition (DAQ) modules and an oscilloscope to its line.

The NI-PXIe 1081 chassis contains 17 hybrid PXI/PXIe slots and a single PXIe-only slot. Having hybrid slots that let you insert everything from the latest PXIe instruments back to legacy PXI instrument cards. Thus, you can keep using cards you have from legacy test systems. A single PXIe slot supports a system-controller module, which can be an embedded controller to an interface card to a host computer. The chassis features 58 W of cooling capacity per slot. Data-transfer rates reach 2 GB/sec.

Two new PXIe embedded system controllers differ in their CPUs. The PXIe-8842 contains a 6-core Intel Core i5 processor while the PXIe-8862 contains an 8-core Intel Core i7 processor. Both controllers have four USB 2.0, two USB 3 Super-Speed, one or two DisplayPort, two USB-C, two 2.5 G Ethernet, and a 9-pin GPIO port. A GPIB is optional for applications requiring box instruments. You can order then with pre-installed Windows 10, Windows 11, or Linux operating systems.

The PXIe-6381 and PXIe-6383 Multifunction DAQ modules bring 18-bit analog input and add analog output, digital I/O, and counter/timers to a PXI-based system. The difference between the two modules comes in the number of channels: analog input (16/32 single ended or 8/16 differential), analog output (2/4), and digital I/O (24/48).

When you need higher-speed signal capture, NI offers the PXIe-5108, a 14-bit, 100 MHz digital oscilloscope. Variants include four or eight channels with 512 MB of acquisition memory and 250 MSa/sec digitizing rate.

The post NI boosts PXI with controllers, chassis, and I/O appeared first on Test & Measurement Tips.

When testing a USB charger or power bank, you can connect a phone or other device and see if it charges. That’s nice, but if you need more data or suspect a failure, you need to interrogate the power lines. While it’s possible to do that with a breakout board, some resistors, and a handheld or bench multimeter, you can use an inline USB meter for very little money.

If you check online, you’ll find dozens of inline USB testers. We tried two from Eversame, available from Amazon and other online sellers. You’ll find what looks like identical products sold under different brand names. Figure 1 shows the two meters and the electronic load.

These inline meters not only measure voltage and current, but they also calculate a USB port’s output power, amp-hours, watt-hours, and load resistance. They also provide chart recorders for voltage, current, and output voltage ripple. Their screens also provide arrows that indicate current flow from input to output. In two of the videos below, we take you through the menus. In a third video, we try out an electronic load, which lets you vary the load so you don’t need a set of fixed resistors. We tried fixed-resistor loads, too.

Both inline testers feature USB-A and USB-C ports, while one adds a micro-USB input port. You’ll also need some cables, and you may require USB breakout boards for connecting to other equipment, such as an oscilloscope. Because UCB-C chargers and power banks produce up to four output voltages, you’ll need something to force them to the desired output voltage unless the default 5 V is sufficient. I used an attachment board that has a set of DIP switches and a two-terminal block for connecting a load, shown in Figure 2. It’s shown connected to a single 10 Ω, 10 W power resistor. For a higher current, we have four resistors connected in parallel to make a 2.5 Ω, 40 W load.

Packaged USB inline tester, Model J7-c

Figure 3 shows the Eversame 2-in-1 Type C USB Tester Color Screen LCD Digital Multimeter Model J7-c, $16.14 on Amazon.

For a low cost, you get a device that’s useful for testing sources such as phone chargers and power banks. You can also use it to see the current a device draws. It has USB Type-A and Type-C input and output connectors. You can plug it directly into a charger or power bank, or use an extension cable, which will result in additional power losses. You’ll likely need a cable to connect the load shown to the Type-C output port in Figure 2.

The color LCD screen is small, and the text, other than the primary voltage and current readouts, is downright tiny. A single button lets you scroll through several screens (see video), including two chart recorders. One chart recorder plots output voltage and current, and the other shows voltage ripple.

Larger screen, no package

Figure 4 shows the Eversame USB C Power Meter Tester, Voltmeter Ammeter Load Tester, $23.39 on Amazon.

This USB inline meter could use some packaging, as it’s just two boards screwed together with spacers. It looks like someone assembles these in a back room. That makes the tester somewhat fragile compared to most other USB inline testers on the market. The good news is it’s easy to take apart. There’s no part number on the product, nor in the Amazon listing. The meter comes with a USB-C to USB-C cable. This tool adds a micro-USB input connector for passing power from an older power bank that has such a connector.

After using the J7-c, I appreciated the larger screen. Four buttons let you select the screen display, change settings, and rotate the screen left or right. That’s handy, should you connect it to a wall charger and find that the screen is upside down or sideways. As the video shows, you also get the chart recorder, power, resistance, watt-hour, and mA-hour calculations. In the video, you’ll see the readout screens and settings menu.

What’s a circuit without a load?

Fixed loads have their purpose, but they’re fixed. You get what you get, no more and no less. Dynamic loads, such as connecting a power source to a phone, can give you some insight into charging, but the load changes are out of your control. A variable load lets you change the current drawn from a charger. Figure 5 shows a
Drok USB Load Tester, Electronic Load Test Resistor Module 25W LD25 USB and Type C Interface, $14.95 on Amazon.

With this electronic load, you can vary the current that the load draws from the power source and thus test your source under differing conditions. If you need a programmable dynamic load, you’ll have to spend more money or design your own.

The Drok load can sink up to 4 A. Our tests couldn’t go nearly that high because that’s more than any of our power sources can deliver. It might be worth trying it with a bench power supply some other time.

The load includes USB-A, USB-C, and micro-USB connectors. There’s an LED display that indicates current drawn, and it can indicate power. A temperature sensor and a fan keep things cool. The digits illuminate in sequence, which makes it difficult to see all of them on the video. To the human eye, they appear to be constantly lit. The board has an on/off button and a setting button that sets the display from amps (default) to watts.

Future work

In future articles, we’ll use these tools, an oscilloscope, and USB breakout boards to compare several chargers and power banks. We will also teardown the Eversame USB-C Power Meter Tester, Voltmeter Ammeter Load Tester, the one with no model number, to see how it displays activity on the data lines.

Related EEWorld Online content

Phone chargers produce EMI: We compared four
USB Type C standardization – what you need to know
A look at USB Type-C in power-only applications
How USB Type-C connectors hold the key to maximizing data speeds and power delivery
The three “C’s” of USB Type-C: connectors, controllers and cables

The post Tryout: two low-cost USB inline meters and a load appeared first on Test & Measurement Tips.

With the retirement of Windows 10, as described in part 1 of this series, users of the operating system to control test and measurement systems have a choice to make. In part 2, we looked at what steps you can take if you want to continue using Windows 10, and in part 3, we looked steps you plan to upgrade to Windows 11.^[1]

One factor that shouldn’t present a major issue is the user interface. According to Noman Hussain, vice president of software and strategic business analysis at Pickering Interfaces, “The transition from Windows 10 to Windows 11 is relatively seamless for experienced users. The core file system remains familiar—for example, the System32 and SysWOW64 folders, as well as Program Files and Program Files (x86) folders, follow the same structure” (Figure 1.)

When asked if operators used to Windows 10 will find difficulty with the switch to Windows 11, Toby Marsden, head of go-to-market strategy and strategic alliances at Keysight Technologies, said, “No, in reality, it is the reverse—that older technology platforms need more specific training; most organizations and users use more modern operating systems. Any specific skills around Windows 10 are dying out.” Added Steve Summers, director and security lead for aerospace and defense at Emerson Test and Measurement, “The user experience is very similar moving to Windows 11. There is a little disruption as operators see the new toolbar for the first time,” but they adjust quickly. And according to Hussain, “From an operator’s perspective…the learning curve is minimal, and day-to-day tasks can be carried out much as before. The main frustration we hear from users is that Windows updates sometimes change the look and placement of icons, which can be disruptive, but is generally a minor inconvenience rather than a training issue.”

Planning for the next operating system

Ultimately, Microsoft will withdraw support for Windows 11, which debuted October 4, 2021, although no end date has been set as of now. (The current version, 25H2, is set to end October 12, 2027, but will presumably be replaced by Windows 11 version 27H2.)

An ongoing effort is the optimal approach for upgrading to any new operating system or other software while planning ahead for any required revalidation process. Marsden described technical architecture modernization as a continuous process to keep systems updated and operating effectively and said that Keysight can help with testing to ensure systems do not regress and continue to perform well. He acknowledged that some customers still rely on legacy technical architectures to run critical business processes, but he added that “most customers are looking for modern, flexible architectures, as legacy systems can result in the business bending to the system’s functionality rather than enabling new operational efficiencies to be achieved.”

According to Summers, “Teams that plan for validation take a professional approach to software design, including building unit tests and other testing routines, so that updates can be validated more quickly.” His company recommends that engineers follow professional software-design processes when developing new systems, but he acknowledged that many existing test systems do not have the  necessary tools implemented. Summers added, “Customers who have followed our advice to design code validation as part of their development process are in a good position for these revalidations; those who have not adopted this approach are facing a more manual validation process.” He recommended, “Update your development process now to incorporate best code practices—CI/CD [continuous integration/continuous deployment], unit test, etc. This takes more time up front but makes ongoing re-validation much faster and easier in the future.”

According to Hussain, “We always design with long-term support in mind. Our driver packages cover multiple generations of Windows, and we test them across different OS versions so you can move when you’re ready. My advice is to plan migration early and involve us in that process — it makes the whole thing smoother and saves a lot of headaches down the line.”

When asked about the inevitable end of Windows 11, Keysight’s Marsden said, “Don’t panic!” He advised a thorough evaluation to ensure that that operating-system migration won’t impact key business processes, but he cautioned that failure to address migration will ultimately lead to higher costs and lower business-efficiency gains. “Business continuity needs to be assessed around key workflows, application functionality, and performance,” Marsden noted, adding that any existing system should be thoroughly tested before migrating to a new Windows version.

Said Hussain at Pickering, “Our advice is to plan ahead rather than react at the last moment. Customers should build a migration strategy that considers both short-term needs and long-term continuity, including hardware readiness, software compatibility, and regulatory requirements.” He concluded, “Change is inevitable. Plan for it now to avoid future hiccups.”

Related EE World content

Contending with Windows 10’s retirement: part 3
Contending with Windows 10’s retirement: part 2
Contending with Windows 10’s retirement: part 1
How modularity benefits test systems
What is an instrument driver and why do I need one?
Tech Toolbox: Test & Measurement
Tech Support Challenges and IoT
Designing The Computational Architecture Of The Future

The post Contending with Windows 10’s retirement: part 4 appeared first on Test & Measurement Tips.

In an earlier series, we discussed how the oscilloscope has evolved over the past three-quarters of a century. We noted that the basic horizontal and vertical controls remain functionally similar, but today’s digital oscilloscopes offer many features, including waveform math functions, that go well beyond what an oscilloscope could do in the 1950s. We touched on triggering in our earlier series,^[1] now we will take a comprehensive look at the topic, addressing some topics that university-level electrical-engineering courses might not cover. ^[2]

Can we begin with a look at triggering basics?
Sure. First, keep in mind what we’re trying to do with triggering. If you apply a fixed-frequency sinusoidal input to an oscilloscope channel without triggering, you’ll see what appears to be multiple waveforms that dance across the screen. What you are actually seeing are successive snapshots of random segments of the same waveform with no particular phase relationship. Triggering stabilizes the display, so the segments overlap and appear static, letting you measure parameters such as amplitude, frequency, and, for multichannel displays, phase.

Figure 1 shows a close-up of a vintage analog oscilloscope we discussed in our earlier series, highlighting the oscilloscope trigger controls. As you can see, you can select a trigger level as well as the slope of the waveform on which the trigger occurs. In addition, you can select from multiple modes, including AUTO and NORM.

Now let’s see how that compares to the trigger controls of a modern digital scope, as shown in Figure 2. For this demonstration, I am using a USB oscilloscope (a PicoScope 2204A) from Pico Technology, which replaces the physical knobs in Figure 1 with on-screen icons. I have clicked the trigger function from the top row, and the trigger menu appears to the left of the trace display. As with the analog scope, I can still select the AUTO trigger mode, as shown at the top of the trigger menu. Moving down, I can choose the trigger type (simple edge), the source (channel A in this case), the trigger threshold level (0 V in this case), and the edge direction at the time of trigger (falling, or negative slope), just as I could have with the analog oscilloscope.

One thing the PicoScope does that the analog oscilloscope couldn’t is allow me to select the pre-trigger range, which I have set to 50%. Consequently, the oscilloscope triggers at the center of the screen — at the position of the yellow diamond — when a negative-going portion of the displayed 1.6 V_PP 1 kHz sinusoid passes through 0.

Could we see another example?
Yes. In Figure 3, I’ve chosen to trigger on a rising edge at a level of 200 mV, and I’ve changed the pre-trigger setting to 20%. The instrument then triggers when the waveform rises to the 0.4 V level at a point one-fifth of the way across the screen.

What’s the EXT input in Figure 1 used for?
Generally, you’ll want to trigger on an internal signal applied to a measurement channel — channel A in Figure 2 and Figure 3. In some cases, however, you may want to trigger on an external signal that you don’t need to view on the scope screen. For example, your system under test may generate a fault signal in the event of an overcurrent or overvoltage condition. You can connect that fault signal to the external (EXT) trigger input and use your measurement channels to display what happens elsewhere, leading up to or in the immediate aftermath of the fault. There are several other times when the use of an external trigger may be useful[3]: when the signal you want to measure lacks sufficient regular features (such as zero crossings) to ensure stable internal triggering, for example.

What else should we know about oscilloscope triggering?
In this post, we’ve looked at the basic functions that could be performed on an analog oscilloscope. Often, you’ll want to trigger on a condition that would be difficult or impossible to detect using analog techniques — when a signal enters or departs from a specified voltage range, for example, or when an anomaly occurs on the transmission or reception of a specific digital code. We’ll take a closer look next time.

References
[1] Nelson, Rick, “Get the most from your oscilloscope: part 1,” Test & Measurement Tips, September 29, 2025.
[2] Etheridge, Ian, “Oscilloscope Triggers: What They Didn’t Teach Me in School,” Digilent, August 18, 2025.
[3] Reed, Callum, “What is an External Trigger in Oscilloscopes? ” Keysight Technologies, September 17, 2025.

Related EE World content
Triggering Enters the Picture
Trigger me this
How to achieve accurate oscilloscope measurements
Basics of oscilloscope roll mode, act-on-event, and the trigger menu
Understanding basic oscilloscope uses
What’s a USB oscilloscope?

The post Make the most of oscilloscope triggering: part 1 appeared first on Test & Measurement Tips.

Engineers designing DisplayPort 2.1 into devices need a way to verify compliance with standards and that the interface will interoperate with other devices. This involves viewing physical-layer (PHY) eye diagrams at both transmitters and receivers. DisplayPort 2.1 PHY compliance test software options from Teledyne LeCroy oscilloscopes add to the tests you can perform with the company’s equipment and QualiPHY 2 software.

Developed by VESA, DisplayPort competes with HDMI for video. Most graphics cards provide both ports. DisplayPort V2.1 can deliver up to 80 Gbps of data, which translates to 8K at 120 Hz 4K at 240 Hz screen update rates.

DisplayPort 2.1 oscilloscope software options include:

• QPHY2-DP2-SOURCE-TX for transmitters (source)
• QPHY2-DP2-SINK-RX for receivers (sink). It supports automated calibration and Sink bit-error rate (BER) compliance testing.
• QPHY2-PC option lets you perform offline analysis on a PC.

The software supports all DisplayPort variants: Ultra-High Bit Rate (UHBR), High Bit Rate (HBR), and Reduced Bit Rate (RBR) test requirements. DisplayPort 2.1 uses four data lanes, whereas the currently most common version (1.4) supports a single lane.

Source: TFT Central

The post Oscilloscope software automates DisplayPort 2.1 PHY testing appeared first on Test & Measurement Tips.

The post USB oscilloscope software adds waveform overlays and other new features appeared first on Test & Measurement Tips.

ESD discharges are tough to locate because they don’t occur periodically. Rather, they occur intermittently. I earlier wrote an application note on how to use an oscilloscope to trace the path of ESD current through your product [1]. In many cases, however, ESD discharges can result from various assembly or production areas in a large industrial facility. I once had to track down ESD events in a tire factory that were gradually degrading the batteries in an autonomous robotic system.

How can we use an oscilloscope to find ESD discharges in a factory environment? I first got this idea from Doug Smith [2] during one of his public seminars. The basic idea is to connect two short identical monopole antennas [3] to separate inputs on an oscilloscope (switch the inputs to 50 Ω impedance) and set the trigger for channel 1, Auto Trigger, and trigger on rise time. Adjust the trigger level for a stable capture by using a piezoelectric BBQ lighter, available in most hardware stores (burn off the gas, and you have a spark generator). Start with the horizontal set at 2 ns/division and the vertical at 200 mV/division.

A tablet oscilloscope, such as the Micsig TO3004 oscilloscope [4], is ideal for this application because all the input ports are along the top. That let me connect short telescoping antennas to channels 1 and 4 (for spacing). Figure 1 shows the oscilloscope with my BBQ lighter used as a test source.

Direction finding of ESD events

The ESD event results in a ring wave due to the inductive wire used as a transmitting “loop” antenna. The important part of the wave is the two leading edges of signals from channels 1 and 4. Note that as the wave ring decreases, the frequency also decreases; thus, for best accuracy, we want to set the cursors up near the start of the ringing.

Figure 2 shows the resulting waves with the ESD source off to the left side. The yellow trace (channel 1 and left-hand antenna in Figure 1) is the first one triggered, thus we see the left antenna received the wave first. Lagging behind by 560 ps is the green wave (channel 4 and right-hand antenna).

An interesting point is the “time-of-flight” calculation. This can be observed from the difference of 560 ps. Electromagnetic waves travel at about 12 inches per nanosecond in free space. The antennas are about 5.5 inches apart, so calculating the time-of-flight between the two antennas is equal to 5.5/12 = 0.4583 or 458 ps. This is in the ballpark of the 560 ps we measured using cursors.

Figure 3 shows the resulting waveforms with the ESD source straight on, facing the oscilloscope, and equidistant from each antenna. Here, we can see both wavefronts are coincident, indicating the ESD source is either straight ahead or directly behind the oscilloscope face and arriving at the antennas simultaneously.

Figure 4 shows the resulting waves with the ESD source off to the right side. The green trace (channel 4 and right-hand antenna) is the first one triggered, thus we see the right antenna received the wave first. Lagging behind by 520 ps is the yellow wave (channel 1 and left-hand antenna).

Because of the portability of the Micsig TO3004 and the fact that each antenna captures the ESD waves, simply lining up the wavefronts will either point you directly towards the ESD source or directly away from the source. By walking several feet one way or the other, you should be able to triangulate the source location.

One note. Because the antennas are relatively short and close together, there may not be sufficient signal capture. You may need to pull out the telescoping antennas further or increase the vertical amplitude sensitivity. I also find angling the two antennas away from each other helps separate the wavefronts a little more.

Another point is that by separating the antennas further apart, a greater difference in the time-of-flight separation in the two wavefronts may be recorded. For example, if the two antennas were connected to equal lengths of coax and handheld three feet apart, the wavefronts would be approximately 3 × 1 ns/foot = 3 ns difference, worst case.

Actual factory ESD example

Real-world ESD discharges will not generally look like nice, clean ring waves as produced by the BBQ lighter, but can look ugly (Figure 5). This was an example captured in that tire factory I mentioned earlier and consisted of multiple ESD bursts. While this may look complicated, remember that the important part of the wave is the front edge as captured by the oscilloscope. The differences in wavefronts will allow you to find direction just as in the examples above.

Summary

While any oscilloscope may be used to “direction find” the source of ESD discharge, a portable tablet oscilloscope like the Micsig TO3004 is ideal, because of the portability, plus with four inputs along the top, antennas may be placed apart from each other for the best directional angle resolution.

References

[1] Tektronix, Troubleshooting ESD failures using an oscilloscope
[2] Doug Smith
[3] Diamond RH789 antenna, Amazon
[4] Review: Micsig TO3004 tablet oscilloscope

The post Locate ESD sources using an oscilloscope and two antennas appeared first on Test & Measurement Tips.

In part 1 of this series, we looked at the use of Windows to control test and measurement systems, and we considered the dilemma posed by the retirement of Windows 10 on October 14. In part 2, we looked at steps you can take if you’re planning to remain with Windows 10 for now. Here in part 3, we’ll discuss how to achieve a smooth migration to Windows 11 if you choose to upgrade soon, which will allow you to safely and securely connect your system to external networks to take advantage of features such as AI-driven cloud-based productivity tools.

Whether you should upgrade to Windows 11 now depends on many details of your specific test system, but the factors listed below can give you a start — if you can check most of the boxes, an upgrade may be your best choice.

• Your existing computer meets the minimum hardware and software requirements to run Windows 11, or you are willing to upgrade your hardware or software
• You do not need to use 32-bit legacy DLLs, or you plan to use Microsoft’s WOW64 emulator
• Your system is not dependent on Windows 10 characteristics that are not present in Windows 11
• You want to connect to an external network to take advantage of cloud-based and AI-enabled processing to boost productivity
• Windows 11 drivers are available for all of the instruments in your system, or you have determined that your 64-bit Windows 10 drivers will work with Windows 11
• Any third-party application programs that you use will run under Windows 11, or you can replace them with Windows 11 versions
• You are willing to undergo a revalidation process if necessary

Proceed with caution, however. Steve Summers, director and security lead for aerospace and defense at Emerson Test and Measurement, warned that upgrades require planning. “Don’t assume that it’s a simple install to do the upgrade,” he said, adding that you’ll need to evaluate the hardware on which you plan to run Windows 11, asking yourself if you could benefit from a higher-performance CPU, more memory, or perhaps a Trusted Platform Module (TPM) chip. Summers added that you’ll also need to determine whether your existing software is compatible with Windows 11.

The minimum requirements for the upgrade, according to Microsoft^[1] include a 64-bit processor with at least two cores running at 1 GHz or faster and a minimum of 4 GB of RAM. The 64-bit requirement is key. Noman Hussain, vice president of software and strategic business analysis at Pickering Interfaces, pointed out that Windows 10 is the last Windows version with native support for 32 bits. “With Windows 11, all systems will be 64-bit,” he emphasized, which could present problems with any legacy software that makes use of 32-bit dynamic link libraries (DLLs). Hussain noted that Microsoft has helped to smooth the transition with its WOW64^[2] Windows-on-Windows emulator, which allows 32-bit DLLs to run on a 64-bit system. “However,” Hussain cautioned, “you would still need to update your system files and hardware configuration to migrate from 32-bit to 64-bit.”

Toby Marsden, head of go-to-market strategy and strategic alliances at Keysight Technologies, said that applications and services that use Windows 10 will require testing to determine whether they rely on specific Windows 10 characteristics that are not present in Windows 11. He cited as an example the Y2K crossover problem, which resulted from date changes hard-programmed into application code. “A similar approach could have enabled applications to operate effectively on the Windows 10 platform,” he said, but there is no guarantee they will operate correctly under Windows 11.

You’ll also need to determine whether Windows 11 drivers are available for all the instruments in your system. If not, you might find that your 64-bit Windows 10 drivers will operate under Windows 11. This is a risky choice, however, because your vendor is unlikely to provide support for its Windows 10 drivers used in this way, and the driver may stop functioning after incremental Windows 11 updates.

Commented Hussein, “All of our PXI and PXIe boards are supported from Windows 7 through Windows 11 — in both 32-bit and 64-bit for Windows 7 through 10 and 64-bit for Windows 11. Some of these boards were originally designed more than 25 years ago and are still fully supported.” He added, “From a driver perspective, the Microsoft Visual C++ 2010 Redistributable^[3] works seamlessly across Windows 7 to Windows 11, allowing the same DLLs to support all these versions.”

Other considerations center on the user interface (UI) and the potential need for test operator retraining. And finally, of course, Microsoft will eventually withdraw support for Windows 11, which debuted October 4, 2021, although no end date has been set as of now. (The current version, 25H2, is set to end October 12, 2027, but will presumably be replaced by Windows 11 version 27H2.) In the final part of this series, we’ll look at UI considerations and ongoing steps you can take to prepare for any operating system’s inevitable obsolescence.

References

[1] Find Windows 11 specs, features, and computer requirements, Microsoft
[2] WOW64 Implementation Details, Microsoft
[3] Microsoft Visual C++ Redistributable latest supported downloads, Microsoft

Related EE World content

Contending with Windows 10’s retirement: part 2
Engineers use AI/ML to improve test
What is an instrument driver and why do I need one?
Real-time vs. a standard operating system & How to choose an RTOS
Embedded security: Do you know what you don’t know?
How do firmware, system software, and application software work together?

The post Contending with Windows 10’s retirement: part 3 appeared first on Test & Measurement Tips.

NAMUR NE 43, from the German User Association of Automation Technology in Process Industries, is a widely adopted industry standard for defining fault levels in the 4-20 mA analog signals used in process automation, enabling the use of live zeros.

Defining fault signals is important in basic process control systems (BPCS), sometimes called a distributed control system (DCS), and in safety instrumented systems (SIS) (Figure 1). The BPCS controls and optimizes daily operations in continuous process industries like petrochemical refining, chemical manufacturing, paper and pulp production, and power generation. The SIS is a parallel system that protects personnel, the environment, and equipment from damaging and dangerous events, thereby enhancing safety and reducing risk.

NAMUR NE 43 basics

Most digital transmitters are compliant with NE 43. The standard establishes a basic structure for transmitter failure signals (A). The structure enables users to separate fault signals from process measurements (M). It’s designed to provide early warnings of faults to support preventative maintenance and safety, and keep the process operating for increased productivity.

Instead of 4-20 mA, NE 43 uses a 3.8 to 20.5 mA signal range for M with ≥21 mA or ≤3.6 mA, indicating some type of instrument failure. That provides guard bands of 0.5 mA at the top end and 0.2 mA at the bottom end, separating measurement information from failure alerts (Figure 2).

NAMUR nuances

NE 43 is intended to solve the dead zero problem, but it doesn’t define specific values for alarm and saturation levels; that’s left to individual equipment manufacturers. In fact, NE 43 defines a saturation zone, not a saturation level. That zone sits between 20.0 mA for nominal high values and establishes a buffer zone between 20.5 mA (minimum high saturation) and 20.8 mA (maximum high saturation) and a differentiation zone between 20.8 and 21.0 mA before reaching the high fault zone (Figure 3).

A similar set of ranges is established for low readings, with 4.0 mA being the nominal 0% reading. The minimum low fault is set at 3.8 mA, with the maximum low fault set at 3.6 mA, which establishes the lower differentiation zone. The maximum low fault measurement is 3.6 mA.

A key to NE 43 is the flexibility to accommodate different manufacturers, different types of transducers, and different sensor technologies. Note that while 3.6 mA is the low fault demarcation line, device makers are required to issue a low fault signal between 3.5 and 3.2 mA. And while 21.0 mA is the high fault threshold, device makers are required to issue a high fault signal between 21.5 and 22.8 mA.

There can be variations in definitions for different transducer types, even from the same manufacturer. For example, one manufacturer of both pressure and temperature measurement instruments defines the various (and very similar, but not identical) limit and threshold levels for specific models as follows:

• Pressure transducer: low saturation limit 3.9 mA, high saturation limit 20.8 mA, low alarm (fault) indication 3.5 mA, and high alarm (fault) indication 22.5 mA.
• Temperature transducer: low saturation limit 3.9 mA, high saturation limit 20.5 mA, low alarm (fault) indication 3.6 mA, and high alarm (fault) indication 22.5 mA.

Equipment designers need to understand the nuances of NE 43 to ensure proper operation. Using NE 43 compliant devices ensures that it will be possible to identify values that will work in each application without overlapping fault signals on the high or low ends. It does not relieve equipment designers from the requirement to ensure proper instrument integration into the final solution.

Summary

A dead zero occurs in a measurement system when the zero value corresponds to a reading of 0 mA from the sensor or transducer. That makes it impossible to distinguish between zero reading and system faults. The NAMUR NE 43 standard was developed to eliminate the dead zero problem in process control and safety systems. NE 43 is a tool designers can use when integrating instrumentation solutions.

References

NAMUR NE 43 Standards and Application Details (often missed), Orion Technical Solutions
The Difference Between 0–20mA and 4–20mA Current Output Transmitters, Zero Instrument
Understanding the Difference Between Live Zero and Dead Zero in 4 to 20 mA Signals, AutomationForum
What does NAMUR NE 43 do for me?, Lesman
What is Live Zero in 4-20 mA Current Loop?, Instrumentation Tools
Why a 0 mA Signal is Not Practical, Precision Digital
Why is 4-20 mA Current Used for Industrial Analog Sensors?, Control

Related EE World content

When are signal conditioners not needed with sensors?
What are simple ways to improve sensor performance?
What sensors are needed to fly hypersonic missiles?
What are the mathematics that enable sensor fusion?
What are the types and uses of position and angle sensors in an EV?

The post What’s the difference between live zero and dead zero? appeared first on Test & Measurement Tips.

In part 1 of this series, we looked at the dilemma posed by the retirement of Windows 10 on October 14, with Microsoft ceasing to provide support or free security patches to customers, including those who use the operating system for test and measurement systems.

As Noman Hussain, vice president of software and strategic business analysis at Pickering Interfaces, put it in part 1, upgrade decisions come down to the cost of change vs. the cost of remaining with the current system. Several factors contribute to the cost of change. To run Windows 11, for example, existing Windows 10 computers may require expensive hardware upgrades. In addition, Hussain noted that Windows 10 is the last version with native 32-bit support, so you may have difficulty using your collection of 32-bit dynamic link libraries in the 64-bit Windows 11 environment. Microsoft has tried to facilitate the 32-to-64-bit migration, which we’ll look at in part 3 of this series.

The decision of whether to migrate to Windows 11 or remain with Windows 10 depends on the specific aspects of your complex test system. However, Table 1 provides a checklist that you can use as an initial guide as you contemplate an upgrade. If you can check most of the boxes, remaining with Windows 10 may be a short-term or even long-term solution.

For example, if your current system is stable and your computer hardware won’t support Windows 11, you might be inclined not to upgrade. For security, if you are willing to pay for Microsoft’s Extended Security Updates program, you can buy yourself three years to plan a transition. Alternatively, if you can air-gap your system to isolate it from external networks and cloud-computing resources, and if you can prevent malware introduction through USB drives, you could remain with Windows 10 indefinitely.

However, Hussain pointed out an additional risk with this strategy. “It’s important to consider that a test system can include hundreds of thousands of ICs,” he said. “It only takes one IC becoming obsolete, reaching end-of-life, or requiring a last-time buy to trigger a chain reaction. The next thing you know, new products may be incompatible with the current system unless updated drivers are installed.” So if you’re committed to your Windows 10 system, you may want to ensure you have an adequate supply of spare parts on hand.

Indeed, drivers are a key consideration—you’ll need to check whether your vendors offer Windows 11 drivers for the equipment you have installed. According to Steve Summers, Director and Security Lead for Aerospace and Defense at Emerson Test and Measurement, the company “…maintains a close relationship with Microsoft so that we can begin testing on new platforms as soon as possible in Microsoft’s development process.” The company maintains an online Windows 11 compatibility guide.^[1]

When asked about driver compatibility, Toby Marsden, head of go-to-market strategy and strategic alliances at Keysight Technologies, said, “We develop our products and support customers based on industry demand to provide ongoing value. We assess the number of customers using our products and advise them to migrate to newer versions to gain productivity increases from the new features.” He added that Keysight would assess specific customer requests to remain on an older operating system version.

Similarly, third-party application software used in your system may not run under Windows 11. Conversely, noted Hussain, many third-party software vendors will stop releasing updates for their Windows 10 software versions, potentially exposing you to vulnerabilities or leaving you with tools that just won’t install any longer.

This last point encapsulates the risks and rewards of migrating to a new operating system. In the consensus view, upgrading is the best approach. “Not addressing the migration will ultimately lead to higher costs and lower business efficiency gains,” said Marsden.

Part 3 of this series will summarize the steps you can take to ensure a successful migration and meet any system revalidation requirements. Finally, we will conclude with general advice on planning for the inevitable obsolescence of Windows 11 or any other operating system you use in test-and-measurement equipment.

References

[1] Product Compatibility for Microsoft Windows 11, Emerson Test & Measurement

Related EE World content

What is an instrument driver and why do I need one?
What’s the difference between network security and physical security?
How modularity benefits test systems
As PCs Decline, Microsoft Betting Its Future on the Cloud
Microsoft to Show Off More Windows 10 Features
Measuring the Universal Serial Bus (USB)

The post Contending with Windows 10’s retirement: part 2 appeared first on Test & Measurement Tips.

Raleigh, NC, October 28, 2025 — Tektronix, well known for oscilloscopes and more recently power sources and power measurements through its acquisitions of Keithley and EA Elektro Automatic, hosted an event at the North Carolina Museum of Art. Attendees included representatives from distributors and customers, plus university professors, students, podcasters, and an editor.

The event, called “Precision Unveiled,” included a keynote address, exhibits of test equipment, and a workshop. I did not attend the workshop, choosing to maximize time at the equipment exhibits, as opposed to the art exhibits. See photos and video below.

In the keynote address, Joe Wang looked at the megatrends that have shaped our world since the mid-1800s. Wang’s slide shows how annual percentage point additions to gross domestic product (GDP) jumped when groundbreaking technologies, including railroads, automobiles, aircraft, and personal computers, became available. If you look at the right side of the graph, you see far greater jumps expected from AI, materials, batteries, and edge computing. That’s where opportunities for test equipment come in,” said Wang, “they all feed on each other.”

Wang continued by focusing on new energy sources. He spoke of batteries the size of tractor-trailer trucks that can store as much energy as contained in a lightning storm, and that a bottle of water, under the right conditions, can contain as much energy as a person could use in a lifetime.

Tektronix has a new parent company, Raleigh-based Ralliant, whose CEO, Tami Newcombe, and Tektronix president, Chris Bohn, then joined Wang onstage.

Bohn spoke of his vision of what engineers want in oscilloscopes. “All oscilloscope companies have specs,” he said, “the difference is in improving workflow.”

Newcombe, an EE and former Tektronix employee, added, “Engineers look to us to help them solve problems.”

Next, Chris White took to the stage, where he spoke about AI, calling it the fifth wave of computing following the PC, internet, cell phone, and cloud computing. He augmented some of Bohn’s comments by adding that the user experience is key and not just by using colors, graphs, and layout. He referred to test equipment taking on an experience similar to the smartphone, but with added automation. By that, he referred to APIs and Python scripts running on an external PC.

White continued by speaking of intelligence at the network edge and how engineers use oscilloscopes to troubleshoot and validate the signal integrity needed for networks to reliably transport data. “Low-cost computing is going into the consumer market,” he said. “Engineers must work to minimize cost, size, and power consumption. The behavioral boundaries cover more than just performance.”

Exhibits

The Tektronix 7-Series oscilloscopes were prominently displayed. Here, it’s demonstrating eye diagrams, a key tool for analyzing signal integrity in data streams. On another table, the acquisition board was on display. You can see the shield traces surrounding the sensitive analog circuits with the signal processor in the center and power components at the far end of the board. The board’s code name was “Falcon,” and the PCB designer left a telltale sign.

Tektronix also displayed a 6-Series oscilloscope. Here’s an application that uses a near-field probe to locate a source of EMI emissions.

Another demonstration focused on power sources and measurements. In the video, Steven Everitt demonstrated the MP5000, a modular rack-mounted system where you can install up to three power supplies or SMUs. The modular SMUs are essentially faceless versions of the classic Keithley 2600. Surprisingly, Everitt noted that engineers are using the system, which was designed for production test, on their benches to achieve high channel counts.

Following the demonstration, I spoke with Rich McFadden and Jennifer Cheney about the MP5000. McFadden noted that a reason for designing it had to do with the need for high channel counts. He cited LED testing on flat-screen TV as an example.

“Every LED needs to be tested. Think of the volume and throughput needed to test every LED in a TV. Test systems take up so much floor space. You could have numerous manufacturing lines just full of those test racks. We needed to design a 1U-size instrument where engineers can stack them in a rack.”

Cheney noted that the MP5000 uses Keithley’s test-script processor, which lets test engineers daisy chain up to 32 MP5000 chassis together, each capable of supporting up to three power supply or SMU modules. Test scripts can run in each chassis, or a host computer can execute them. Often, test engineers use a host computer for data collection and analysis.

Other exhibits included measurements for sensor design and for materials testing. Yes, there were art exhibits.

The post Tektronix event highlights precision measurements appeared first on Test & Measurement Tips.

The test-and-measurement industry has long leveraged consumer technology to take advantage of economies of scale and lower costs. Consequently, it’s not uncommon to see sophisticated test and data-acquisition systems incorporating Windows computers. This approach works well until interrupted by inevitable discontinuities in the trajectory of Windows, with each version having a lifespan measured in years, compared with test systems’ lifetimes measured in decades.

Such a discontinuity is upon us now. Effective October 14, Microsoft ended technical support and free security updates^[1] for Windows 10, which debuted July 29, 2015.^[2] Windows 10 joins Windows XP^[3], Windows Vista,^[4] Windows 7,^[5] and Windows 8,^[6] which have all reached end-of-life, as illustrated in the Figure 1 timeline. Note that there was no Windows 9.

The end of Windows 10 support and free security upgrades presents a dilemma to those using the operating system to control test and measurement systems. The obvious step is simply to upgrade to Windows 11, but that may not be optimal. According to Toby Marsden, head of go-to-market strategy and strategic alliances at Keysight Technologies, if a system is stable, migration to a new operating system risks a regression that can cause core business processes to fail, and therefore a customer may choose not to upgrade.

Steve Summers, director and security lead for aerospace and defense at Emerson Test and Measurement, added another perspective, noting that customers upgrading to Windows 11 would see few if any new features that specifically benefit test-and-measurement equipment. “Windows 10 is sufficiently stable, has the features needed, and works great for test systems,” he said. But there is a key driver favoring the upgrade. “The lack of support is the forcing function that is making test systems migrate to Windows 11,” Summers noted.

Noman Hussain, vice president of software and strategic business analysis at Pickering Interfaces, said his customers’ biggest concern with the retirement of Windows 10 is the lack of security updates, which makes systems susceptible to malware attacks. “It always comes down to the cost of change versus the cost of continuing to run the current system,” Hussain said. “I still have customers using Windows XP in air-gapped environments [that is, disconnected from external networks]. For them, the cost of change is higher than the perceived risk of keeping their existing systems running—but as you can imagine, those PCs are now a nightmare to support.”

In addition, air-gap security is far from perfect. Summers said, “If you look at the attacks on operational systems over the past 10 years, many of them have been on air-gapped systems. It’s just not a solid security defense. More effective is to move to a zero-trust defense strategy, where you have multiple defenses at every node in the system, so that if the air gap is breached, there are still defenses to protect the systems.”

Microsoft is offering a temporary reprieve with regard to security. It offers for purchase an Extended Security Updates (ESU) program for Windows 10.^[7] The price begins at $61 per device for the first year, with the price doubling each successive year. The ESU is planned to last a total of three years, giving you time to plan a smooth migration to Windows 11 (as shown in the figure, Microsoft offered a similar ESU program for Windows 7).

In subsequent parts of this series, we’ll describe steps you can follow if you plan to stay with Windows 10, whether or not you plan to purchase the ESU. Then, we’ll describe steps you can take if you plan to immediately upgrade to Windows 11. And finally, we’ll discuss how to plan for the inevitable demise of Windows 11 or any other operating system you plan to use in test-and-measurement systems.

Continue to part 2.

References

[1] Windows 10 support ends on October 14, 2025, Microsoft
[2] Windows 10 Home and Pro, Microsoft
[3] Windows XP, Microsoft
[4] Windows Vista, Microsoft
[5] Windows 7, Microsoft
[6] Windows 8, Microsoft
[7] Extended Security Updates (ESU) program for Windows 10, Microsoft

Related EE World content

Sorting out PC-based instrumentation
Work-at-home tools: PC-based instruments address demanding applications
Microsoft to Give Vista Buyers Free Windows 7
Developments with Microsoft’s New Windows 8 System
Microsoft Skips Windows 9 to Emphasize Advances
Microsoft launches Windows 10: Here’s what that means

The post Contending with Windows 10’s retirement: part 1 appeared first on Test & Measurement Tips.

Congratulations to the LEAP Awards for Test & Measurement winners, who are profiled here:

Gold

Liquid Instruments
Moku:Delta

Liquid Instruments’ Moku:Delta is the first test and measurement device to introduce Generative Instrumentation, combining an advanced multi-instrument test platform with the power of agentic AI. This entirely new approach to test enables engineers to design and deploy custom instruments and automatically configure complex systems with natural language prompts. Instead of hours spent manually optimizing numerous parameters or creating custom code, users simply describe what they want, and Moku:Delta with Generative Instrumentation configures itself. Beyond Generative Instrumentation, Moku:Delta delivers the highest-resolution, 2 GHz Oscilloscope and the only Spectrum Analyzer providing full 2 GHz bandwidth down to 0 Hz for precise 1/f noise measurements to fully characterize device performance. The platform also includes the highest-channel count, ultra-low noise, microwave Lock-in Amplifier for advanced research. Moku:Delta is designed to deliver a level of performance previously unseen in software-defined instrumentation. At its core, it offers 2 GHz analog bandwidth, eight analog inputs and eight outputs at up to 5 GSa/s, and 32 high-speed digital I/O channels. It supports a reconfigurable suite of 15 standard instruments and can run up to eight of them simultaneously, delivering over 2 billion possible configurations in a compact 2U form factor. This flexibility and performance empower users to respond to changing requirements and vastly improve productivity for applications like quantum research, semiconductor validation, and aerospace and defense. Supercharged by Generative Instrumentation, Moku:Delta both simplifies everyday tasks and enables entirely new capabilities on demand, unlocking previously impossible test solutions.

Silver

Pico Technology
PicoScope 3000E Oscilloscope

The PicoScope 3000E Series redefines PC-based oscilloscopes with up to 500 MHz bandwidth and 5 GS/s in a compact, portable, USB-powered package. PicoScope 3000E oscilloscopes are small, portable, and deliver high-performance specifications ideal for engineers working on advanced electronics and diverse embedded system technologies, either in the laboratory or on the move. Offering four analog channels with a choice of 100 MHz, 200 MHz, 350 MHz, and 500 MHz bandwidth options, plus 16 digital logic analyzer channels on MSO models, there’s a model to suit every application and budget. Supported by the advanced PicoScope 7 Test and Measurement software, the PicoScope 3000E Series enables the rapid, cost-effective debugging and performance validation of complex analog and power electronic designs. It also offers an ideal package for many other applications, including embedded systems design, research, testing, education, service, and repair. Where power and performance meet portability Up to 500 MHz, 5 GS/s, USB powered PC oscilloscopes and MSOs 100, 200, 350 or 500 MHz with 5 GS/s Up to 10-bit resolution (14 bits using enhanced resolution) 2 GS ultra-deep capture memory 16 digital channels (on MSO models) Function/arbitrary waveform generator included Compact, portable and USB powered Over 40 serial protocol decoders included as standard Segmented memory, persistence and fast waveform updates Advanced math, measurements, masks and digital triggering PicoScope 7 for Windows®, macOS® & Linux® with free updates Support for LabView®, MATLAB® and writing your own code 5-year warranty and free technical support.

Bronze

ITECH ELECTRONICS
IT2700 Multi-channel Modular Power System

The ITECH IT2700 series multi-channel modular power system, as a next-generation power testing solution, is designed to provide engineers with more flexible and efficient testing options, driving advancements in power electronics and battery technology. The IT2700 multi-channel modular power system integrates high precision, multifunctionality, and modularity, taking innovation to the next level. It is ideal for R&D and production lines, enhancing productivity, improving energy efficiency, and actively contributing to global green and low-carbon development. The IT2700 series multi-channel modular power system brings ultra-high power density. The 1U main frame can include up to 8 modules (200W each) or 4 modules (500W each). Different modules can be grouped and synchronized. The modules could be bidirectional DC power supplies, DC power supplies, or regenerative loads. They have built-in LAN, USB, CAN, digital I/O, and free PC software. It can be widely used in ATE integration in R&D, design verification, and manufacturing of DC-DC converters, communication power semiconductors, 3C products, like smartphones, PCBA, battery simulation and test, chips, BMS chips, etc. The IT2700 Series next-generation power system enables ITECH to offer more competitive testing solutions in bidirectional power and multi-channel testing. The launch of the IT2700 undoubtedly fills a gap in the market, providing strong support for efficient testing in R&D and production lines. Its high integration and diverse functionalities are set to revolutionize the field of power testing.

The post Design World presents the 2025 LEAP Awards Winners: Test & Measurement appeared first on Test & Measurement Tips.

This interference does not just travel in one way. Figure 1 illustrates the four primary methods by which EMI can travel from its source to a victim device:

1. Conducted EMI, as the name suggests, is interference that travels along physical conductors. Figure 1 shows this as noise moving along the wire connecting the two devices. This is common in power cords and data cables, where the noise from one device is directly fed into another.
2. Radiated EMI is interference that travels through the air (or space) as electromagnetic waves, much like radio or Wi-Fi signals. The “Source Device” acts as a tiny, unintentional radio transmitter, broadcasting noise that the “Device Under Test” picks up like an antenna.
3. Inductive EMI is a near-field effect also known as magnetic coupling. As shown by the transformer symbol, a changing electric current in the source creates a changing magnetic field, which in turn induces an unwanted current in the nearby circuits of the test device.
4. Capacitive EMI is the other near-field effect, also known as electric field coupling. As represented by the capacitor symbol, a changing voltage (an electric field) on the source device can couple with the test device, inducing an unwanted voltage and disrupting its signals.

Understanding these four pathways is the first step in achieving Electromagnetic Compatibility (EMC), the goal of designing electronics that can function correctly without either causing or being affected by EMI.

How can EMI be prevented?

Preventing EMI is a core part of the entire electronics design process and involves three main strategies: reducing the noise at its source, blocking its transmission path, and making the receiving device less sensitive.

1. Reducing the source is the best approach to create less noise from the start. This can be done by choosing electronic topologies (circuit designs) that are inherently quieter, such as quasi-resonant flyback converters. Another method is softer switching, which involves slowing down the fast voltage and current changes that are the root cause of EMI.

This concept is visually represented in Figure 2. The “Hard switching” (red line) path shows high voltage and current at the same time, creating a large, abrupt energy spike that generates significant EMI. In contrast, “Soft switching” (blue line) ensures that either voltage or current is near zero during the switch, resulting in a smooth transition that drastically reduces the source noise.

2. Block the path—if noise is generated, the next step is to stop it from traveling. This can be achieved in the following ways:
   • For conducted EMI, the solution is an EMI Filter, which is illustrated on the left side of Figure 3. The filter is placed on the power line, taking in the signal + noise and diverting the unwanted noise to an electrical ground, allowing only the clean signal to pass.
   • The right side of Figure 3 shows what this looks like on a real circuit board. The copper-wound coils (chokes) and gray blocks (capacitors) are the physical components of a two-stage filter designed specifically to block this conducted interference.
   • For radiated EMI, the paths are blocked using physical layout and shielding. This includes making high-current loops on the circuit board as small as possible to reduce magnetic (inductive) coupling. It also involves placing components at curved angles or 45° bends to each other and using metal shields to block electromagnetic waves, just like a Faraday cage.

3. Hardening the receiver is the final strategy to make the “Device Under Test” immune to any noise that still gets through. This is achieved by designing sensitive circuits with low impedance (making them harder to disrupt) or by using differential signaling. Differential signaling works by detecting the voltage difference between two complementary signals, effectively rejecting noise that affects both signals equally.

Summary

EMI is an unavoidable side-effect of modern electronics, traveling as unwanted noise through conductive, radiative, inductive, and capacitive paths. Effectively managing it requires a comprehensive approach: designing quieter circuits from the start, using filters and shielding to block transmission, and building robust devices that can ignore interference.

References

EMC Considerations for Auxiliary Inverters in Electric Vehicles Applications, KEB Automation KG
Electromagnetic Interference (EMI) in Power Supplies, EMC FastPass
A Soft-Switching SEPIC with Multi-Output Sources, MDPI
Electromagnetic Interference, HardwareBee

EE World related content

EMC/EMI design and the use of board-mount dc/dc converters
How can power converters be designed to minimize EMI?
Safety capacitors for EMI filtering and voltage isolation
Choking off EMI/RFI in off-line switchers
What are some common EMI/EMC tests?
EMI control for power and signal lines

The post EMI: what it is and how to keep it in check appeared first on Test & Measurement Tips.

If you design, test, or troubleshoot circuits, you surely need an oscilloscope, and two eight-bit channels are not enough anymore. Indeed, the MXO 3 oscilloscope comes in four-or-eight channels with 12-bit vertical resolution, letting you see more of your analog or digital circuits on the display. In HD mode, the oscilloscope has up to 18 bits of vertical resolution.

The MXO 3 replaces the RTM 3000 series and brings features from MXO 4 and MXO 5 series to a lower-grade maximum bandwidth. It’s available at bandwidths that cover the low-to-midrange — 100 MHz, 250 MHz, 350 MHz, 500 MHz, and 1 GHz — making it suitable for analog circuits such as audio amplifiers and filters or for embedded/IoT circuits. The bandwidth is field upgradable. Standard acquisition memory is 125 Msamples standard, 500 Msamples optional. The MXO 3 is a lower-cost version of the MXO 5 (also 4 or 8 channels), which tops out at 2 GHz bandwidth and 1 Gsamples of data memory. Compare the MXO 3 to the MXO 4 and MXO 5 here.

Rohde & Schwarz included features that can save you time and help you catch signal anomalies that the company says other oscilloscopes might miss. For example, it features a screen update rate of 4.5 million updates/sec, which matches that of the MXO 4 and MXO 5, and which Rohde & Schwarz claims is the fastest in the industry. That speed means you can see up to 99% of all signal events.

Then there’s triggering. After all, if you can’t trigger your oscilloscope on a specific event, you’ll likely not see it at all. To that end, the MXO 3 uses hardware-accelerated triggering found in a custom ASIC that can rearm in 21 ns. As you’d expect, the MXO 3 uses common triggers such as level, pulse width, and runt, but it also lets you trigger on math such as power (voltage × current) calculations.

Because so many engineers use oscilloscopes to look at signals in the frequency domain, the MXO 3 lets you trigger on occurrences that are easily visible there but hard to see in the time domain. That can help you identify problems such as EMI issues. Besides setting a level trigger, you can use a zone trigger to capture an acquisition should a frequency spike occur in a particular range. You can also initiate a trigger should an event not occur at a specified time or at a specified frequency.

With its small footprint and and 11.6 inch (30 cm) display, the MXO 3 takes up little bench space yet still provides plenty of viewing area. If you have no available bench space whatsoever, you can mount the instrument using its VESA mounting holes.

Options include:

• 50 MHz single-channel arbitrary waveform generator
• A wide array of probes, including current probes, differential probes, and EMI probes
• 500 Msamples of acquisition memory (125 Msamples standard)
• Power analysis
• Bode plot frequency response analysis
• 16 digital-logic inputs
• Bus decoding that includes I²C, SPI, UART, and automotive Ethernet 10Base-T1S

The post Midrange oscilloscopes focus on speed and resolution appeared first on Test & Measurement Tips.