Electronics EveryWhere

Semi Conductor Tester - SemTest Part 4

noreply@blogger.com (Administrator) — Thu, 29 May 2014 10:10:00 +0000

Testing was done in part 3, this last part is consist of following:

Final assembly

The front panel assembly can now be lowered down onto the case. Make sure that the three ribbon cables are folded neatly into the space above the lower PCB and not caught between the edges of the case or lid. Fasten the case together with four M4 screws into the corner holes, then fit the knobs to the rotary switch and the pot and the assembly is complete.

Testing the HV crowbar

It’s now necessary to check that the HV crowbar circuit is working correctly. To do this, power up the unit, wait a few seconds and then press the Menu Select button. You will get a display like this:

Device to Test:▲

1:Diode/Zener▼

Press Enter and then the Up button. The display will then show:

Test parameter:▲

Irev(OPV)▼

Press Enter again. Set the Device Operating Voltage to 25V, using the right-hand knob. Then press the Test On/Off button to start the test.

Now carefully measure the voltage across the top and bottom A and K terminals in the diodes and LEDs section of the test socket. You should get a reading close to 25V. If it’s much lower (say, 12V) then either the crowbar circuit has triggered prematurely or there is a fault in the DC/DC converter circuit. You will need to switch off, open up the unit and check the crowbar and converter circuits for faults, such as incorrectly oriented components.

If you get a much higher reading than 25V, there is a problem with the DC/DC converter section. Switch off and measure the voltage across the A and K terminals until it drops to a safe level. Then open the unit up and look for the source of the problem.

Assuming all is well, press the Test On/Off button to terminate the test. You can now do a high-voltage test. The procedure is similar to before, except you want to do an IREV(BV) test. So when you get to this stage:

Device to Test:▲

1:Diode/Zener▼

press enter twice and start the test.

Carefully measure the voltage across the A and K terminals again. It should be several hundred volts and it will rise to close to 600V after a number of seconds. Now press the Test On/Off button again to terminate the test while monitoring the voltage between the A and K terminals. It should immediately fall to just a few volts when the test is terminated.

If it remains high and only decreases slowly, the crowbar has failed to operate and you will need to wait for the capacitors to discharge before opening the unit up and checking for faults.

If the crowbar is not working (eg, if it fails), a warning will be displayed on the LCD immediately after performing a high-voltage test. This indicates that there is still a high voltage present at the test socket. If you get this warning then you should open the unit up and repair the crowbar circuit.

HV DC CROWBAR

An add-on crowbar module instantly kills the high voltage applied to the ZIF socket at the conclusion of any breakdown voltage test. As a further safety measure, it also kills the high voltage in the event that the SemTest is inadvertently turned off before a test has properly concluded.

This minimises the chance of the user getting a shock from the test terminals when removing the DUT or a possible breakdown of the DUT itself when the power is inadvertently removed.

The crowbar module is wired to three points on the main (lower) SemTest PCB. On our prototype, these wires have been soldered to specific component leads, but the final

SemTest PCB has pads for these wires.

The crowbar board senses the 11.4V supply rail to the MC34063 DC/DC converter IC1. This drops very quickly to around 6V when a test finishes or more slowly if the unit is switched off during a test. Either way, this is the trigger for the crowbar to discharge the capacitor bank from 600V to a few volts in around 20ms.

Circuit description

The full crowbar circuit is shown in Fig.7. It could potentially be used in other devices, but for use with the SemTest, link LK1 is installed, to short Vin (the sense input) and V+ (its power supply) together.

The +HV and GND terminals at CON1 are connected across the SemTest’s high voltage capacitor bank. Fig.8 shows a fragment of the SemTest circuit and demonstrates how the crowbar module is connected. The V+ terminal goes to pin 6 of IC1, which is at around +11.4V when the DC/DC converter is running and drops to 0V when it is switched off.

While the DC/DC converter is running, current flows from this rail, through diode D1, charging the 100μF capacitor. As this capacitor charges, the gate (AG) of programmable

unijunction transistor PUT1 is pulled up too, via the 10kΩ and 100kΩ resistors.

At the same time, the anode (A) is pulled up via a 330Ω resistor. The 10nF capacitor between PUT1’s anode and gate is initially discharged and this helps to keep the gate at

anode potential, preventing false triggering if there are any initial glitches in IC1’s power supply (eg, due to relay contact bounce).

A PUT is essentially a small anodegate SCR. While a conventional SCR is turned on when its gate is pulled above its cathode (K), a PUT turns on when its gate is pulled below its anode, sinking current from the gate. Both SCRs and PUTs remain on once triggered, until their anode-cathode current flow drops below the ‘holding’ current, in this case much less than a milliamp.

As long as V+/Vin are held at around 11.4V, the crowbar circuit remains deactivated. But once Vin drops precipitously, the 10nF capacitor begins to charge while the 100μF capacitor retains its charge, by virtue of diode D1.

Once Vin drops below the ~6V threshold, sufficient current flows from PUT1’s gate to trigger it on. It then dumps the charge in the 100μF capacitor into SCR1’s gate (KG), via the 330Ω current-limiting resistor. This happens in less than 100μs if Vin drops fast, as when a test ends normally.

The 330Ω resistor limits the current into SCR1’s gate to around 25 to 30mA, enough to trigger it reliably. SCR1 then rapidly discharges the high voltage capacitor bank through

the 100Ω resistor. The peak discharge current is 600V/100Ω = 6A.

PUT1 switches off as soon as it has finished dumping the charge of the 100μF cap into SCR1’s gate. But SCR1 stays on until the current through it drops below 40mA (its holding current) so the capacitor bank discharges to around 4V.

The specified TYN816 SCR is rated for 800V and 16A. Do not use an SCR with lower ratings.

Construction and testing

Refer to the overlay diagram, Fig.9. Fit the two small resistors first, followed by diode D1, with its cathode stripe towards the right side of the board. Use a lead off-cut for link LK1 and solder it in place. Then install the two 1W resistors.

Wiggle the middle lead of SCR1 back and forth until it snaps off. If there is any lead remaining, remove it with side-cutters. Bend the remaining two leads down and insert

them through the holes on the PCB, then use the machine screw to attach the metal tab with a shakeproof washer, both under the screw head and under the nut. Do it up tightly since the screw conducts the current when the crowbar activates. Then solder the two pins.

Fit the 10nF capacitor and then PUT1, bending its leads out with pliers to suit the pad spacing. Push it down as far as it will go before soldering and trimming the leads. Next, mount the 100μF capacitor, with its longer (positive) lead towards the left side of the PCB.

Bend its leads so that it lays down flat on the board before soldering them – see photo.

Don’t fit a terminal block for CON1, since we have limited clearance to fit the unit into the SemTest. Instead, solder a red wire to HV, a yellow wire to V+ and a black wire to 0V. Make sure there are no stray copper strands.

Wire the unit up to the SemTest as shown in the main overlay diagram (previous parts). Trim each lead so that you don’t have a lot of extra length. The photos show the best place to fit it.

Once it’s wired up, slip the crowbar module into the heatshrink tubing and apply gentle heat. Make sure there is no exposed metal when you are finished. Some silicone sealant can then be used to hold the unit in place, so it doesn’t rattle around inside the case.

Once the SemTest unit itself is complete, the HV crowbar must now be tested for correct operation, as described in the main article.

Using the SemTest

The SemTest is used as follows:

STEP 1: place DUT in ZIF socket and switch on.

STEP 2: Press Menu Select.

STEP 3: Use Up/Down buttons to select device type and press Enter.

STEP 4: Use Up/Down buttons to select test and press Enter.

STEP 5: For OPV tests, use right-hand knob to select test voltage.

STEP 6: Press Test On/Off to start test (red LED on) and read result.

STEP 7: Press Test On/Off again to finish test (red LED out).

STEP 8: check red LED is out and there is no high voltage warning on the LCD before removing DUT.

Exercise caution when testing components for high-voltage breakdown. Up to 600V DC is present on the device leads during such tests, so be careful not to touch them!

The biggest problem in using the SemTest is knowing the various lead configurations of the devices it can test. To that end, we have prepared a connections chart showing commonly used diodes, LEDs, BJTs, MOSFETs, SCRs and PUTs. It can be stuck on a wall or to the underside of the SemTest case for easy reference.

For less common devices, you’ll need to look up the connections in a data book or by downloading a data sheet from the manufacturer’s website.

Finally, here are a few tips to guide you when you’re doing some of the more specific tests:

When reading the forward voltage drop VF of a diode or LED, or the voltage drop VAK of an SCR when it’s conducting, be aware that the accuracy of this measurement is not very high due to measuring circuit limitations. So if you need to make really accurate measurements of VF or VAK, you’ll need to use an external DMM with its leads connected across the device’s ‘A’ and ‘K’ leads.
Remember that during the same tests, it’s OK to increase the device operating voltage to a higher setting in order to see the voltage drop at higher current levels.

When you want to measure the hFE of a BJT, start on the setting with the lowest IBIAS level (ie, 20μA), because this is the setting with the highest hFE range. Only swing down to one of the higher IBIAS settings if the hFE reading you get is very low (ie, below 300). This should only be necessary with medium-to-higher power devices, which often have their ‘peak’ hFE at higher currents.

When you want to measure the IDS vs VGS characteristic of a MOSFET to get an idea of its transconductance or ‘gm’, start by selecting the highest device operating voltage which will not exceed the device’s VDS ratings. That’s because the VGS bias voltage (adjusted via VR10) is derived from he actual device operating voltage, which inevitably tends to drop once the device begins to draw drainsource current (due to voltage drop in the current-limiting resistors).

If you don’t set the switch for a reasonably high voltage to start with, you’ll find that it won’t be possible to provide much VGS once the device starts to conduct.

Actually, although you need to set the operating voltage within the device ratings when you start this test, it’s OK to increase the setting to 100V during the test itself, if you need to do so in order to achieve a higher VGS.

This won’t cause any problems if you only increase the voltage setting once the device is conducting.

EPE

Downloads: PCB n code files

Infrared audio headphone link for TV

noreply@blogger.com (Anonymous) — Fri, 11 Jan 2013 17:04:00 +0000

To listen via headphones preferably good ‘surround your ears’ muff-type headphones, which not only deliver the wanted sounds directly to your ears and hearing aid(s) but also cut back the competing sounds at the same time. If you pick the right kind of headphones, with some acoustic damping in the earmuffs, they don’t cause your hearing aid(s) to emit feedback and whistle sounds either.

The result is comfortable listening at a volume level that’s right for you, where you can hear and understand everything that’s being said.

Headphone jack
Some TV sets do have earphone jacks, so you could simply fit a pair of stereo headphones with their own volume control (if necessary), plus a long cord and plug to mate with the jack on the TV. But many sets do not have a headphone jack, and many that do have it wired so that when headphones are plugged in, the speakers are disabled.

That’s fine for you, but no good for everyone else. In any case, being hooked up to the TV via a long cable has its own problems: you can forget to take ’em off when you get up for a comfort break or someone else can trip on the cable when they move about the room.

Cordless headphones
A much better solution is to use ‘cordless’ headphones, either via a UHF or infrared link. This means that you have a transmitter or sender unit that sits on the top of the TV, plus a small battery-operated receiver to drive the headphones at your end.

Of course, IR-linked cordless headphones are available commercially and these can give you some improvement. But there are drawbacks, the main one being that the receiver unit is built into the actual earphones and/or their headband, so it can’t be used with any other headphones. That means you’re stuck with the ones you get and in most cases, they are not the ‘surround-your-ears’ muff type. Nor do they have any acoustic damping.

As a result, you not only have to throttle back your hearing aid to stop it from whistling, but also the headphones allow quite a lot of competing sounds to enter as well.

So that’s the reasoning behind the development of this project – by building it, you get to choose the best type of headphones. However, there is one more feature – it works in mono only. This has been done deliberately because stereo sound is a real drawback to those who have trouble making out speech from the TV.

This applies particularly to those films, documentaries and sportcasts where there is a lot of background music or other sounds. By using a mix of the left and right channels, we cancel most of these extraneous sounds, making the speech much easier to discern. In addition, we have applied a small amount of treble boost to the audio signal, which further improves intelligibility on speech – see Fig.6.

There’s one more bonus with using mono sound – it also simplifies the circuit considerably.

How it works
The method of transmission is simple and effective. Basically, the signal is transmitted using pulsewidth modulation (or PWM). This converts the audio signal directly into a pulse stream of constant frequency, but with the pulse width varying with the instantaneous amplitude of the audio signal.

Fig.1(a) shows the method. First, the left and right stereo signals are mixed together to give a mono signal. This signal is then passed through an input amplifier stage (IC1b) and then via a four-pole low-pass filter (IC1a and IC4a), which sharply rolls off the response just above 12kHz.

This is done for two reasons. First, if you are partially deaf, signals above 12kHz are not much use anyway. Second, it prevents any spurious ‘alias’ signals from being generated during the digital modulation process – which is equivalent to digital sampling. We are using a fairly high sampling frequency of about 90kHz which tends to reduce aliasing, but the low-pass filtering is also worthwhile, because it ensures that virtually no signal frequencies above 15kHz are fed to the modulator.

Sampling signal
Next, the audio is fed directly to the non-inverting input of a comparator (IC5) where it is compared with a 90kHz triangular wave ‘sampling’ signal on the inverting input. This 90kHz triangular wave signal is generated by feeding a 180kHz clock signal into a D-type flip-flop.

This then produces a very symmetrical square-wave signal at half the clock frequency, or 90kHz. This 90kHz signal is buffered and fed through an active integrator stage, which converts it into a linear and very symmetrical triangular wave.

But how does the comparator use this 90kHz triangular wave to convert the audio signal into a PWM stream? To see how this works, take a look at the waveforms of Fig.2. Here the green sinewave represents the audio signal fed to the positive input of the comparator,
while the higher frequency red triangular wave shows the sampling signal fed to the comparator’s negative input.

In operation, the comparator’s output is high when the audio signal level is higher than the 90kHz sampling signal. Conversely, the comparator’s output is low when the sampling signal’s level is the higher of the two. A switching transition occurs when ever the two waveforms cross. The resulting PWM output waveform from the comparator is shown as the lower black waveform.

Note that the comparator output is a stream of 90kHz pulses, with the pulse widths varying in direct proportion to the audio signal amplitude. The average value of the pulse stream is directly proportional to the instantaneous value of the incoming audio, as shown by the dark blue dashed curve.

Referring back to Fig.1, this PWM pulse stream is fed to a PNP switching transistor, which drives a string of IR-emitting LEDs. As a result, the digitised audio is converted into a stream of IR light pulses, directed towards the receiver unit.

Receiver block diagram
The receiver is even simpler than the transmitter because of the fact that the average value of the PWM pulse stream varies in direct proportion to the audio modulation.

As shown in Fig.1(b), a silicon PIN photodiode is used to detect the IR pulse stream from the transmitter. Its output current is then passed through a current-to-voltage (I/V) converter and amplifier stage (IC1b and IC1a) to boost its level. The resulting pulse waveform is then fed through a limiter stage (IC2) to produce a stream of clean, rectangular pulses of constant amplitude.

Next, the pulses are fed through a multi-stage low-pass filter (IC3b and IC3a) to remove all traces of the 90kHz sampling/modulating signal. This simply leaves the audio signal which was carried in the average signal level of the pulses.

From there, the recovered audio passes to a volume control pot and finally to a small audio amplifier (IC4) to drive the headphones.

Power for the receiver circuit comes from four AA cells, which can be of either alkaline or NiMH rechargeables.

Circuit description
Refer now to the full circuit for the transmitter – see Fig.3. As shown, the incoming line level stereo signals are mixed together using two 47kohm resistors, while trimpot VR1 sets the level. The resulting mono signal is then fed to op amp stage IC1b, which operates with a gain of 23, as set by the 22kohm and 1kohm feedback resistors.

Next, the signal is passed through op amps IC1a and IC4a, which form a 4-pole low-pass filter (or two 2-pole active filters in cascade, to be more precise). Together, these roll off the response above 12kHz. The filtered signal then emerges from pin 1 of IC4a and is fed directly to the non-inverting input of comparator IC5.

The 180kHz ‘twice sampling clock’ signal is generated by IC2b, a 4093B CMOS Schmitt NAND gate wired as a simple relaxation oscillator. A 12kohm resistor and 680pF capacitor set the operating frequency. This is not particularly critical, although for best performance it should be between 160kHz and 200kHz (corresponding to a sample frequency of 80 to 100kHz).

Flipflop stage IC3a is used to divide the clock pulses by two and generate the symmetrical 90kHz square wave. Its output at pin 1 is then passed through Schmitt NAND gates IC2a, IC2c and IC2d which are connected in parallel as a buffer. The buffer output is then coupled via a 100nF capacitor to op amp IC4b.

IC4b is configured as an active integrator to convert the 90kHz squarewave into a linear symmetrical triangular waveform of the same frequency. This triangular wave is then fed directly to the inverting input of comparator IC5, to sample and convert the audio signal into the PWM pulse stream.

IC5’s PWM output appears at pin 7 and is used to drive transistor Q1 (BC328). This in turn drives series-connected infrared LEDs (LEDs 1 to 3 and LEDs 5 to 7), along with LED4 (green) which serves as a ‘power on’ indicator. The 47ohm resistor in series with the LED
string limits the peak pulse current to around 45mA, resulting in an average current drain for the complete transmitter circuit of about 25mA.

Transmitter power supply
Power for the transmitter circuit is derived from a 12V AC or 15V DC plugpack. This feeds diode bridge D1 to D4, which rectifies the output from an AC plugpack. Alternatively, the bridge rectifier allows a DC plugpack to be used with either polarity.

The output from the bridge rectifier is filtered using a 1000uF capacitor and then fed to 3-terminal voltage regulator REG1 to produce a 12V DC supply rail.

Receiver circuit
OK, so much for the transmitter circuit. Now let’s take a look at the receiver circuit – see Fig.4.

In operation, the transmitted PWM infrared signal is picked up by PIN photodiode PD1 (BP104). This device produces output current pulses in response to the incoming IR signals and these are then fed to the inverting input (pin 6) of op amp IC1b. The non-inverting input (pin 5) of IC1b is biased to half-supply (ie, 3V) by two 22kohm resistors connected in series across the 6V supply rail.

IC1b operates as an active I/V (current-to-voltage) converter. In operation, it converts the input current pulses to voltage pulses, which appear at its pin 7 output. These pulses are then coupled via a 2.2nF capacitor to op amp stage IC1a, which operates with a gain of –10. The resulting amplified output pulses appear at pin 1 and are fed directly to pin 3 of IC2.

IC2 is an LM311 comparator and is used here as the limiter. Note that its non-inverting input (pin 2) is biased to half the supply voltage using the same voltage divider (2 × 22kohm resistors) that’s used to bias IC1a and IC1b. This ensures that the pulses from IC1a are compared with a voltage level corresponding to their own average DC level. And that in turn ensures that the limiter ‘squares up’ the pulse stream in a symmetrical fashion.

In addition, the 2.2Mohm feedback resistor and the 10kohm resistor in series with the bias for IC2 together provide a small amount of positive feedback hysteresis, to ensure clean switching. Because the LM311’s output (pin 7) is an open collector, it must be provided with a resistive pull-up load. This is provided by power-on indicator LED1, together with its 390ohm series resistor.

The restored PWM pulse stream appears at pin 7 of IC2 and is then fed through the receiver’s low-pass filter circuitry. This comprises passive 47kohm/180pF and 100kohm/100pF RC filter stages, voltage follower IC3b, active low-pass filter stage IC3a and
finally, a 4.7uF coupling capacitor and a 1kohm/10nF passive filter connecting to the top of volume control VR1.

As a result, the signal appearing across VR1 is a very clean replica of the original audio signal fed into the transmitter unit.

IC4 is the audio amplifier output stage and is based on an LM386N. It amplifies the signal from the volume control (VR1) and drives a stereo phone jack via a pair of 33ohm current limiting resistors (one to the tip and one to the ring).

Finally, the receiver is powered from a 6V battery consisting of four AA cells connected in series. These cells can be either standard alkaline primary cells or rechargeable NiMH (or NiCad) cells if you prefer. The average current drain is typically around 20mA, so even ordinary alkaline cells should give at least 80 to 100 hours of listening.

Mounting the LEDs
As can be seen on Fig.7 and in the photos, LEDs 1 to 7 are all mounted with their leads bent down through 90°. This is done so that the LED bodies later protrude through their matching holes in the front panel.

In each case, it’s simply a matter of bending the leads down through 90° exactly 5mm from the LED’s body, then installing the LED with its leads 8mm above the PC board (see photo). Make sure that each LED is correctly orientated – the anode lead is the longer of the two.

The easiest way to get the LED lead spacings correct is to cut two cardboard templates – one 5mm wide and the other 8mm wide. The 5mm template is then used as a lead bending guide, while the 8mm template is used to correctly space the LEDs off the board.

The transmitter board assembly can now be completed by installing the two RCA phono connectors (CON1 and CON2) and the DC power socket (CON3).

Receiver board assembly
Fig.8 shows the assembly details for the receiver board. Once again, begin by soldering in the resistors and the small non-polarised capacitors, then install the larger electrolytics and the ICs. Note that the large 2200μF electrolytic capacitor is mounted on its side, with
its leads bent down through 90°.

Note also that the ICs are all different, so don’t mix them up. Take care to ensure they are correctly orientated.

Once the ICs are in, install the volume pot (VR1), the headphone socket and power switch S1. Follow these by installing PC pins at the A and K positions for PD1 (the BP104 photodiode) and at the power supply inputs.

The BP104 photodiode can now be installed by soldering its leads to its PC pins (see side-view diagram in Fig.8). Be sure to install this part the right way around. Its cathode (K) lead has a small tag, as shown on its pinout diagram in Fig.4.

It’s also vital to install this device with its sensitive front side facing out from the PC board.

Finally, LED1 can be mounted in position. This part must be mounted with 13mm lead lengths, so that it will later protrude through the lid of the case. A 13mm wide cardboard template makes a handy spacer when mounting this LED. Be sure to orientate it with its anode (A) lead (the longer of the two) towards IC2.

Final assembly – transmitter
The final assembly involves little more than installing the PC boards inside their respective cases.

Fig.10 shows the drilling details. The best approach is to photocopy these diagrams and then attach them to the panels so that they can be used as drilling templates. Note that hole ‘D’ is the adjustment access hole for trimpot VR1.

Once the panels have been drilled, they can be dressed by attaching the relevant artworks. These artworks are attached using double-sided adhesive tape. Once they are attached, they can be protected by covering them with clear self-adhesive film (eg, wide sticky tape) and the holes cut out with a sharp utility knife.

Final assembly – receiver
Now for the final assembly of the receiver. Use Fig.11 as a drilling template and attach the front panel artwork as described above.

As shown in the photos, the PC board is mounted on the underside of the lid on four M3 × 15mm tapped spacers. Four M3 × 6mm countersinkhead screws secure the spacers to the lid, while the PC board is secured using four M3 × 6mm pan-head screws.

The power LED (LED1) and toggle switch (S1) both protrude through matching holes in the lid. Once the PC board is in place, one of the switch nuts is fitted to the top of the threaded ferrule, to help hold everything securely together.

The two holes in the side of the box accept the shaft of the volume control (VR1) and the collar of the headphone socket (CON1). Another hole at one end of the box provides the ‘window’ for photodiode PD1.

As shown in the photos, a short length of PVC conduit was fitted around this hole, on the end of the box, to make a light shield ‘hood’. Although not strictly necessary, it does improve the signal-to-noise ratio of the link when you are using it in a fairly large room that’s lit with compact fl uorescent lamps (CFLs) – ie, when there’s a long link path. CFLs produce a signifi cant amount of noise at IR wavelengths and the hood stops most of this noise from reaching PD1.

For the prototype, the hood was made using a 15mm length of 16mm OD PVC conduit. This was glued to the box end (concentric with the hole) using fast-setting epoxy cement.

The battery holder, with its 4 × AA cells, is mounted at the other end of the box. This can be held in place using a strip of electrical insulation tape. It’s then wedged firmly in position by the end of the PC board when the lid goes on.

Note that the lid assembly must be introduced into the box at an angle, so VR1’s shaft and the headphone socket can enter their matching holes. It’s then swung down and fastened to the box using self-tapping screws.

Set-up and adjustment
Getting the transmitter unit going is straightforward. Basically, it’s just a matter of connecting the audio input leads and applying power. However, if you have an oscilloscope or a frequency counter, it’s a good idea to check the frequency of the clock oscillator before you close up the case.

This is easiest done by checking the frequency of the triangular wave signal at test point TP2 (just behind IC5). The frequency here should be between 80kHz and 100kHz. If it’s well outside this range, then you’ll need to change the value of the 680pF oscillator capacitor to
correct it.

The capacitor concerned is easy to find on the transmitter board – it’s just to the right of IC2.

In practice, a value of 680pF (as shown on the circuit) should be suitable if a Motorola MC14093B device is used for IC2. However, if an ST Micro 4093B is used, this capacitor will probably have to be reduced to 470pF or 390pF. Conversely, for a Philips 4093B, the capacitor may need to be increased to 820pF or even 1nF.

The basic idea is that you increase the capacitor’s value to lower the clock frequency, and reduce its value to increase the frequency.

If you don’t have a frequency counter, but have a modest uncalibrated oscilloscope, you can still check and adjust the clock frequency fairly easily by using the waveform at TP2 as a guide. The waveform here should be a very linear and symmetrical sawtooth, with a peak-to-peak amplitude of about 10.5V and only a very tiny ‘pip’

on each positive and negative peak. If you find that the waveform is a clean sawtooth, but much lower in amplitude than 10.5V p-p, this means that the clock oscillator’s frequency is too high. To fix this, simply increase the value of the 680pF capacitor.

On the other hand, if the waveform does have an amplitude of 10.5V p-p or more but is clipped or truncated rather than being a clean sawtooth, this means that your clock oscillator’s frequency is too low. That’s fixed by reducing the value of the 680pF capacitor.

If you don’t have a counter or an oscilloscope, leave the capacitor’s value at 680pF and wait to see if the link’s performance is satisfactory. We’ll discuss this option shortly.

The receiver unit needs no adjustments; all you have to do to get it going is to plug in your headphones, switch it on and point it towards the transmitter. The small green power LED should light and it’s then simply a matter of adjusting the volume control for a comfortable listening level.

Testing the link
To test the link, first connect the left and right channel audio signal leads to the transmitter’s inputs. These signals can come from the stereo line outputs on your TV. You can also use the line outputs on your VCR or DVD player, but only if you are actually using this equipment.

Note that piggyback RCA phono socket leads may be required to make these connections if the audio outputs are already in use (eg, Jaycar WA-7090).

Next, use a small screwdriver to adjust the ‘Set Level’ trimpot (VR1) at the rear of the transmitter to mid-position. That done, position the transmitter (eg, on top of the TV) so that it faces towards your viewing position and apply power. The transmitter’s green centre LED should immediately light (assuming an audio signal is being applied) but the IR LEDs will remain dark to your eyes.

It’s now just a matter of checking that the link actually works. To do this, initially set the receiver’s volume control to minimum, then plug the headphones in and switch the receiver on. The receiver’s green power LED should either blink briefly (if you’re not pointing the
receiver towards the transmitter) or light steadily if PD1 is able to ‘see’ the infrared signal.

The idea now is to place the receiver in a convenient position so that it gets an unobstructed ‘view’ of the transmitter. In most cases, it can simply be positioned on an armrest, an adjacent coffee table on even on the back of the sofa.

Now turn up the volume control and you should be able to clearly hear the TV sound. If so, your link is finished and ready for use.

If the sound is overly loud and distorted, even when the receiver’s volume control is down near zero, it’s probable that the audio input signals from the TV are overloading the transmitter. In that case, try adjusting trimpot VR1 anticlockwise using a small screwdriver, to lower the input level. This should allow you to remove any audible distortion and bring the volume down to a comfortable level.

If you find that distortion is still present, even when the audio level is turned well down, this probably means that your clock frequency is either too high or too low. This can occur if you weren’t able to previously check the transmitter’s oscillator frequency – eg, if you don’t have a counter or an oscilloscope.

In this case, try altering the 680pF capacitor’s value one way or the other, to see if the distortion gets better or worse. If it gets worse, go back the other way. If it gets better, keep changing the value in that direction.

In practice, you shouldn’t need to increase the capacitor value above 1nF or reduce it below 390pF in order to remove all audible distortion.

EPE

Downloads:Schematics

Ultrasonic Transmitter and Receiver

noreply@blogger.com (Anonymous) — Fri, 04 Jan 2013 14:59:00 +0000

THE Ultrasonic Transmitter circuit shown in Fig. generates ultrasonic sound waves at a frequency of 40kHz. The key component is a 555 timer, IC1, which is wired as an astable to produce an output frequency of 40kHz. This stream of electrical signals drives the ultrasonic
transmitter transducer TX1.

Potentiometer VR1, used as a variable resistor, enables you to make fine adjustment to the frequency when setting up the control system to ensure that TX1 is resonating at its optimum natural frequency, thereby ensuring maximum range. This adjustment is described in the ‘synchronising’ section at the end of the receiver section.

Ultrasonic Receiver:
The circuit shown in Fig. is designed to receive and process the 40kHz pulse of sound waves generated when the transmitter’s pushswitch is pressed. On the first push the relay is energised, and de-energised on the next push, enabling an electrical device to be switched on and off via the relay contacts. The Receiver circuit comprises the following building blocks:

Integrated circuits IC1 and IC2, which together amplify the ultrasonic sound waves received by sensor RX1 from the transmitter. Just like the Bat Detector project (Part 6), two coupled
amplifiers are used to boost the voltage gain and ensure remote control over a useful range of around 25 metres.

The amplified 40kHz pulses are fed to the base (B) of NPN transistor TR1 via capacitor C3. When a burst of ultrasonic waves is received, the voltage at the collector (C) terminal of TR1 suddenly falls as current flows through resistor R2.

The sharp fall in voltage is transferred to the trigger pin 2 of IC3, which is a 555 timer connected as a monostable circuit. Once triggered it produces a short pulse from its output
terminal (pin 3) of about one second duration. This pulse triggers flip-flop IC4 via its clock input pin 13, each successive pulse ‘toggling’ the output (pin 15) between logic high and logic low.

The final stage is the relay driver using NPN transistor TR2. When a logic high is produced at pin 15 of IC4, the transistor switches on, energising relay RLA1 and turning on the optional LED1. The next burst of 40kHz sound waves sets the output of IC4 to logic low, switching off the transistor and denergising the relay and LED1.

Do not omit capacitor C5, as it helps to stabilise the power supply voltage to the circuit.

Synchronising the system
The transmitter needs adjusting so that it generates ultrasound at a frequency of 40kHz. This is easy if you have an oscilloscope or frequency counter handy. If you do not have one,
proceed as follows.

Make sure the transmitter is connected to the 9V battery and its sensor is pointing towards the receiver placed about a metre away and also connected to its 9V battery. LED1 should be on or off depending on whether the output of IC4 is high (on) or low (off) when powering up the circuit. Now it’s a matter of trial and error!

Press and release pushswitch S1 on the transmitter while watching LED1 on the receiver. Make small adjustments to preset VR1 on the transmitter until repeatedly pressing and releasing S1 causes LED1 to go off and on. Gradually increase the distance between the
transmitter and receiver to 20 metres and continue to make fine adjustments to VR1. If the receiver is to be used to switch on and off an electrical device, a relay needs to be connected in parallel with LED1 and R5, as shown, and its switching contacts used to control the device in a separate circuit.

12V-24V High-Current Motor Speed Controller Part-3

noreply@blogger.com (Anonymous) — Fri, 21 Dec 2012 10:00:00 +0000

In first part we discussed about the detail of circuit, back EMF and Mosfet protection; and in second part we covered the display and menu, in this third and last part we will build and test our project.

The DC Motor Speed Controller is built on two PC boards: a main board, and a display board. These are joined together via a 12-way flat ribbon cable, which plugs into a pin header on the main board.

The main board can be assembled first – see Fig.9. Start by checking the PC board for hairline cracks and for any visible shorts across the copper tracks, especially between the ground plane and any adjacent tracks. In addition, check the hole sizes for the larger hardware items by test fitting these parts into position.

Making a link
That done, begin by installing the 17 wire links. These must go in first, since some of them run underneath some components.

To straighten the link wire, first clamp one end in a vice, then stretch it slightly by pulling on the other end with a pair of pliers. It’s then just a matter of cutting the links to length and bending their leads down through 90° to match the holes in the PC board.
Note that resistor R1 should be 1kohm if the supply voltage will be higher than or equal to 16V. Alternatively, use a 100ohm resistor if the supply voltage is going to be less than 16V. The 22ohm 1W resistor (on the lefthand side of Fig.9) should be mounted 3mm to 4mm proud of the PC board to aid heat dissipation.

Once the resistors are in, install the diodes and the Zener diodes. Take care to ensure that these are all correctly oriented and note that diode D2 (near inductor L1) must be a 1N5819
Schottky type.

Take care also with the Zener diodes. ZD1 to ZD5 are all 16V 1W types, while ZD6 and ZD7 are 33V 5W types. The fast recovery diode (D3) can be left until later, as there are two
options for this.

MOSFETs
The next thing to do is to solder in the four power MOSFETs (Q5 to Q8). These all come in a TO-220 package and sit horizontally on the PC board. Note that the source(s) lead of each device extends further than the other two. Before mounting each device, you will have to first position it on the PC board and bend its leads down through 90° so that they mate with the holes in the board. That done, fasten each device to the PC board using an M3 x 6mm screw and nut before soldering its leads.

Do not solder the leads before bolting the metal tab down. If you do, you risk stressing the soldered joints as the screw is tightened, and this could crack the copper or lift the pads.

The next thing to do is to solder in the two IC sockets. Make sure these are oriented correctly as per the component overlay. It is important that you use IC sockets because calibration of
the +5V rail is done with microcontroller IC1 out of circuit. This is described later during the setting-up stage.

Now the four bipolar transistors can be soldered in. There are three BC337 NPN types (Q2 to Q4) and one BC327 PNP type (Q1) – don’t get them mixed up. Their leads will only go in one way, and it’s just a matter of pushing them down onto the board as far as they will comfortably go before soldering the leads.

The single 3mm red LED can go in next, again taking care to orient it correctly. Its anode lead is the longer of the two; the cathode can also be identified by a ‘flat’ on the LED’s body.

Follow this with trimpot VR2 and the piezo buzzer. Note that the buzzer must be installed with its positive terminal towards the bottom edge of the board – see Fig.9.

Once these parts are in, you can solder the capacitors in place. The ceramic, monolithic and MKT polyester types can go in either way round, but be sure to orient the electrolytics
correctly. The large low-ESR 2200uF capacitor sits horizontally on the PC board, with its leads bent at 90° to go through the holes in the board.

The SPDT horizontal toggle switch (S1) can be soldered in next. Alternatively, this part can be mounted on a panel and connected back to the PC board via flying leads – it all depends
on how you intend mounting the board.

The 220uH toroid inductor (L1) sits vertically on the PC board – see photo. Be sure to strip the enamel from its leads and tin them with solder before attempting to solder it in position.

CON1, the 12-way header, is next on the list. This should be oriented as shown on Fig.9 and the photographs. That done, mount the two heavy-duty 3-way screw terminal blocks. Note that these are dovetailed together to make a 6-way block before mounting them on the board.

Potentiometer VR1 can now be installed. There are two possible locations on the board for this part. Mount it in the position that’s the most convenient for your application (but
don’t install a pot in both positions).

Alternatively, it can be mounted off the board and connected to its mounting pads via flying leads. Leave the two ICs out of their sockets for now. We’ll cover their installation later.

Fast recovery diode
Now for the fast recovery diode (D3). If you are using the 40EPF06PBF diode (rated at 40A), then this can be soldered straight in as shown in Fig.9 and the photo at left. Make sure that it goes in with its metal face towards the screw terminal block.

Alternatively, if you are using the two MBR20100CT 20A diodes, then these must be connected in parallel and installed as shown in Fig.10 and its accompanying photos. These particular diodes come in TO-220 packages and each package itself contains two 10A diodes, which are used in parallel.

The middle lead connects to the tab and is the cathode. This middle lead must be cut off using a small pair of sidecutters, as the tabs are used to make the cathode connections to the
board. The two outer leads of each device are the anode connections. The procedure for mounting these two diodes is as follows:
1) Drill the hole for the cathode connection on the PC board to 3mm.
2) Lightly tin the copper around the two mounting holes (this is necessary to ensure good contact with the screw heads).
3) Secure the metal tab of the first device to the PC board using an M3 × 12mm screw and nut, with the screw passing up from the underside of the board. Tighten the nut down firmly.
4) Install an M3 × 12mm screw through the hole at the anode end of the diodes, secure it with a nut, then fit a solder lug and secure it with a second nut. Again, make sure the nuts are done up tightly.
5) Solder the two outer leads of this first device to the solder lug, then fit another solder lug and nut.
6) Secure the tab of the second device in position and solder its outer leads to the top solder lug.

Building the display board
The display board is optional, but will be useful in many applications. If you decide not to build it, you will not be able to change the settings, and the default values will have to be used. You will also have no way of knowing what percentage of full speed the motor is running at.

As before, check the board for defects. In particular, check for shorts between copper tracks or between the ground plane and any adjacent tracks. Start the assembly by installing the
23 wire links. Some of these sit under the 7-segment LED displays, so make sure these links sit flat against the PC board. Don’t forget the short links immediately to the left of CON2 and at bottom right.

The resistors can go in next, followed by the 100nF capacitor and the four transistors (Q9 to Q12). If you are using an IC socket for IC3, then this can also now go in. Install it with its notched end positioned as indicated on the diagram, then install IC3 (74HC595).

Alternatively, IC3 can be directly soldered to the PC board. Be sure to install the IC with the correct orientation – ie, its notched end goes towards CON2.

Finally, complete the display board assembly by installing a 12-way pin header (this part is optional), the two pushbutton switches and the four 7-segment displays. Be sure to orient
the displays with the decimal points at bottom right.

Similarly, make sure that the push-button switches are correctly oriented. As shown in Fig.11, they must each be installed with the flat side towards the displays.

Connecting the boards
Once the board assemblies are complete, make up a 12-way ribbon cable to connect the two boards together. This should be terminated at either end to a 12-way header plug. Be sure to arrange this cable so that pin 1 of the header of the main board connects to pin 1 of the
header of the display board and so on.

Set-up
Once you have completed the construction, the next step is to go through the setting up procedure. You also need to adjust trimpot VR2 on the main board, so that the output from the MC30463 IC (IC2) sits at exactly +5V (this rail must be at exactly +5V to ensure that voltage measurements made by the microcontroller are accurate).

The set-up procedure is as follows:
1) Check that the two ICs on the main board are out of their sockets and that toggle switch S1 is in the OFF position (ie, the switch should be in the opposite position to that shown Fig.9).

2) Connect a 12V to 24V DC supply to the screw terminal blocks (check the supply voltage before you do this). Note that if you are connecting the supply leads directly to a battery,
there could be a spark when you first connect power due to the low-ESR 2200uF bypass capacitor across the supply.

For this reason, connect the supply leads to the battery first, then to the terminal blocks, as it’s never a good idea to generate a spark near a battery.

As noted previously, resistor R1 should be 1kohm if you are using a supply greater than or equal to 16V. Alternatively, R1 should be 100ohm if you are using a supply less than 16V.

3) Apply power to the circuit by toggling S1 to ON.

4) Check the voltage on D1’s cathode. It should be about 0.6V less than the supply voltage.

5) Check the voltage on ZD1’s cathode. This should be very close to +16V if you are using a power supply that’s greater than 16V. Alternatively, it should sit between +12V and +16V if
you are using a 12-16V power supply.

6) If these voltages are OK, switch off and install the MC34063 switchmode IC (IC2) into its socket. Make sure that the notch on the IC matches the notch on the socket – ie, the notch
must face towards inductor L1.

7) Apply power and check the voltage at pin 1 of the 12-way header on the main board. This is the +5V rail, but it may not yet be at exactly +5V (the exact voltage depends on the setting
of trimpot VR2).

8) Adjust VR2 until the voltage on pin 1 of the header is exactly +5V.

9) Assuming that the above voltage is now correctly set, switch off and insert microcontroller IC1 into its socket. Make sure that its notched end goes towards link LK11.

10) Plug the display board into the 12-way header and apply power. If everything is working correctly (and assuming a motor isn’t connected), the 4-digit display should immediately read ‘P00.0’, indicating the current speed.

11) If it all works correctly, skip the following troubleshooting section and go straight to the software initialisation procedure.

Troubleshooting
If you strike problems, the first thing to do is go back and check the board for any missed or bad solder joints. Check also that there are no shorts between tracks or between the ground plane and any adjacent tracks.

If these checks don’t reveal the problem, we suggest that you go through the following checklist:
1) Check that all 17 links are installed on the PC board. A missing link could mean that parts of the ground plane are floating and this will lead to improper operation!
2) Check that all polarised components, including the diodes, Zener diodes, electrolytic capacitors and ICs are correctly oriented as per the component overlay.
3) Check that you have used the correct value for R1, depending on your input supply voltage.
4) Make sure that the four transistors on the main board are the correct types. Q2 to Q4 are all BC337s but Q1 is a BC327 type.
5) Make sure that the input voltage connections to the 6-way terminal block are correct.

Software initialisation
If your boards are working, it is now time to initialise the software settings before connecting a motor. To do this, you must have the display board connected to the control board.

Note that more detailed user instructions for the DC Motor Speed Controller appeared in Part 1 and Part2. This section simply explains how to change the battery level alarm and the frequency of the PWM before you use the unit for the first time. These settings will be retained in non-volatile memory.

When the microcontroller begins executing the firmware (from a power-on reset) you should be taken straight to the main menu. This will indicate the percentage of full speed
that the motor is currently running at. Without a motor connected, it should read ‘P00.0’.

From here, press ‘short R’ twice (see Part 1) to arrive at the battery level alarm menu. This menu shows an ‘A’ followed by a 3-digit voltage value. which indicates the voltage level
below which the low battery alarm will sound.

In this menu, press ‘long L’ to set the level using potentiometer VR1. The ‘A’ should start flashing and you should then be able to vary the pot to change the level. Once you are happy
with the current level, press ‘short L’ to update the setting and return to the main menu.

Next, press ‘short R’ three times to arrive at the frequency menu. This will show an ‘F’ followed by a 3-digit frequency in kilohertz.

Once you are in this menu, press ‘long L’ and set the frequency of the PWM (pulse width modulation) using VR1. The ‘F’ should flash while you are setting the frequency. In practice, the frequency can be set to one of 256 values between 488Hz and 7812Hz.

Once you have set the desired frequency, press ‘short L’ to store the setting and return to the main menu. That’s it! For more detailed instructions on the other software modes, refer back to Part 1, and Part2.

Connecting the motor Once you have verified that the DC Motor Speed Controller is working
correctly, you can connect a motor. This should be connected with its positive terminal to the second terminal block from the top, while its negative terminal can go to either the
third or fourth terminal from the top.

Note that all supply and motor connections to the terminal block should be run using heavy-duty 56A wire.

The top terminal block is used to terminate the positive supply lead from the battery. This lead should connect via the 50A in-line fuse. Either of the bottom two terminals can be used for the negative battery lead (ie, one is left unused).

Your DC Motor Speed Controller is now ready for action.

EPE

Downloads:Code n PCB Files

12-24v High Current Motor Speed Controller Part 2

noreply@blogger.com (Anonymous) — Sat, 15 Dec 2012 06:17:00 +0000

In first part we discussed about the detail of circuit, back EMF and Mosfet protection; in the second part we will cover these:

Other protection measures
As already mentioned, diode D1 provides reverse polarity protection for microcontroller IC1 and the switchmode supply (IC2). Zener diode ZD1 is self-protecting in the case of reverse supply connection. However, if the supply is reversed, there will be a heavy conduction path via fast recovery diode D3 and the internal substrate diodes in the four power MOSFETs. If you are lucky, the 50A fuse will blow before the MOSFETs are damaged, but there is no guarantee of this. SO dOn’T rEVErSE THE bATTErY cOnnEcTIOnS!

In a similar vein, if the outputs are shorted while power is applied, high current will flow through the MOSFETs. Again, if you are lucky, the 50A fuse will blow before the MOSFETs go up in smoke. In reality, the 50A fuse is there to stop a fire! SO dOn’T SHOrT THE OUTPUTS TO THE MOTOr.

If the motor is under heavy load and becomes stalled, high currents will flow in its armature. Depending on the motor’s rating, this may or may not blow the fuse. If the fuse does not blow during stall conditions of the motor, the MOSFETs should survive, although they may get very hot.
Warning buzzer
If the circuit is overloaded, the battery voltage should drop to the point where the warning buzzer sounds.

LED1 and its 470ohm current-limiting resistor are switched by a high level on the output of pin 3 of the microcontroller. This is configured as a simple digital output. It also turns on transistor Q4 and the piezo beeper. This output is controlled by the firmware and can be disabled.

A 1kohm pull-up resistor is used on pin 4 (reset) of the PIC16F88-I/P. This ties the reset pin high, which means that the microcontroller is reset only at power-on.

Finally, the rest of the outputs of the microcontroller, namely pins 7 to 13 and 15 to 17, are used to drive the optional display board.

Display board
Fig.1 shows the optional display board circuit. It connects to the main board via 12-pin header CON1 and a ribbon cable.

The display board consists of two pushbutton switches, four 7-segment displays, which are multiplexed by the firm-ware, four transistors and some resistors, as well as a 74HC595 shift
register (IC3).

Pins 1 and 2 of the 12-way connector CON2 supply +5V to the display board. Pin 3 is connected to a digital input of the microcontroller (pin 17) and is pulled high by a 1kohm resistor on the main board. Conversely, it is pulled low by the display board.

This is used by the microcontroller to detect whether the display board is connected or not.
Pins 4 to 7 of CON1 are used to drive transistors Q9 to Q12 on the display board. These transistors switch the 7-segment display cathodes (K).

Pins 8 to 10 of CON1 are respectively the CLK, DATA and OUTPUT ENABLE lines and these go to the 74HC595 shift register (IC3). The microcontroller drives these lines to load a new 8-bit value into the shift register. The outputs of the shift register are connected across the four 7-segment displays and drive the anodes.

Finally, pins 11 and 12 are connected to pushbutton switches S2 and S3 on the display board. They are also connected to digital inputs on the microcontroller (which have internal pull-ups enabled) and these inputs are used to monitor the pushbuttons.

DC Motor Speed Controller: Software Features and Set-up:
ThE STRuCTuRE of the firmware for the DC Motor Speed Controller is shown overleaf in Fig.2. The transitions between the various menus are made using the switches on the display board and are indicated with labelled arrows.

There are four possible switch presses, either ‘Short’ or ‘Long’ and either the Left (L) or Right (R). Thus, for example, ‘Short R’ refers to a short press of the right pushbutton.

Main menu
The Main menu is as shown in Menu 1. It consists of the letter ‘P’ (for ‘percentage’) and three digits with a decimal point indicating the range 00.0% to 99.9%. The percentage value indicates the fraction of full speed the motor is currently running at.

In this mode, the motor’s speed can be adjusted by varying the pot. The letter ‘P’ will flash while the motor’s speed increases or decreases to the new setting. When the current speed reaches the speed set by the pot, the letter ‘P’ will stop flashing and there will be a short beep (if enabled).

Since there is a small periodic window when the pulse width modulation (PWM) is turned off by the firmware in order to read the back-EMF, at full speed the reading will not indicate 99.9% but will achieve its maximum at around 98% or so.

Monitoring the input voltage
From the Main menu, press ‘Short R’ once and you will be taken to the display shown in Menu 2. It consists of a ‘b’ (for battery) followed by three digits with a decimal point indicating a level from 00.0V to 99.9V, to monitor the battery. For good voltage accuracy, it is important that the +5V supply rail be precisely set using trimpot VR2.

In practice, with the supply rail to the microcontroller set at 5V, the level will not register any higher than around 40.1V. This is because the voltage divider used to derive the voltage reading consists of 33kohm and 4.7ohm resistors. The relatively high series resistance of 37.7kohm was chosen to avoid damaging the input of IC1 if the input voltage goes any higher than around 40V.

To go back to the Main menu, either press ‘Short L’ or press ‘Long R’. If you press ‘Long L’, you will set the low-battery alarm level to 91.6% of the current voltage input level (and then return to the Main menu). This is a shorthand way to set the low-battery alarm level when you know that the batteries are fully charged.

For a typical 12V battery, they are fully charged at around 13.8V (with charger connected) and should not be discharged beyond 11V.

Press ‘Short R’ to go to the low battery alarm level menu.

Setting the low-battery alarm
From the Main menu, press ‘Short R’ twice. You will be taken to the low-battery alarm level menu as shown in Menu 3. It consists of an ‘A’ (for alarm) followed by three digits, which indicate a level between 00.0V and 41.6V. This will show the current setting of the low-battery alarm or rather, the voltage level below which the alarm will sound (if enabled).

Whenever the input voltage is below this level, the display will flash (with increasing frequency as the voltage drops) while if the alarm sound is enabled, there will be a flash from LED1 and a beep.

To set the low-battery alarm level press ‘Long L’. The ‘A’ will start flashing and then the low-battery alarm level can be modified by adjusting the pot setting. To turn the alarm off completely, simply set the level to 00.0V.

When you have reached the required level, simply press any button and the level will be recorded (and stored in EEPROM). Then there will be a beep (if enabled) and you will be taken to the Main menu.

Note that the motor will be turned off automatically when setting the low-battery alarm level.

Setting the PWM frequency
From the Main menu, press ‘Short R’ three times. You will be taken to the frequency menu, as shown in Menu 4.

This consists of an ‘F’ (for frequency) followed by three digits with a decimal point indicating a level between 0.48khz and 7.81khz. This is the current PWM frequency. As the frequency increases, the resolution of the PWM setting decreases.

At 0.48khz (actually 488hz) the resolution is 10 bits. This decreases to six bits at 7812hz. Thus, the resolution is at worst six bits or 64 levels, and at best 10 bits or 1024 levels.

While in this menu, press ‘Long L’ and you will be able to set the frequency. The ‘F’ will start flashing and then the frequency will be modified according to the pot setting.

When you have reached the required frequency, simply press any button and the level will be recorded and stored in EEPROM. Then there will be a beep (if enabled) and you will be taken to the Main menu.

Note that the motor will be automatically turned off when setting the frequency.

Enabling and disabling audible cues
From the Main menu, press ‘Long L’. You will be taken to the settings menu as shown in Menu 8. It consists of ‘A’ (for alarm) followed by either ‘0’ or ‘1’ (0 = disable, 1 = enable) and a ‘b’ (for beep) followed again by either ‘0’ or ‘1’ (0 = disable, 1 = enable). In this menu, pressing ‘Short L’ will toggle the alarm setting (enable/ disable) and pressing ‘Short R’ will toggle the beep setting (enable/ disable).

When the alarm setting is disabled, there will be no beeping when the input voltage falls below the alarm level. There will still be a warning flashing on the display, however. To disable the latter, simply set the alarm level to 00.0V. When the beep setting is disabled, audible beeps emitted by the firmware at certain points (as when setting certain values or when the desired speed is reached) will be blocked.

If you do not want any beeping from the piezo buzzer, simply set ‘A’ to 0 and ‘b’ to 0. In this menu, pressing ‘Long L’ will take you to the Reset Menu, as explained below. Pressing ‘Long R’ will take you back to the Main menu.

Reset menu
From the Main menu, press ‘Long L’ twice. You will be taken to the Reset Menu, as shown in
Menu 9.

It consists of the letters ‘CL’ (for clear) followed by two digits and a decimal point of the form X.X. The X.X represents the current version of the firmware, which for this release stands at 3.0. It is possible that future releases of the firmware will add new features or
refinements to critical sections of the code.

While in this menu, press ‘Short L’, ‘Short R’ or ‘Long R’ to go back to the Main menu.

Note, however, that pressing ‘Long L’ will reset all settings to the default values and the speed controller will lock until power is turned off. When a power-on reset next occurs, the default values for the frequency, low-battery level alarm and audible beeps will be restored.

This feature is useful for initialising the firmware variables and for making sure that you begin from a known state. Most of the time, it will not be used.

Memory speed mode
From the Main menu, press ‘Short L’ to enter memory mode. The display will be as shown in
Menu 6.

It consists of the letter ‘C’ (for constant) followed by a digit from 1 to 8 (indicating one of the eight available memories), in turn followed by two dashes.

Now adjusting the pot will select one of the eight memories. When the pot becomes stable for a short period, the speed of the motor will be set according to the current value of that memory.

The display will change as shown in Menu 7. This display still consists of the letter ‘C’ followed by the number of the memory, but it will then have a decimal point followed by two digits representing the speed percentage from 00% to 99% (the first two letters will flash
until the set speed is reached).

Adjusting the pot will now change the selected memory and the speed setting will be recalled from one of the eight stored memory speed settings (after a short beep, if enabled).

To go back to normal mode, where the motor speed is controlled directly by the pot, simply press any key, long or short.

Setting the memory
To set one of the eight memory speed values you press ‘Long R’ from the Main menu. The display will change as shown in Menu 5. It consists of the letter ‘C’ (for constant) followed by a digit from 1 to 8 (indicating one of eight memory settings) and two dashes.

Now adjusting the pot will select one of the eight memory settings to store the current value of the speed of the motor.

When the pot becomes stable for a short period, the speed of the motor will be stored at that
particular memory. This can be recalled later by entering memory mode, as explained in the previous section.

There will be a short beep (if enabled), indicating that the value has been stored and you will be taken back to the Main menu.

EPE

DIY Desktop-Table LED Lamp

noreply@blogger.com (Anonymous) — Sat, 08 Dec 2012 05:39:00 +0000

Hi guys lets do something woody!!! something you may wanna do!. If don't then,.... what can i say its your choice.Ok lets get started.
In this DIY i ll tell you the story of my diy wooden desktop and wall led lamp. So first of all the things required are: 1)round slick/rod of one inch dia.
2) Some nuts n bolts
3) some washer
4) 24 High bright LEDs
5) 8 100ohm resistors
6) 12v adaptor
7) small piece if MDF board
8) on/off switch
9) piece of flat thin wood (as you can see in fig bellow, actually i don't know the name of this wood!!! )

Thin Wood stick n round wood piece

Steps for DIY LED lamp:

Fig 2

Cut the round stick about one foot, you can increase or decrease the height its all depends on you.
Now mark the one inch, cut the round edges and leave the center; can't understand?? i knew it, see the picture above.
After this cut the two pieces of one foot flat wood stick and round the edges with sandpaper (see in above fig).
Now drill them all but remember the size of drill bit is slightly more than the dia of the bolt you are using; in my case its about 8mm.
Sandwich the round rod/stick with the pieces of flat wood sticks with the help of nut, blot n washer. The arrangement is as follow:
First insert washer in bolt then these into wood stick then washer then round stick then again washer then the other piece of wood stick
then washer again then the nut. That is all, as you can see in the picture 2.
Tight this assembly so it can easily be adjusted.
Now connect the LEDS as shown in the circuit. I am using 24 LEDs eight parallel combination of three series leds.
For resistor value: each led forward voltage is 3v and can handle 30ma easily, so three leds makes 3vx3=9v and supply is 12v.
so it gives r=(12v-9v)30ma=100 ohm. If you are using different supply voltage then calculate the value of resistor using the above equation.
And for wattage use w=i*i*r=.03*.03*100=.09 so quater watt is ok, use0.25watt resistors.
Also note the supply voltage must be greater than 9v.
Now assemble the circuit and cut the MDF board of same size. Then connect the LEDs circuit with the MDF board with the help of nut n bolts.
After this cut the pieces of flat wood to cover the leds and plase and wire the on/off switch.
Cut the two inch round slick and same as in step two(see picture). And attach it with the MDF board assembly as told in step 5 n 6,see fig 3 n 4.
Now fix the whole assembly on the heavy base, i have iron foot as base!!! see picture 5.
Connect the circuit with adaptor n hurrayyyyyy where is your complete table/desktop lamp.

fig 3

fig4

fig 5

For the wall lamp every thing is same except the base. For that cut the MDF board about 3"x6" and the round stick as in third last step.

The other difference is; there are two points of ajdustment for table/desktop lamp, one is the flat wood and round stick junction and the other is the flat wood and MDF board. Where as the fixed wall lamp have three points of ajdustment, the third one is on the MDF board base.

So its span is 180 degree on x-axis more than 180 degree in y-axis and about 180 degree for LEDs on y-asix. While the first one is not included
in the desktop lamp, but you can add it the same way, if you want.Picture of wall lamp in action!!!!

Rest of the Pictures:

nut bolt washer sample

LED assembly

switch and some tape!!!

wall lamp adjustable points

Lamp in action.

12-24v High Current Motor Speed Controller Part 1

noreply@blogger.com (Anonymous) — Sat, 24 Nov 2012 06:30:00 +0000

This 12V or 24V high-current DC Motor Speed Controller is rated at up to 40A (continuous) and is suitable for heavy-duty motor applications. All control tasks are monitored by a microcontroller and as a result, the list of features is extensive.

This high-current motor speed controller is based on a PiC16F88 microcontroller. This micro provides all the fancy features, such as battery monitoring, soft-start and speed regulation. it also monitors the speed setting potentiometer and drives a 4-digit display board, which includes two pushbuttons.

The 4-digit display board is optional, but we strongly recommend that you build it, even if you only use it for the initial set-up. it unlocks the full features of the speed controller and allows all settings to be adjusted.

The microcontroller will detect whether the display board is connected, and if not, the speed controller will support only the basic functions. in this simple mode, it will function as a simple speed-regulated controller with automatic soft-start and with the speed being directly controlled by a rotary potentiometer (VR1). All the other settings will be the initial defaults or as last set (with the display board connected).

When connected, the 4-digit display allows you to monitor the speed and the input voltage (useful when running from a battery). it also enables you to navigate through the various menus to adjust the settings.

The circuit can run from 12V or 24V batteries and can drive motors (or resistive loads) up to 40A. Furthermore, this is our first DC speed controller (except for train controllers) incorporating speed regulation under load. in other words, a given motor speed is maintained, regardless of whether the motor is driving a heavy load or not.

Monitoring the back-EMF
in speed controllers which do not have good speed regulation (ie, the vast majority of designs), the more a motor is loaded, the more it slows down. In order to provide speed regulation, the circuit must monitor the back-EMF of the motor, since this parameter is directly proportional to its speed.

As a result, this speed controller monitors the back-EMF of the motor. ‘Back-EMF’ is the voltage generated by any motor to oppose the current through its windings. EMF stands for ‘electromotive force’ and is directly proportional to the motor speed and so by monitoring this parameter, we have a means of controlling and maintaining the motor speed.

In practice, the main control loop of the microcontroller tries to match the speed of the motor (back-EMF) to the speed set by the pot or a value recalled from a preset memory. If the measured speed is lower than the set speed, the duty cycle of the pulse width modulation (PWM) signal used to drive the power MOSFETs that control the motor is gradually increased. In other words, if the speed tends to drop, more power is fed to the motor
and vice versa.

The frequency of the pulse width modulation can be set from 488Hz to 7812Hz. This is a useful feature, since different motors will have different frequency responses, as well as different resonant frequencies. It is important to reduce the audible buzzing from the pulse width modulation, as these frequencies are well within the range of hearing.

Window of opportunity
By now, you’re probably wondering how the microcontroller monitors the back-EMF of the motor, considering that the motor is continuously driven with pulse-width modulated DC. The answer is that the micro periodically turns off the PWM signal to the motor for just enough time for the back-EMF to stabilise.

This ‘window’ needs to be wide enough to ensure that we are measuring back-EMF and not the spike generated by the last PWM pulse. On the other hand, we don’t want the window so wide that the maximum power to the motor is significantly reduced or that the motor noticeably slows.

The compromise value is that the motor is monitored for 200m s every 7.4ms (ie, about 135 times a second), as shown in the scope screen shots in Fig.3 to Fig.7. As a result, the fact that we do monitor the back-EMF around 135 times a second means that the power applied to the motor is slightly less than the theoretical maximum, although this effect is negligible.

A low-battery alarm is also incorporated to warn when the battery level drops below a preset value. This is especially useful for applications like electric wheelchairs.

There are also eight memory speed settings. All settings are persistent, meaning they are retained in nonvolatile memory.

Soft-start
When the motor is switched off, perhaps by an external switch in series with one of its terminals, the voltage at the drain (D) of the MOSFETs will be 0V (this is due to the voltage divider used to scale the back-EMF voltage to within the operating range of the microcontroller). The microcontroller converts this analogue value to a digital value using an on-board ADC (analogue-to-digital converter).

The firmware detects this 0V condition and sets the duty cycle of the PWM to 0%. This ensures that when the motor is switched in, its speed will increase gradually from the stationary state to the desired speed setting.

Turn-on currents for motors can be very high and it is desirable to reduce these surge currents as much as possible. That is why the automatic softstart feature has been incorporated into the firmware. It will ensure that the motor is brought up to the set speed gradually.

Fast switch-off feature
Another feature that has been incorporated into the firmware is the so-called ‘fast-off’ feature. This means that the duty cycle of the PW modulation is set to 0% (turning off the motor) whenever the selected speed setting of the pot goes to 0%. Rather than decreasing the speed gradually, setting the pot to its lowest setting turns the motor off immediately.

This design also incorporates our extensive experience with previous speed controllers. As a result, it uses four high-current MOSFETs to do the switching (pulse width modulation), uses very wide tracks on the PC board and heavy-duty (40A) terminal blocks to carry the large currents.

User interface
Two pushbuttons on the display board are used to navigate through the menus, while the potentiometer (VR1) is used both to vary the speed and to vary certain settings.

The two pushbuttons are sensitive to two types of presses, short and long. A short press is of the order of half a second or less, while a long press is around one second.

To change a setting, a long press is usually needed. This prevents unwanted changes to the settings, which are stored in EEPROM and thus recalled at the next switch on. Because of the capabilities offered by the PIC microcontroller, we have been able to incorporate a large
number of features into the firmware, as described in the separate panel later.

Circuit description
The circuit for the speed controller is shown in Fig.1. As noted previously, it can work with 12V or 24V batteries, but has been optimised for operation at 24V. Within the circuit itself, there are two separate voltage rails: +5V for the microcontroller and +16V for driving the gates (G) of the MOSFETs. Both are derived from the +24V input supply.

The main input supply is filtered by a 2200uF low ESR capacitor, to minimise highvoltage transients which can be produced by the inductance of the battery connecting leads. This capacitor is absolutely vital to the proper operation of the speed controller at high currents.

Power switch S1 and diode D1 protect the lowpower part of the circuit (IC1 and IC2) from reverse polarity. A 22W 1W resistor, a 33V 5W Zener diode (ZD7) and a 100n F capacitor also protect the MC34063 DCDC converterIC from transients on the supply rail.

The filtered supply is fed to the MC34063 (IC2), which operates in a standard step-down converter configuration to provide the +5V rail. Three 1W resistors between pins 6 and 7 are used to set the maximum switching current. The output voltage is set by the voltage divider associated with trimpot VR2. Only about 200mA is ever drawn from this supply, and most of this is used to drive the display.

IC1 is the heart of the circuit and is the popular PIC16F88 microcontroller, which incorporates a number of peripheral functions. Of these, the timers, hardware PWM (pulse width modulation) and three ADC inputs are used.

The three ADC inputs used are at pins 1, 2 and 18. As these need to be within the 0V to 5V range, voltage dividers consisting of 33kW and 4.7kW resistors are used to scale both the input voltage rail (which could be as high as 29V) and the backEMF from the motor, to be fed to the ADC inputs at pins 1 and 18. The ADCs convert the monitored voltages to 10bit values.

Gate driver
The +16V rail is used as the gate drive supply for the MOSFETs and is derived from the 24V supply via a 1kohm resistor and a 16V 1W Zener diode (ZD1). Bypassing of this rail is particularly important and is accomplished using 100uF and 100nF capacitors near ZD1 and adjacent to the transistors Q1 and Q2.

If the battery supply is to be 12V, the 1k ohm resistor feeding ZD1 should be reduced to 100ohm. In this case, the supply will actually be between 12V and 14V (depending on the actual battery voltage); still enough to provide adequate gate drive for the MOSFETs and ensure minimum heat dissipation (low onresistance).

The PWM output of the PIC16F88 (adjusted by firmware) appears at pin 6 and drives transistor Q3, which then drives complementary transistors Q1 and Q2. Transistor Q1, Q2 and Q3 thus provide buffering and voltage level translation for IC1’s PWM output to drive the gates of MOSFETs Q5 to Q8, via 15ohm resistors.

Note that these resistors need to be relatively low in value (ie, 15ohm) in order to ensure quick charging and discharging of the gate capacitances. That’s because the gate capacitance of these MOSFETs can be quite high, of the order of 5000pF to 10,000pF each. If the gate charging time is too long, the MOSFETs will spend too much time between the on and off states and this will lead to much higher heat dissipation.

In fact, the gate voltage transitions need to be very fast, of the order of 1m s or less. This has been accomplished, as shown by the oscilloscope screen grab of Fig.4.

The specified MOSFETs are from International Rectifier, type IRF1405. This is a 55V 169A N-channel HexFET with an exceptionally low on-resistance (Rds) of 5.3 milliohms (5.3mohm) typical. Their pulse current rating is a stupendous 680A.

The IRF1405 is specifically intended for automotive use, in applications such as electric power steering, anti-lock braking systems (ABS), power windows and so on, and is therefore ideal for this speed control application.

Why four MOSFETs?
In fact, since the ratings of this MOSFET is so high, you might think that just one device on its own would be enough to handle the 40A rating of this speed controller project. So why are we using four MOSFETs in parallel?

As always, real world use brings us down to earth. For a start, we are using these MOSFETs without heatsinks, apart from the heatsink effect of bolting them to the copper side of the PC board – not much heatsink benefit there. Their thermal characteristic is 62°C per watt (junction to ambient), if they are mounted in free air (which they are not).

Assuming an ambient temperature of 25°C and an on-resistance of 10mohm (conservative), we can approximate the temperature of the MOSFETs at their highest operating current (10A per MOSFET for a total of 40A). At 10A and 10m ohm on-resistance, the power dissipated is: 10^2 × .01 = 1W Therefore, the temperature of the case will be approximately: 25 + 62 × 1 = 87°C

This means that at full current, the MOSFETs will be very hot to the touch. Careful: they will burn you. Our measurements produced a top temperature of around 77°C after a test period of half an hour.

In practice, even with much higher ambient temperatures, the MOSFETs should not get quite this hot because in ‘real world’ operation, the speed control is not likely to be providing full power to the motor on a continuous basis. At 24V and 40A, the motor would have 960W applied (ie, more than 1HP) and this equates to relatively high power operation.

Protection
Zener diodes ZD2 to ZD5 are included to protect the MOSFETs from excessive gate voltages. In normal circuit operation, these Zener diodes do nothing.

Additional protection for the drains of the paralleled MOSFETs is provided by 33V 5W Zener diode ZD6, in parallel with a 100nF capacitor. The Zener is there to clip any residual voltage transients which get past the 2200uF low-ESR input filter capacitor.

The MOSFETs are further protected by fast-recovery diode D3 and its parallel 220nF capacitor. These parts are wired across the motor terminals and are used to suppress the high back- EMF spikes caused by the armature inductance when the motor is switched off by the MOSFETs.

These components are crucial to the operation of the speed controller. Without them, the high voltages generated can and probably would destroy the MOSFETs.

EPE

Note: In above text where you find 'W' with resistor values, its actually Ohm not watt; mistake, sorry for that.

AVR ISP ScoketBoard

noreply@blogger.com (Anonymous) — Thu, 01 Nov 2012 03:30:00 +0000

Most Atmel AVR microcontrollers can be programmed via their in-built serial programming interfaces (SPI). This method is ideal for in-situ programming, such as might be used in manufacturing or for firmware development or field upgrades.

In this scenario, the micro remains in its socket on the application board and a low-cost in-system programmer (ISP) is plugged into a dedicated programming header. In other words, the microcontroller does not have to be removed from its socket and plugged into a parallel programmer each time a firmware update is required.

However, in some cases it is desirable to programme a microcontroller stand-alone, such as when the application board is unavailable or doesn’t include an ISP (or JTAG) header. A low-cost method of stand-alone programming might also be useful where a batch of chips is needed for a small prototype run and the cost of a commercial parallel programmer is prohibitive.

This is where the AVR ISP SocketBoard comes in. It provides the minimum of functions necessary to support in-system programming, including a regulated power supply,clock source and microcontroller IC socket. Just connect your in-system programmer to a PC, plug its ISP cable into the SocketBoard’s on-board header and add a DC plugpack. You’re then ready to start programming!

Programming sockets

As you can see from the photos, the SocketBoard contains five programming sockets. Why so many? Well, we’ve provided one programming socket for each group of micros with common SPI pinouts. This allowed us to eliminate the switching logic that would have been required if we’d used just a single, 40-pin socket, so greatly simplifying design and construction.

We expect that many constructors will install just one or two programming sockets (depending on their requirements), to keep costs as low as possible. The overlay diagram (Fig.2) lists specific device types and the sockets (SK1 to SK5) that support them. For example, to program the ATM ega16, socket SK4 must be installed.

For cases where quantities of chips need to be programmed, the board will accept standard zero insertion force (ZIF) sockets as well. There is absolutely no need to install ZIF sockets (as shown in our photos) for occasional programming; this would simply be expensive overkill.

The unit can be powered from a 12V DC 150mA (or higher) unregulated plugpack, which also powers the ISP programmer when it’s plugged into the on-board header.

Operation
As mentioned, the SocketBoard provides the minimum of functions necessary to support in-system programming. As stated, this includes a series of programming sockets to accommodate the different types of AVR micros, a regulated power supply, and a clock source.

The power supply is based around two series-connected LM317 adjustable positive regulators (see Fig.1). The first regulator acts as a current limiter. In normal operation, it performs no function. However, should the current through R1 increase to a level where about 1.25V is dropped across it, REG1 begins to reduce the voltage at its OUT terminal. In effect, REG1 then acts as a constant current source, limiting output current to a
maximum of 125mA.

In normal operation, the complete setup consumes an average of about 20 to 40mA, depending on the type of in-system programmer connected. The remaining capacity (85 to 105mA) leaves a comfortable margin, which in most cases is still low enough to preserve any micro that might be accidentally reversed in a socket. It also protects other components if an internally short-circuited micro is plugged into a socket.

The second regulator (REG2) is configured as a conventional voltage regulator. Without JP1 installed, it produces +5V to power the system. Installing JP1 reduces this to +3V. Some constructors may find this lower voltage useful for verifying the memory in micros that are destined for 3V systems.

As well as power, AVR micros require a clock source for their internal programming circuits to operate. This is provided by a Pierce oscillator, which is composed of a 4MHz crystal (X1), two resistors and one gate of a 74HC04 hex inverter (IC1a). A second gate (IC1b) buffers the clock signal before it is applied to all of the programming sockets. A 47W resistor provides series termination and current limiting.

All that now remains to be described is the ISP interface. This is extremely simple indeed, as it consists only of a 10-pin DIL header (CON2) and five resistors. The four 100W series resistors act as peak current limiters, in case the ISP cable or a chip is accidentally inserted with power applied. These also help to protect the programmer if a faulty micro is inserted in a socket. The remaining resistor (47kW) pulls down the interface’s RESET line, so that the micro is held in the reset state if a programmer is not connected or is non-functional.

Testing
Connect a 12V DC source to the DC socket (CON1), noting that the centre pin is the positive input. If the power connections are accidentally reversed, nothing bad will happen as a series diode (D1) provides polarity protection.

Now apply power by sliding switch S1’s actuator towards the edge of the board. The power LED should light immediately. If it doesn’t, either the power connections are reversed or there is an assembly error. Carefully recheck the board against the overlay diagram and look for dry or missed solder joints.

Next, use your multimeter to measure the voltage between pins 7 and 14 of IC1’s socket. Expect a reading of 5V ±5%. Temporarily insert a jumper shunt on JP1 and measure the voltage again. This time, you should get the lower reading of 3V ±5%. When done, remove
the jumper, as in the majority of applications, a 5V supply is preferred for programming.

If the power supply checks out, switch off and insert IC1 into its socket. Naturally, the position of the notched (pin 1) end of this IC must match that
of the IC socket.

Using it
It doesn’t take a lot of grey matter to use the SocketBoard. Simply switch power off, plug your in-system programmer into the AVR ISP connector (CON2), and insert the microcontroller to be programmed into the designated socket. After switching on, the micro can be programmed following the instructions supplied with your ISP.

Important: always switch the power off before inserting or removing a microcontroller from its programming socket.

Note that 8-pin micros present a special case. Instead of a separate socket, all 8-pin devices are programmed in the first 20-pin socket (SK1). In addition, jumper shunts must be installed on JP2 and JP3 to route signals to the correct places for these diminutive
devices.

After programming an 8-pin device, the two jumper shunts (JP2 and JP3) should be removed if you also intend to program 20-pin devices in the same socket. This ensures that there is no possibility of damage to the larger devices.

If a faulty micro is inserted in a socket or if a working device is inserted backwards, the current-limit function will swing into action. In most cases, the current passed through the part should not be destructive – if the problem is noticed right away and power is switched off immediately!

EPE

Downloads: PCB layout

Phase Angle Control Of SCR Using AT89C51

noreply@blogger.com (Anonymous) — Sat, 15 Sep 2012 06:40:00 +0000

Silicon-controlled rectifiers (SCR) are solidstate semiconductor devices that are usually used in power switching circuits. SCR controls the output signal by switching it ‘on’ or ‘off,’ thereby controlling the power to the load in context. The two primary modes of SCR control are phase-angle fired—where a partial waveform is passed every half cycle—and zerocrossing fired—where a portion of the complete waveforms is passed to regulate the power.

In the phase-angle controller, the firing pulse is delayed to turn on the SCR in the middle of every half cycle. This means that every time a part of an AC cycle is cut, the power to the load also gets cut. To deliver more or less power to the load, the phase angle is increased or decreased, thereby controlling the throughput power.

There are several ways to control the firing angle of SCR. This article describes a microcontroller AT89C51-based phase-angle controller. A microcontroller can be programmed to fire SCR over the full range of half cycles—from 0 to 180°—to get a good linear relationship between the phase angle and the delivered output power.

Some of the features of this microcontroller-based phase-angle controller for SCR are:

1. Utilises the zero-crossing detector circuit

2. Controls the phase angle from 0–162°

3. Displays the phase angle on an LCD panel

4. LED indicators are used for displaying the status of SCR

5. Increases or decreases the phase angle with intervals of 18°

Basically, the zero-crossing detector circuit interrupts the microcontroller after every 10 ms. This interrupt commands the microcontroller to generate some delay (in the range of 1ms to 9 ms). The user can increase or decrease the delay in intervals of 1 ms using switches. the SCR is then fired through the opto-coupler. This repeats after every 10 ms.

Circuit description:

The complete circuit is divided into two sections:
1. The zero-cross detector section
2. The control section

The zero-cross detector section.

Fig.1 shows the circuit diagram of the zero-crossing detector and the power supply. The main sections of the circuit are a rectifier, regulated power supply and zero-crossing detector. The 230V AC mains is stepped down by transformer X1 to deliver the secondary output of 9V, 500 mA. The transformer output is rectified by a full-wave bridge rectifier comprising diodes D1 through D4 and then regulated by IC 7805 (IC3). Capacitors C2 and C3 are used for bypassing the ripples present in the regulated 5V power supply. A capacitor above 10μF is connected across the output of the regulator IC, while diode D6 protects the regulator IC in case their input is short to ground. LED5 acts as the power-on indicator and resistor R5 limits the current through LED5.

This regulated 5V is also used as biasing voltage for both transistors (T1 and T2) and the control section. A pulsating DC voltage is applied to the base of transistor T1 through diode D5 and resistors R1 and R2. When the pulsating voltage goes to zero, the collector of transistor T1 goes high. This is used for detecting the pulse when the voltage is zero. Finally, the detected pulse from ‘C’ is fed to the microcontroller of the control section.

The control section:
Fig.2 shows the circuit diagram of the control section for the phase-angle control of SCR. It comprises a microcontroller AT89C51, opto-coupler MCT2E, LCD module and a few discrete components. Port 0 (P0.0 through P0.7) of AT89C51 is used for interfacing data input pins D0 through D7 of the LCD module.Port pins P2.6, P2.5 and P2.7 of the microcontroller control the registers select (RS), read/write and enable (E) input pin of the LCD module, respectively. Preset VR1 is used for controlling the contrast of the LCD module. Push-to-on switches S1, S2 and S3 are connected with the pins P1.0, P1.1 and P1.2 through diodes D9, D10 and D11, respectively. External interrupt pin (P3.2) of the microcontroller is connected to S1, S2 and S3 through D12, D13 and D14, respectively. The role of different switches is shown in Table I.

The output of the zero-crossing detector from ‘C’ is fed to the external interrupt pin (P3.3) of the microcontroller.

Port pin P2.0 is connected with pin 2 of the opto-coupler (MCT2E). The output pin 5 of MCT2E is used for triggering the gate of SCR TYN604. The anode of SCR is connected to the load (bulb) with the 230V AC supply.

A 12MHz crystal along with capacitors C5 and C4 are connected to the microcontroller pins 18 and 19 to provide the basic clock to the microcontroller. Power on reset is derived by using capacitor C6 and resistor R6. Switch S4 is used for a manual reset.

The operation

The complete operation can be well understood with the help of waveforms in Fig.3.

1. The waveform at point ‘A’ is a fully rectified wave that is fed to the base of T1.

2. When the base voltage falls below 0.7V, transistor T1 is switched off, pulling the output higher. This
results in a very short positive pulse, which is available at the collector, (at point ‘B’) as shown in the second waveform.

3. As this positive pulse is inverted by transistor T2, it produces one negative pulse of the same width
at ‘C.’ This is shown as the third waveform.

4. This negative pulse is fed to the interrupt pin of the microcontroller, which acts as an interrupt for the microcontroller. The microcontroller then generates a positive pulse on P2.0 (at point ‘D’) after some delay. This turns ‘off’ the internal LED of the opto-coupler (MCT2E) and a positive pulse is produced at output ‘E’. This is used for triggering (fire) SCR1.

5. Depending on the time delay in between the interrupt and the pulse on port pin P2.0 of the microcontroller, the SCR is fired in the middle of the half wave cycle.

6. Two different waveforms—one for 4 ms delay and the other for 8 ms delay—are shown in Fig.3. In the case of 4ms delay, the output positive cycle of the AC wave is 60 per cent of the input. Therefore, nearly 60 per cent of the power is delivered to the load (the dotted line shows part of waveform that has been cut). In the second case of 8 ms delay, the output cycle is 20 per cent of the input cycle, so only 20 per cent of the power is delivered to the load.

This change in delay is done using switches S1 and S2. Different LEDs are used for indicating different functions as shown in Table II.

The diodes D12 through D14 are connected in such a manner that whenever any of the three push-to-on switches are pressed, it generates an external interrupt .

When switch S1 is pressed for the first time, it enables external interrupt and displays the message ‘SCR on.’ So after every 10 ms, external interrupt is generated which starts the entire operation. Pressing switch S1 again disables external interrupt 0 and the message ‘SCR off’ is displayed. the complete SCR operation gets shut off.

On pressing S2, the delay increases by 1 ms (firing angle will shift by 18°) and firing of SCR is delayed by 1 ms. The power delivered to the load is also decreased by 10 per cent. The maximum delay that can be applied is 9 ms which will delay firing by an angle of 162°. When the limit is reached, it is indicated by LED3 and a message ‘Max. phase angle’ is displayed on the LCD. The glowing of the bulb goes off.

Similarly, when S3 is pressed, the delay is decreased by 1 ms and the load current increases by 10 per cent. The minimum delay is 0 ms, which means a full positive cycle is applied. However, when the limit is reached, it is indicated by LED4 and a message ‘Min. phase angle’ is displayed.

Software program:

The software code for this project is written in ‘C’ programming language and compiled using the Keil μVision3 compiler. After compilation, the final.hex code is downloaded to the microcontroller using a suitable programmer. The source program is well commented and easy to understand.

The main function initialises the timer, ports and LCD. Finally, after enabling the external interrupt 0, it enters into a continuous loop.

Int0 function is an interrupt function and is automatically called when any of the three switches S1 through S3 is pressed.

1. If switch S1 is pressed, it checks if it is pressed for an even/odd number of times. Accordingly, it either switches ‘on’ or switches ‘off’ the SCR. Basically, it enables/disables external interrupt 1. The state of the SCR is displayed by a message on the LCD and an indication comes on LED1 and LED2 also.

2. If switch S2 is pressed, the delay is increased by 1 ms and the angle is increased by 18°. the light intensity of the bulb also increases. If the limit is reached, the message is displayed on the LCD.

3. For switch S3, the operation remains the same as with S2, but the delay is decreased by 1 ms and the angle is decreased by 18°.

Int1 function is also an interrupt function and is automatically called when the zero-crossing detector gives the pulse after every 10 ms. It feeds one pulse to the gate of the SCR after the desired delay (set by switch S2 and S3). The pulse applied is indicated on LED1.

writecmd function sends the command byte to the LCD. It takes one argument byte and sends it to port P1

writedata function sends data bytes to be displayed on the LCD. It also takes one argument byte and sends it to port P1.

writestr function writes a whole string (message) on the LCD. It takes the pointer as an argument that points the address of the first character of the string. Then through the pointer, it sends all the characters, one by one, to port P0.

busy function checks the status of the busy flag of the LCD. If the flag is set, it means the LCD is not ready and the programs remain within the loop. When the flag is reset, it means the LCD is ready and the program comes out of the loop.

keydly function, used for key debouncing, is the fix delay by approximately 100 ms.

delay function is a variable delay generated by timer 0. The basic delay is of 1 ms, which is rotated in the loop from 1 to 9 times to generate a minimum of 1 ms and a maximum of 9 ms delay.

display function separates each digit of the angle and converts them into an equivalent ASCII number, before sending it to the LCD, one by one, for display. EFY

Downloads: Code Files n PCB

Solar LED Lantern with battery charger

noreply@blogger.com (Anonymous) — Sat, 08 Sep 2012 09:08:00 +0000

This solar LED lantern can be used as an emergency light. Its 6V battery can be charged either from 230V, 50Hz AC mains or a 12V, 10W solar panel. Two LED indicators have been provided—red LED (LED1) indicates battery charging and green LED (LED2) indicates fully-charged battery.

You can choose to charge the battery either from the mains power or the solar panel by using the single-pole, double-throw (SPDT) switch. Capacitor C1 (1000µF, 35V) removes ripples from the power supply and regulator IC LM7809 (IC1) provides regulated 9V DC to the emitter of pnp transistor T1 (TIP127/BD140) and pin 7 of op-amp IC CA3140 (IC2), which is configured in comparator mode.

The reference voltage of 6.3V at pin 2 of IC2 is obtained through the combination of resistor R7 (1-kilo-ohm) and zener diode ZD1 (6.3V). The comparator controls charging of the battery. Pin 3 of IC2 is connected to the positive terminal of the battery to be charged through resistor R5. When the battery is fully charged, it stops charging and the green LED (LED2) glows to indicate the full-charge status.

When the battery voltage is low, diode D1 (1N4007) forward-biases and the battery connects (through resistor R3) to the collector of T1 for charging (indicated by the glowing of red LED1). Three high-wattage white LEDs (LED3 through LED5), such as KLHP3433 from Kwality Photonics, are used for lighting. These are switched on using switch S3.

Cellphone-Based Device Control With Voice Acknowledgement

noreply@blogger.com (Anonymous) — Sat, 01 Sep 2012 09:04:00 +0000

Here is a circuit that lets you operate your home appliances like lights and water pump from your office or any other remote place. So if you forgot to switch off the lights or other appliances while going out, it helps you to turn off the appliance with your cellphone. Your cellphone works as the remote control for your home appliances. You can control the desired appliance by pressing the corresponding key. The system also gives you voice acknowledgement of the appliance status.

Circuit description
Fig.1 shows the circuit for cellphone based device control with voice acknowledgement. It comprises microcontroller AT89C51, DTMF decoder MT8870, voice recording/playback device APR9600 and a few discrete components.

Microcontroller AT89C51 is at the heart of the circuit. It is a low-power, high-performance, 8-bit microcontroller with 4 kB of flash programmable and erasable read-only memory (PEROM) used as on-chip program memory, 128 bytes of RAM used as internal data memory, 32 individually programmable input/output (I/O) lines divided into four 8-bit ports, two 16-bit programmable timers/counters, a five-vector two-level interrupt architecture, on-chip oscillator and clock circuitry. A 11.0592MHz crystal (X_TAL1_{) is used to provide basic clock frequency for the microcontroller. Capacitor C3 and resistor R3 form the power-on reset circuit, while push-to-on switch S20 is used for manual reset.

Port pins P1.0 through P1.7 of the microcontroller are configured to get the input from push-to-on switches S1 through S8. Pins of Port P1 are pulled high via resistor network RNW1. Port pins P2.0 through P2.4 are configured to receive the decoded DTMF signal from DTMF receiver MT8870. The functions of the corresponding switches (S1 through S8) and cellphone keys are shown in Table I.}

_{The DTMF decoder is used for decoding the mobile signal. It gets DTMF tone from the mobile headset’s speaker pins and decodes it into 4-bit digital signal. The DTMF decoder is operated with a 3.579MHz crystal (}X_{TAL2 )}In DTMF receiver MT8870 (IC3), capacitor C12 is used to filter the noise and resistors R6 and R7 help to amplify the input signal using the internal amplifier.

Pin 16 of IC3 connected to resistor R5 provides the early steering output. It goes high immediately when the digital algorithm detects a valid tone pair (signal condition). Any momentary loss of signal condition causes ESt to return to low state.
Pin 17 of IC3 connected to capacitor C11 is bidirectional, acting as steering input/guard time output (St/GT). A steering logic V_TSt detected at St causes the device to register the detected tone pair. The guard time output resets the external steering time constant, and its state is a function of ESt and the voltage at St.

Port P3 pins P3.6 and P3.7 of IC1 are configured to select the control source for the devices. These are connected to DIP switches S17 and S18 and pulled high via resistors R2 and R1, respectively. Here, we are using two control sources, switches and mobile’s key. DIP switches S17 and S18 select the control sources as shown in Table II.

Pin 2.5 of Port P2 is configured to show the rest status. That is, if none of the control sources is selected by DIP switches S17 and S18, LED1 glows. Resistor R14 limits the current through LED1.

Voice acknowledgement is provided by the APR9600 (IC2). It is a single-chip voice recording and playback device that can record and play multiple messages at random or in sequential mode for 60 seconds. The user can select sample rates with corresponding quality recording lengths. Microphone amplifier, automatic gain control (AGC) circuits, internal anti aliasing filter, internal output amplifier and message management are some of the features of the APR9600.

Here the APR9600 is configured in random-access mode, which supports two, four and eight messages of fixed durations. The length of each message is the total recording length available divided by the total number of memory segments/tracks enabled.

Audio processor APR9600 can store up to eight voice messages. Port P0 pins and P2.7 are configured to communicate with IC2. Port P0 pins trigger selection of the message. Port pin P2.7 is the input signal to identify whether the voice message is playing or not.

Pins P3.0 through P3.5 of Port P3 control the devices with the help of relays RL1 through RL6 via relay driver IC4.

A speaker is connected to IC2 for audio output. The speaker output drives the mic input of the mobile for audio acknowledgement. An electret microphone MIC1 is connected to IC2 to record the voice in IC2. LED2 flashes to show the busy status of IC2 during recording and playback. The audio messages to be recorded in APR9600, by using trigger switches S9 through S16, are shown in Table III. SPST switch S19 is closed for recording and switch S19 is opened for playback.

Fig.3 shows the power supply circuit. The 230V AC mains is stepped down by transformer X1 to deliver the secondary output of 9V, 500 mA. The transformer output is rectified by a full-wave bridge rectifier comprising diodes D1 through D4, filtered by capacitor C16 and then regulated by IC 7806 (IC5). Capacitor C15 bypasses the ripples present in the regulated 6V power supply. LED3 acts as a power-on indicator and resistor R16 limits the current through LED3.

Recording and playback

To record the voice in IC2, follow Table III. Close SPST switch S19 to make pin 27 of IC2 low. Thereafter, press and hold switches S9 through S16 to record corresponding voice messages. LED2 flashes to indicate audio recording.

For playback of any device status, open SPST switch S19 and press the corresponding switch (S9 through S16). The recorded audio can be heard from the speaker connected to pins 14 and 15 of IC2. Fig.2 shows the pin configuration of mobile headset.

Software

The program (Device_Control.BAS) for the microcontroller is written using BASCOM microcontroller programming software. In the program, first, initialise the port (P0-P3) for corresponding controls. Thereafter, declare the variables for the program. After declaration, assign some initial value to variables. Here, microcontroller ports are initialised to make all the devices ‘off’ initially.

After that, the main function checks through ‘Do’ loop which control source has been enabled by using DIP switch pins. If you select switch S17, it searches the input from the mobile only. If you select switch S18, it searches the input from the switches (S1 through S8) only. If you enable both switch S17 and switch S18, it searches the inputs from switches and mobile. Else, the rest-status LED1 glows. Refer to Table II to select the control source.

The mobile signal is decoded into the DTMF signal by IC3. The DTMF output for each mobile key (used in this project) pressed is shown in Table IV.

After getting the input from the switches or mobile, the program goes to the device_action subroutine and executes the corresponding action (refer Table I).

The device_action subroutine changes the status of the device and calls the voice_alert subroutine. The voice_alert subroutine checks the device status and device name from the source input and controls the corresponding pins of IC2. First, it selects the voice signal for the device name. After playing that, it selects on/off status of corresponding device as mentioned in Table III.

If you press ‘*’ key followed by the device number on your mobile handset, it will not change the status of that device and inform the current device status. If you press device number followed by ‘*’ key on your mobile handset, it will change the status of that device and inform the changed device status. ‘#’ key controls the voice_control subroutine and acts like a mute key.

Downloads:code files +PCB

Versatile 200KHz Function Generator using ICL8038

noreply@blogger.com (Anonymous) — Wed, 20 Jun 2012 17:14:00 +0000

ONE of the most useful facilities in the electronic experimenter’s workshop is an ability to generate a.c. test signals of various waveforms, frequencies and amplitudes. This is where this Function Generator project comes in, the “function” in the name referring to the waveform of its output signal.

The Generator can be used for testing or driving many circuits, from below audio up to a couple of hundred kilohertz, and may even be used as a variable speed clock for logic circuit testing. It has sine, square and triangle-wave outputs plus a separate 0V to +5V squarewave output for logic driving. This may also be used with a “sync” input when, for example, inspecting low-level signals on an oscilloscope.

Circuit Detail:

The full circuit diagram is shown in Fig.1. Starting with the power supply, this is of the dual rail type. The very low output frequencies require d.c. coupling in the output circuit, which in turn makes a circuit using separate positive and negative supplies about a central 0V or ground much simpler to design.

Transformer T1 has a 15V-0V-15V 100mA centre-tapped output, which after rectification by REC1 produces raw positive and negative supplies of about 22V each across capacitors C1 and C3. The voltage regulators IC1 and IC2 reduce these to plus and minus 12V.

Function Generator Schematic

Since the supply voltages must be symmetrical to avoid a d.c. offset voltage at the output an adjustable regulator is used for IC2, which can be trimmed with preset VR1 to achieve this. These regulators are 1A and 1·5A types which can dissipate the small amount of internally generated heat without additional heatsinks. Their outputs have the usual decoupling capacitors, C5 to C8, and the l.e.d. D1 indicates when the unit is powered.

Waveform Generation:

Most of the waveform generation is carried out by IC3, an 8038 dedicated waveform generator chip. The output frequency is determined by the voltage applied to the Frequency Sweep input at pin 8 and the value of the capacitor connected between the negative supply rail and pin 10.

The required range of control voltage is applied by the panel mounted Frequency control VR3 with minimum and maximum values adjustable with presets VR2 and VR4 for the desired frequency range. Note that the frequency rises as the control voltage is lowered and vice-versa, which explains why preset VR4, Set High Freq., is at the bottom.

Seven frequency ranges are provided, in decade steps, by rotary switch S2 with associated capacitors C12 to C18. Some of these capacitors can be obtained in 1% tolerance, others may have to be 5% or 10%, and for the bottom two ranges electrolytics are used. These are notoriously inaccurate, so where these ranges are to be provided it helps if a capacitance meter is available to pick ones reasonably close to the correct value.

The top range is set with capacitor C11, which is slightly lower than the expected value to compensate for stray capacitance in the circuit. The value shown proved to be about right in the prototype.

Although the 8038 is capable of quite good waveform linearity without adjustment, some worthwhile improvement can be achieved by external trimming. Presets VR5, VR6 and VR7 are provided for minimising sinewave distortion. The sinewave output is taken from pin 2 and the triangle output from pin 3.

Square Wave:

The squarewave output is usually taken from pin 9. This is an “open-collector” output which requires a pull-up resistor to the positive supply so the speed of the rising edge depends to some extent on the value of this resistor. A sufficiently low value was found to result in some distortion of the sinewave output during development of this project so it was decided not to use it. Instead, the triangle output at IC3 pin 3 is fed to the inputs of the LM393 dual comparator IC4 at pin 3 and pin 5, via resistors R10 and R11.

One of the comparator outputs, from pin 1, is taken to the Waveform selector switch S3. The other, pin 7, drives the base of transistor TR1 which gives a 0V to +5V squarewave output from SK1 for driving logic circuits. This output may also be used for synchronising external equipment such as oscilloscopes when working with low level signals.

A couple of minor precautions are included in this part of the circuit to minimise breakthrough of the squarewave into the other waveforms. Resistors R10 and R11 eliminate a problem of feedback which appeared to be from the inputs of IC4 back into the triangle waveform, and use of a two-pole switch for the waveform selector S3 allows the squarewave signal from resistor R17 to be “grounded” through S3b when not selected.

Range Selection:

The three signals, Square, Triangle and Sinewave, all have different peak-to-peak amplitudes at this point. Op.amp IC5 has a gain set by resistors R20 and R21 so that the sinewave leaves it at 10V pk-pk, then the two resistors R16 and R17 are used to attenuate
the other two signals in conjunction with resistor R19 to obtain the same pk-pk level.

Op.amp IC5 is an Elantec EL2045 which is a high-speed type. A TL071 can be used here, but the faster component offers less distortion of the triangle and squarewave signals at high frequencies. Potentiometer VR8 provides variable control of the output amplitude with decade ranges added by the attenuator network built around Amplitude Range switch S4 and resistors R22 to R26.

Finally, a buffer stage using transistors TR2 to TR5 and resistors R27 to R33 gives the circuit a 50-ohm output. The complementary design of this stage minimises quiescent current and keeps distortion to a minimum, even at high frequencies.

First Tests:

Always remember to disconnect the unit from the mains before making any adjustments. Do not come into contact with the 230V a.c. input to the transformer.

Normally the author recommends initial testing with a current-limited bench power supply but this is a bit difficult with a dual supply rail design so this project was just connected to its transformer (T1) and powered up, testing each part of the circuit in turn. It is assumed that a DVM and an oscilloscope will be available as a function generator is usually used in conjunction with a ’scope anyway.

The first check is to connect the transformer and power it up. Around 22V d.c. should be present with respect to ground or 0V, positive at the top of capacitor C4 and negative at the bottom of C2.

If these voltages appear correct IC1 can be fitted and on powering the board again the regulated negative 12V should appear at the bottom of C5. It should also appear at pin 4 of the sockets for IC4 and IC5, and pin 11 of the socket for IC3.

Next IC2 should be fitted. This time the regulated positive voltage should be present at the top of capacitor C6 and on pin 8 of IC4’s socket, pin 7 for IC5, and pin 6 for IC3. It probably won’t be exactly 12V, but it can now be adjusted to approximately this value with preset VR1.

Wave Form Check:

Frequency control VR3 should now be temporarily connected along with a 22nF 1% capacitor across the two connection points for switch S2. All of these points can be readily identified from the connections diagram shown in Fig.3. IC3 should now be inserted into its socket.

When the circuit is now powered up, the sinewave (around 5V pk-pk) and triangle wave (8V pk-pk) should be present at their connection points for switch S3. The 22nF capacitor is the value for 200Hz to 2kHz, so Frequency control VR3 should give a range somewhere around this, though it may not be accurate until the presets VR2 and VR4 have been adjusted.

Next IC4 can be fitted. Both outputs of this, pins 1 and 7, should produce squarewaves with an amplitude of about 20V pk-pk. If transistor TR1 is now fitted, the 5V pk-pk squarewave output should appear from the connections for socket SK1. Note that this swings between ground (0V) and +5V, not symmetrically about 0V as the other waveforms will.

If IC5 is now fitted and an oscilloscope used to view its output (pin 6, but helpfully it also appears at the connection point for the top of amplitude control VR8), connecting the input (the connection point immediately above it) to any of the three waveform outputs for S3 should produce the appropriate output waveform at about 10V pk-pk.

The last stage of testing is to fit the remaining four transistors, taking care with type and orientation as there are two pnp and two npn types, and to connect the output from IC5 directly to the input to this stage (the connection point for the pole of S4). When the input to IC5 is connected to each waveform as before, it should appear at the final output, connection point for SK2, with a 10V pk-pk amplitude. This completes the board testing. It may be found that IC1 to IC3 run very slightly warm, this is normal.

Output Attenuator:

The resistors (R22 to R26) for the output attenuator are soldered directly to switch S4 as shown in the wiring diagram Fig.3. This also shows the connections of the frequency range capacitors C12 to C18 to frequency range switch S2, though these are shown positioned radially for clarity. In fact there is not room for this, so they are assembled pointing backwards from the switch, using a piece of fairly thick tinned copper wire soldered to an unused switch connection (position 11) for their common or negative connection point. This should be clear from the photographs.

Frequency switch S2 is a single-pole 12-way switch with its limiting device set to give seven ways. Although C12 is shown as a single component in the circuit diagram it actually consists of a 150pF and a 33pF capacitor connected in parallel, this combined value being selected for the correct range by trial and error after the rest of the circuit was complete and fully adjusted.

Final Adjustments:

The project is now ready for final adjustment, where the preset resistors are trimmed to obtain the optimum performance. Before commencing the 1kHz frequency range (using C14, 22n 1%) should be selected with switch S2, and Frequency control VR3 should be set to around midtravel so that the unit is operating at about 1kHz.

Preset VR1 should have previously been set to give a positive supply of about 12V, and the remaining presets VR2, VR4, VR5,VR6 and VR7 should all be set to about half travel. Multi-turn types seem to be supplied already set to this, but if doubt exists they can be turned in one direction until a “click” is heard and then turned back again for half their total number of turns.

The triangle output waveform should be selected with S3 and the project allowed a few minutes to settle to its normal operating temperature following power-up. The first adjustment consists of connecting a meter between ground (0V) and the output of IC5 (pin 6), and carefully adjusting preset VR1 to obtain an average d.c. voltage of zero at this point, i.e. no d.c. offset in the output waveform.

Sinewave adjustment is carried out using presets VR5, VR6 and VR7. Presets VR6 and VR7 adjust the shapes of the positive and negative output half cycles respectively, clockwise rotation giving them a more “pointed” shape, and anticlockwise a more “rounded” one. Preset VR5 has less effect but actually adjusts both duty cycle and frequency to a small extent.

Connect an oscilloscope to the output at SK1 and adjust VR6 and VR7 for the best sinewave shape. Presets VR6 and VR7 interact to some extent so repeated adjustment will be required.

EPE

Versatile Bench Power Supply

noreply@blogger.com (Anonymous) — Tue, 12 Jun 2012 17:08:00 +0000

The power supply described here provides an output of up to 20V with current in two ranges of 100mA or 1A. Ideally a workbench supply should have an output voltage that can be adjusted right down to zero as it is occasionally useful to be able to power a circuit gradually from this when fault-finding. There should also be a fast and effective current limiting facility, again adjustable from zero as this also provides valuable protection when testing circuits.

If the supply can be set to deliver a constant output current instead of a voltage it can also be used as a charger for the many types of re-chargeable battery that are available nowadays, both alkaline and small sealed lead-acid types. The latter are normally charged to a constant voltage but until this is reached a current limit is often required to prevent an excessive charge rate.

Preferably the supply should have two meters so that the voltage and current supplied to the load can be seen at a glance, and most users would probably prefer analogue meters to the digital type for this.

Power supplies meeting these specifications can be expensive, but this project can be constructed at reasonable cost, especially if some of the parts such as the case and meters are already to hand, or are purchased from inexpensive sources such as surplus stores or amateur radio rallies.

Design Points:
The design of a good power supply is more complex than might be supposed. It must provide a constant voltage or current whilst being able to react rapidly to any changes in the load it is supplying. This means that the circuit must have a fast response and remain stable for a wide range of output loads, voltages and currents, so the circuit is inevitably a compromise between speed and stability.

The design used for this project should be fast enough for most purposes and has good output stability.

Better Control:
An arrangement which overcomes the design difficulties is shown in Fig.1. It uses two power supplies, a main one of about 25V capable of supplying up to 1A of current for the output, and an auxiliary split supply of +12V for the controlling circuit. The 0V rail of this is the ground for the control section. It operates as follows:

A reference voltage is produced using regulator IC1 and applied to resistor R1. During normal operation op.amp IC2 uses feedback to maintain equal voltages at its inputs, so a constant current flows into R1. This current also flows through resistor R2, developing a voltage across it directly proportional to its value. If a linear variable resistor (potentiometer) is used for R2 it will provide linear control of the output.

Perhaps the easiest way to understand the action of the circuit is to consider what would happen if the voltage at the junction of R1 and R2 were to rise slightly. The output of IC2 would start to rise and thus turn on the MOSFET TR1 a little more. This would cause the 0V rail to rise with respect to the main negative supply rail so that more current would flow through R2 to bring the input to IC2 down again, restoring the balance.

The 0V rail, of course, is the positive output so the voltage developed across R2 is what is delivered to the load. Naturally, as the 0V rail rises with respect to the main supply, so do the positive and negative 12V rails, so that even when the output voltage is close to the maximum of the main supply there will still be a full 12V available for IC2 to use in controlling the gate of power MOSFET TR1.

Output Current:
It is also simple to measure the output current in this arrangement using the sensing resistor R3, which develops a current dependent voltage with respect to the control circuit ground. The only disadvantage of this arrangement is that the return path for the reference current from resistor R1 is via the main supply, TR1 and R3, so this current should be kept small to prevent any noticeable effect on the measured output current. This also applies to the voltmeter which is connected directly across the output.

Mains Supply:
The full circuit diagram for the Versatile Bench Power Supply is shown in Fig.2. The incoming 230V a.c. mains passes through a 2-pole isolating switch S1 to the two transformers T1 and T2, whilst the transient suppressor VDR1 removes any brief high voltage spikes that may occur.

No supply fuses are fitted to the prototype, a 3A fuse in the mains plug being considered adequate. Transformer T1 is a 20V 20VA (1A) type and, with bridge rectifier REC1 and reservoir capacitor C2, produces a no-load voltage of about 30V. Under full-load conditions this drops to about 24V which is still sufficient to maintain the 20V output. Capacitor C1 and resistor R1 also help to eliminate transient voltage spikes.

Transformer T2 is a smaller 100mA type with a centre-tapped 9V-0V-9V output. Arranged as shown with bridge rectifier REC2, it develops both positive and negative supplies of about 12V each. A 5V reference voltage is generated by IC1, a standard 78L05 regulator.

Earthing:
The earthing arrangements used in this project are slightly unusual and require some explanation. Earthing of a mains powered project is essential both for safety and because capacitive coupling between windings in the transformer can transfer potentially damaging a.c. voltages to the output, even though from a very high impedance.

However, the output of a bench power supply is often required to have d.c. isolation from earth so that other earthed equipment, such as oscilloscopes, can be safely connected to any part of the circuit on test.

Commercially produced bench supplies often overcome this problem by using transformers with internal foil screens between the primary and secondary windings which are earthed to eliminate the capacitive coupling. These are not readily available to the home constructor, so a different method must be used.

Switch S2 allows the mains earth to be connected directly to either the positive or negative output terminal or left floating, where capacitively coupled a.c. voltage is grounded through capacitors C3, C4 and resistor R2. Although not ideal, this system works well in practice and reduces noise and a.c. voltage at the output to a few millivolts at most. For safety, capacitors C3 and C4 must be 250V class X2 suppression types.

Voltage Control:
The voltage controlling part of the circuit is constructed around op.amp IC2 and operates as described earlier. The resistance to which the reference voltage is applied includes the preset VR1 so that the maximum output voltage may be set to exactly 20V.

The input to IC2 has a pair of protection diodes, D1 and D2. Capacitor C12 ensures stability but is small enough to allow a fast response to output load changes. Power MOSFET TR1 handles the full output current. Output voltage is indicated by meter ME1 with VR3 allowing preset adjustment for calibration to 20V full-scale.

Load current is sensed by the one ohm resistor R10. At full output this dissipates a watt of heat, so a 7W type is used to dissipate this safely. A potential divider across R10, consisting of R12 with R13 and R14 in parallel, allows selection of either the full potential across it or one tenth through switch S3, giving the 100mA and 1A current ranges.

The voltage from S3 is compared by op.amp IC3 with an adjustable voltage derived from the reference by divider network VR4, R16 and the current control VR5. When the S3 voltage exceeds this secondary reference voltage, the output of IC3 goes low, pulling the gate voltage of TR1 down through diode D3 to limit the current flow through this transistor.

It also turns on transistor TR2, which illuminates light emitting diode (l.e.d.) D7 to indicate that current limiting is taking place. Preset VR4 allows the current ranges to be set precisely to 100mA and 1A, whilst VR5 gives control from zero to the selected maximum.

Op.amp IC4 also receives the voltage from S3, and converts it into a 1-100uA drive for the current indication meter ME2. Preset VR6 allows calibration of this meter for full-scale corresponding to the 100mA and 1A signal inputs. The high value of resistor R21 prevents any possibility of meter damage through overdriving.

Double-pole switch S4 allows the output to be switched off independently and also provides instant isolation from the circuit connected to it at any time.

Zener diode D6 protects TR1 from any high voltage pulses arising from the connected load. This can happen with some types of inductive load and, whilst normally tolerant of brief high currents, semiconductors are easily damaged by excessive voltage.

Testing:
The aim of using separate op.amps for the various functions is partly to make testing and trouble shooting easier, so hopefully this will be the case.

The first test is to connect the 9V-0V-9V transformer as shown in Fig.3. Using 0V (the centre-tap connection) as a reference, the circuit should be powered up and the presence of +12V and -12V checked at pins 7 and 4 respectively of the three d.i.l. sockets.

The presence of +5V from regulator IC1 can also be checked at its output pin. Next, the MOSFET TR1 and VR2, the Voltage control, should be connected, along with the two leads from the main power supply, from REC1 and C2. IC2 should now be inserted into its socket.

Whilst monitoring the output with a voltmeter, the circuit should be powered up, and it should now be possible to vary the output from zero to around 20V with Voltage control VR2. If so, VR2 can be turned right up and the voltage set to exactly 20V with preset VR1. If the voltmeter ME1 is now connected this can be adjusted for an indication of 20V (full scale) with preset VR3.

Next the current range switch S3 should be connected and set to 100mA, and the current control VR5 should be connected. The current control op.amp IC3 should be inserted into its socket, and a meter set to read a current of about 100mA connected in series with a 100 ohm resistor across the output. The circuit should again be powered and the voltage control turned right up.

Potentiometer VR5 should now control the current, from zero to around 100mA. Preset VR4 can be adjusted for a maximum output of exactly 100mA. The 1A range should automatically be correct following this. The l.e.d. D7 can be connected, this should light whenever current limiting is active.

Finally, the current meter ME2 should be connected, op.amp IC4 inserted, and preset VR6 set for full scale at 100mA. As before, the 1A range should now automatically be correct following this adjustment.

Current Limitations:
It should be noted that the current limit takes a finite time to operate so there are some applications where external limiting is needed. The output of IC3 has to come out of saturation and slew through about eight volts before limiting begins, and this will take a microsecond or two!

The limit protects the supply itself against short circuits, and in almost every practical fault situation it is fast enough to prevent damage to faulty circuits on test as these are rarely total short circuits
anyway.

However, it should be remembered that the MOSFET output device can handle very high currents and has a large capacitor behind it so, for the brief time it takes for the limit to operate, many amperes can be delivered into a short! An obvious example is the testing of l.e.d.s.

The limit should never be relied upon on its own for this, a suitable series resistor should always be used.

EPE

High Efficiency LED Torch

noreply@blogger.com (Anonymous) — Sun, 03 Jun 2012 07:12:00 +0000

THE high forward voltage of l.e.d.s is a challenge for efficient battery use. In the circuit shown in Fig.1a, a 74HC14 inverter is used to “double” the voltage of a 6V rechargeable ex-mobile-phone battery and drive a string of three 5mm l.e.d.s. The oscillator around IC1a and IC1b generates a square wave at about 2·8kHz, and its output is buffered by IC1c and IC1d used in parallel to maximise current to the doubler.

Booster
Using a 6V supply and with 1N4148 diodes in the rectifier, which comprises D1 and D2 plus capacitors C2 to C4, results in a rectified voltage of 9·6V having a 12mA output capability for a 24mA supply current – an electrical efficiency of over 80%. Using a 6·3V battery the output voltage is 10V and the available current increases to 16mA.

If Schottky diodes are used for D1 and D2, there is an increase in voltage of 100mV, resulting in a 20% increase in available current, to 19mA (6·3V supply), at similar efficiency, and a much brighter light. Using these diodes there might be benefit in increasing the oscillator frequency substantially. This is set by the values of R1 and C1.

At night, the torch casts a pool of white light two metres wide to a range of over 6m without a reflector, perfect to walk by. Side leakage is useful but seal the l.e.d.s at the rear as back-scatter is irritating. Nearby colours are vibrant compared with the gloomy monochrome of incandescent torches.

Pulsed Light
Anecdotal evidence suggests that a subjectively brighter light is obtained by low-frequency pulsing the l.e.d.s at currents higher than their continuous rating. There are two spare inverters, IC1e and IC1f, which can be wired as shown in Fig.1b. They provide variable-duty voltage pulses which directly drive a logic-level MOSFET and the string of three l.e.d.s, D7 to D9, via current limiting resistor R8.

With resistor R7 and potentiometer VR1 setting a total resistance of 3·9M , the frequency is 15Hz at 20% duty cycle. At 3mdistance from the subject the illumination appears continuous, but closer in the flicker is very obvious, like a strobe. Negotiating confined spaces, e.g. a spiral staircase, by this light is a challenge as the walls move as you climb. The output voltage is about 10·8V with a 0·8V ripple at the pulse frequency. The output current is limited by the inefficiency of the doubler.

Increasing the frequency to 35Hz (R7 + VR1 = 1·6M ) results in 35Hz, producing a good light, even to read by, though some flicker can still be seen as the duty cycle is 30%. Average current consumption is 21mA at 6·3V, with an l.e.d. current about 28mA.

Right Choice
In this application, 40106 or 74C14 CMOS chips are so markedly inferior to the 74HC14 that they are unsuitable. The published maximum voltage rating of an 74HC14 is 7·0V. The high initial voltage of a freshly charged battery may be tamed to within this limit by switching in a diode (e.g. 1N4001) until the terminal voltage has fallen somewhat.

PROGRAMMABLE TIMER FOR APPLIANCES

noreply@blogger.com (Anonymous) — Tue, 15 May 2012 06:13:00 +0000

This programmable timer is usefu for domestic, commercial as well as industrial applications. It automatically turns the appliance on/off after a preset time. The time period can be varied from 8 seconds to 2 hours with the help of rotary switches S2 and S3. The circuit works in two modes: off mode and cyclic mode. Slide switch S4 is used for mode selection.

In the off mode, the appliance turns on after a preset time (set by rotary switch S2), remains on for another preset time (set by rotary switch S3) and then turns off. In the cyclic mode, this process repeats again and again.

The circuit is built around three quad two-input NAND gate ICs CD4011 (IC1, IC3 and IC5), two 14-bit binary ripple counters CD4020 (IC2 and IC4) and a relay driver transistor (T1). It works off a 12V DC, 500mA power supply. You can also power the circuit from mains by using a 12V DC, 500mA adaptor in place of the 12V DC power supply.

Let’s assume that you want an appliance to turn on after two minutes and keep it on for another two minutes. For this set the rotary switches S2 and S3 to positions as shown in the figure.

Initially, when power switch S1 is closed, a small charging current pulse through capacitors C2 and C3 resets both the counters (IC2 and IC4) to make all their outputs (Q4 through Q14) low. The high output at pin 10 of NAND gate N3 starts the first oscillator comprising NAND gates N1 and N2, which provides clock pulses to IC2 at the rate of one pulse per second. The glowing of red LED (LED1) indicates that this oscillator is working well and timer is ‘on.’

During the first 2 minutes, relay RL1 remains de-energised by the control circuit formed by NAND gates N7, N8 and N9 and LED2 is off, which indicates that the appliance is in ‘off’ codition. The second oscillator built around NAND gates N4 and N5 (which provides clock pulses to IC4 at the rate of one pulse per second) is inhibited by the timing control circuit formed by NAND gates N6, N10 and N11.

After 128 pulses (approximately two minutes), the Q8 output of IC2 goes high to perform the following three functions:
1. Make the output at pin 10 of NAND gate N3 low via rotary switch S2, which inhibits the first oscillator.
2. Energise relay RL1 via NAND gates N8 and N9 and relay driver transistor T1 to make appliance ‘on’.
3. Make the output at pin 10 of NAND gate N10 low, which is connected to the inputs of NAND gate N11 to make its output at pin 11 high. This high output is further connected to the input (pin 1) of NAND gate N4.

Now the second oscillator starts oscillating and provides clock pulses to pin 10 of IC4 at the rate of one pulse per second.

Now, after 128 pulses (approximately two minutes), the Q8 output of IC4 goes high. This de-energises the relay via NAND gates N7 and N9 and relay driver transistor T1, provided the mode-selectio slide switch S4 is towards off position. The high Q8 output will inhibit the second oscillator via NAND gates N6, N10 and N11 to stop clock pulses to pin 10 of IC4. Thus the relay is energised only once (for 2 minutes) since clock pulses to both IC2 and IC4 are stopped altogether and their outputs get latched.

In case the mode-selector switch S4 is towards ‘cycle on’ side, clock pulses to IC4 would continue and the relay is alternately energised and de-energised for two minutes each. This continues until the circuit is switched off and started again, or the mode-selector switch is slided towards ‘cycle off’ side.

Rotary switch S2 is used for start time selection and rotary switch S3 is used for hold time selection. The start and hold time can be increased up to 24 hours by changing the values of R and C components of the oscillator circuit of first and second oscillator.

Time Delay Touch Switch

noreply@blogger.com (Anonymous) — Fri, 11 May 2012 10:55:00 +0000

Fitting switches which turn off automatically after a preset time in these locations can reduce the electricity consumed quite significantly. A single 100W bulb left burning all night (say eight hours) would consume 800W/hrs. The same bulb fitted with a switch which turned off automatically after say five minutes would consume only 83W/hrs, even in the unlikely event that it were to be activated ten times during this period.

Triac Control
There are two major problems to solve when designing a touch switch to replace a conventional mechanical light switch and these concern the power supply and noise immunity.

The main switching element used to control the light is a triac, a semiconductor a.c. switch. This is normally in its off or non-conducting state but can be switched into conduction by means of a small trigger current fed to its gate terminal. Once triggered it will remain conducting until the current through it drops below a certain value (known as the holding current). This occurs normally on a.c. supplies when the voltage drops to zero at the end of every mains half-cycle, and the triac must be retriggered again if the light is to remain on.

As with any switch, the voltage across a triac when it is in its off state will be the supply voltage (in this case 230V a.c.). When the triac is on, the voltage across it will be around 1V a.c., depending on the current. The triac therefore dissipates very little power so no heatsink is required.

With normal house wiring, only the Live wire and its feed to the lamp are accessible at the wall-mounted light switch. The Neutral wire from the lamp is left inaccessible in the ceiling. Thus it is not possible to simply reduce and rectify the mains to provide a d.c. voltage for the triac control circuit, which must therefore be powered “through” the lamp, i.e. from the voltage across the triac.

An adequate d.c. voltage can be derived by delaying the point at which the triac is triggered in a mains half-cycle. This allows a higher voltage to develop across it which can be rectified and used to power the rest of the circuit.

The disadvantage is that it gives rise to EMI (electromagnetic interference) which can interfere with radio reception and computer systems. As a result, bulky and expensive chokes and capacitors have to be fitted to suppress it. This is unavoidable in light dimmers where we have to delay the triggering so that the power to the lamp can be controlled. For an on/off control, though, triggering should occur as early as possible so that full power is available to the lamp and without interference being generated.

In Control:
In this circuit, a small Zener diode is connected in series with the triac. This causes sufficient voltage drop so that after rectification a 3V to 4V d.c. supply is available for the control circuits. The Zener has the load (light) current flowing through it during operation and therefore dissipates power, but for the kind of currents used in lighting this only amounts to a few hundred milliwatts, a value well within the ratings of small wire-ended Zener diodes.

The circuit provides sufficient current to enable the triac to be triggered by d.c. pulses instead of the short a.c. trigger pulses normally used. As a result, the triac does not turn off between half-cycles, and so does not cause interference to the mains supply. This point is important as it also enables the circuit is to be used with low power fluorescent lamps if required.

Touch Control:
Touch control circuits, by their very nature, tend to be prone to spurious switching in response to the many glitches and pulses which normally occur on the mains supply due to other equipment being switched on or off. This is often difficult to overcome, but it can be very much reduced by building a delay into the circuit so that short pulses, which characterise most of the interference, are ignored and the circuit only responds to longer inputs.

By fitting a delay of a few hundred milliseconds, users will not notice any delay in switching, and interference, which rarely lasts for more than a few tens of milliseconds, will be largely ignored by the circuit. This is not a complete solution to the problem, but in this case there is also the advantage that even if the circuit should switch on accidentally, it will only remain on for a few minutes anyway!!!.

Circuit Description:
The circuit diagram for the Time Delay Touch Switch is shown in Fig.1. Triac CSR1 and Zener diode D4 are connected in series with the lamp as previously discussed. The voltage appearing across the Zener is rectified and smoothed by diode D3 and capacitor C4 to provide a d.c. supply of around 4V.

As this voltage only appears when the triac is on, capacitor C5 is connected as a capacitive mains dropper to provide a small current to the Zener to maintain the d.c. supply when the triac is off. Resistor R6 is included to limit the Zener current should the circuit be powered when the capacitor is discharged (very likely) and also to limit the discharge current which could destroy the triac when it switches on (also very likely).

The rest of the circuit is quite conventional and is based on a quad CMOS Schmitt trigger NAND gate, although the gates are used as logic inverters by having both inputs connected together. This type of gate has the advantage of switching cleanly once the input threshold voltage has been exceeded.

The input sensor is formed by placing a conducting foil behind the insulating front of a standard light switch blanking plate. The foil forms one plate of a capacitor. When it is approached by a hand, which becomes the other plate, the capacitor formed effectively connects the base of transistor TR1 to earth. (The human body is a fairly good conductor and exhibits a capacitance to earth of around 100pF, depending on size and area!)

Since the emitter circuit of npn transistor TR1 is connected via resistor R2 to the Live rail, which is at 230V a.c. with respect to earth, this is equivalent to a 230V input to the base of TR1 via the combined capacitance of the body plus that the hand and foil.

Not surprisingly, transistor TR1 turns on when the input polarity is correct and charges capacitor C1. Resistor R1 and preset resistor VR1 set the sensitivity of the circuit, allowing it to be activated by placing one finger or the whole hand on the plate. Resistor R2 discharges the capacitor when the hand is removed.

When the input threshold of gate IC1a is exceeded, the output switches low causing capacitor C2 to charge via resistor R3, which introduces the switch-on delay required to improve the noise immunity of the circuit. Should the output of IC1a go high again because TR1 turned on for only a short period due to noise, C2 would be quickly discharged via diode D1 so that the input of IC1b would not have time to go low.

Assuming a moderately long input period during which the output of IC1a remains low, however, IC1b’s input will also become low, causing its output to go high, so quickly charging capacitor C3 via diode D2. This, of course, causes the output of IC1c to go low and allow gate current to flow to the triac via pnp transistor TR2 and resistor R5. In turn, this causes the triac to be triggered and so switch on the lamp.

Note that the circuit uses negative triggering (i.e. a current flows out of the triac gate rather than into it) because triacs are generally more sensitive in this mode and require less gate current to ensure triggering.

When the hand is removed from the touch plate, diode D2 will become reverse biased and capacitor C3 will only be able to discharge via resistor R4 and preset VR2. These two components, together with C3, therefore determine the length of time for which the triac remains triggered and the lamp turned on.

With the components specified, the turn-on time can be made variable from around 20 seconds to 2¼ minutes. This may be increased if required by increasing either R4 or C3, or both.

Note also that the time can easily be extended by touching the plate during the on period. This will recharge C3 so that if you stop on the stairs for a chat, you need not be plunged into darkness half way through the conversation!

In any event, even if this should happen, with the triac turned off 230V a.c. will appear across it and the small neon lamp LP1 will light. This is arranged to shine through a hole in the printed circuit board, illuminating the touch area so that it should be an easy matter to find the switch in the dark. Resistor R7 limits the current through the neon, which generally strikes at about 90V.

Touch Plate:
The touch plate assembly is made by using a piece of aluminium kitchen foil about 4cm square stuck behind the centre of an electrical blanking plate, as shown in Fig.2. Most blanking plates have a central area made of thinner plastic than the surrounding plate. This forms an ideal area for the neon light to shine through. (Do not make a hole in the blanking plate.) The aluminium foil should be removed from this area as shown in Fig.2. The surface of the foil should then be covered with strips of insulating tape.

A short piece of wire with about 1cm of insulation removed should be trapped beneath the insulation to enable the foil to be connected to the transistor on the circuit board. A small knot tied in the wire under the insulating tape may help to prevent the wire from being pulled out during final assembly.

Installation:
Once this has been completed, the unit can be mounted in the wall box. The mains must be switched off before the existing switch is removed, and when making any adjustments. The two cables inside the wall box should be connected to the terminal block on the unit. Because of the way the circuit operates, it is important that the Live wire is connected to the L terminal of circuit.

This wire is normally coloured red but it may not be, depending on the installation. If you are not sure which of the two wires is the Live one, connect them either way around. If you have chosen the wrong one no damage will result but the circuit will not switch on and the procedure will need to be repeated with the wires reversed.

Note that modern installations will also have an earth wire which is not connected to the switch terminals but is usually connected to a terminal on the wall box. This should be left in place and not disturbed.

Once this is done, the switch can be secured to the wall box and the mains switched on. Make sure that the light can be switched on when the plate is touched and ensure that it goes out at the end of the previously set time.

If the circuit cannot be switched on, increase the sensitivity of the circuit by turning VR1 clockwise (switch off the mains before doing this). If it is still impossible to switch the circuit on, turn the preset fully anticlockwise and reverse the wires to the terminal block before repeating the procedure.

Do not increase the sensitivity beyond the point at which it switches on easily as this can make it more susceptible to random switching, or even prevent the unit from switching off at the end of the preset period. The latter can be increased by turning VR2 clockwise.

Should you wish to use the unit with a two-way switching circuit, to enable the lights to be switched on and off from two locations, this can be done quite easily with no re-wiring being involved. Fig.3 shows a typical two-way switching arrangement and the new switches may be used to replace the existing switches as shown.

The wire colours stated in Fig.3 are those normally used although they may be different depending on the installer. In any event, one of the existing wires will not be required and the touch switches should be wired in parallel as shown. It is important to ensure that the mains Live is connected to the correct terminal on each switch.

MAINS Failure Alarm

noreply@blogger.com (Anonymous) — Mon, 07 May 2012 16:26:00 +0000

MAINS failure alarms are often employed in situations where the removal of the mains supply from a piece of equipment can have adverse or even disastrous consequences.

Very often, the “failure” may simply be the result of the switch on the outlet socket being operated inadvertently and this may have no immediately noticeable effect on the equipment. The results of the mistake may only become evident a few hours or even days later, when it is too late.

The circuit described here overcomes all of these problems at a stroke and also does away with the need for mains plugs, or indeed any connections to the mains at all. It does so by monitoring the electric field which exists around a cable connected to the a.c. supply (whether it is carrying a current or not).

By placing it on or near to the cable of the appliance to be monitored, it will also sound the alarm if the fuse in the plug blows, the outlet is switched off or the plug disconnected. It will only fail to detect the situation where the equipment itself has been switched off via its own built-in switch. However, since many appliances such as freezers do not have on/off switches, this is not really a problem.

The circuit is extremely simple and inexpensive to build. Its simplicity and lack of any specialized components should make it attractive to many constructors who will probably already have most of the components to hand. The absence of any mains connections should make it an ideal project for a beginner.

As with many simple circuits, however, the advantages and possible uses take longer to describe than the operation of the circuit diagram which is shown in Fig.1.

The heart of the circuit is an oscillator built around two CMOS NAND gates (IC1c and IC1d) which drive a small piezo sounder, WD1. With the values specified for resistor R5 and capacitor C3, the circuit oscillates at around 200Hz, producing a fairly loud sound from WD1.

Since an intermittent sound is better at gaining attention than a louder continuous alarm, this oscillator is driven by another similar circuit built around IC1a and IC1b. Here the frequency determining components, R3 and C2, have been chosen to give a lower frequency so that when the circuit is activated, the sounder produces a tone of 200Hz in bursts at around 2Hz (i.e. two bursts per second).

Mains Field
Oscillator IC1a/b is normally disabled, however, because of the action of transistors TR1 and TR2. These are switched on in the presence of the alternating electric field which exists around all cables carrying a mains potential, manifesting itself in high gain amplifier systems as “mains hum”.

In this circuit, transistor TR1 is used as the high gain amplifier, providing base current for TR2. There is a combined current gain of around 10,000.

With a high value of collector load resistance (VR1 plus R1), transistor TR2 does not need to pass very much current before its collector voltage falls below the nominal logic-high level for CMOS gates (approximately half of the supply voltage).

When the input to TR1 is placed near enough to a cable carrying a mains voltage, the positive mains half-cycles are amplified and TR2’s collector voltage falls, so disabling the oscillator around IC1a/b.

Capacitor C1 prevents the collector voltage from rising again during the negative half cycles.
Should the mains field cease, TR1 and TR2 will remain off and C1 will charge to the circuit’s d.c. supply voltage via VR1 and R1, allowing IC1a/b to oscillate and the alarm to sound.

Sensitivity
The sensitivity of the circuit depends on the value of the collector load and is best determined experimentally, so it has been made variable using preset VR1. In most cases, unless the mains field is generally very high, it will probably not need to be reduced from its maximum 470k ohm value.

Resistor R1 has been included to prevent a direct short-circuit between the two power lines via TR2 should VR1 be set to zero resistance.

Because capacitor C1 takes time to charge via VR1 and R1, there will be a delay of about one second (depending on the setting of VR1) before the alarm sounds following a mains failure.

The circuit can be powered by a PP3 9V battery, or two AA cells in series providing 3V, depending on the user’s preference. With the lower supply voltage, however, the sound produced will not be as loud. Since the current consumption in the stand-by mode is around 45uA (at 9V) the battery should last many months.

Testing
For initial tests, keep the unit well away from any potential source of mains electrical field. Solder a short length (say 100cm) of single core insulated wire to the base connection to TR1 to act as an “aerial”. Set VR1 to its maximum resistance.

When the battery is fitted, the alarm should sound after capacitor C1 has had time to charge. If it does not sound, there may be sufficient local mains electrical radiation for the unit to pick it up undesirably. If so, reduce the resistance of VR1 to reduce the circuit’s sensitivity, although the need to do this is unlikely.

Now the oscillator should only stop when you grip the aerial insulation. Once the circuit is working correctly it should be mounted in a plastic box of a size that will accommodate it and the battery. The piezo sounder was purchased mounted in a plastic package suitable for fixing to the box but uncased elements are also available and if this type is used it should be glued to the inside of the box lid. It may be found necessary to drill a small hole in the lid to allow the sound to be heard, although the lid will generally act as a sounding board.

The sensitivity of the unit may be high enough for the aerial wire to be left inside the box and the box simply placed alongside the mains cable. If this is not the case, the wire should be brought out and wrapped around the appliance mains lead a few times. This will not only increase the apparent sensitivity of the circuit but will also ensure that the unit remains in close proximity to the cable.

Alternately, a Bulldog type paper clip attached to the box and wired to the base of TR1 could be used as an aerial and a means of securing the device to the cable.

Once complete and in proximity to the mains cable of the appliance, or clipped to it, final testing of the circuit is very simple. Unplugging the mains lead or switching off the power at the mains socket should cause the alarm to sound. Restoring the power should mute the alarm. It is probably better to use a small appliance such as a table lamp to do this initially, rather than your freezer!

Other Applications
As well as a mains failure alarm, the circuit will be found to be a useful addition to any electrician’s tool box. It may be used in identifying which circuit is connected to which fuse in the main fuse box or indeed to ensure that the cable which is about to be cut is not carrying a mains voltage.

The circuit can be attached to the cable and the fuses removed in turn until the alarm sounds. Most non-contact cable testers provide only a visual indication and cannot be attached to a cable.

The circuit is also useful in finding a break in a cable or mains circuit. One application which comes to mind is the annual quest to discover which lamp in the Christmas tree lamp chain has mysteriously become disconnected during its yearlong storage.

Another is checking the fuse in a mains plug without the need to undo the whole thing – only to find that the fuse is OK!

Voltage Booster – High Voltage from a 5V Supply

noreply@blogger.com (Anonymous) — Fri, 04 May 2012 15:50:00 +0000

Digital circuits operating from 5V regulated supplies are common but occasionally a higher voltage is required, perhaps for a bio-medical circuit, or for liquid level measurement or for monitoring high resistance contacts. For such circuits a means of generating a high voltage from the 5V supply can be a solution. Diode/capacitor multipliers can offer advantages over switched-mode circuits, since they do not use inductors, are easier to design and troubleshoot and often generate less radiated interference.

The principle of the voltage multiplier is fairly well known. A capacitor is used with a square wave drive signal to “pump” current through a pair of diodes, roughly doubling the supply voltage. A series of such stages can be cascaded to raise the voltage in multiples of the supply, but it is possible to improve efficiency and reduce the number of stages by using two driving signals with opposite phases. However, each diode incurs a drop of about 0.6V so with two diodes per stage, and with an initial supply of just 5V this becomes significant, leading to poor efficiency and an impractical number of stages.

These problems are overcome in the Voltage Booster circuit design of Fig.1 by increasing the voltage before multiplication with IC1, an SI7660 “negative rail generator” (not the ICL7660 – ARW). The additional negative supply is generated very efficiently since switching is performed not by diodes but by CMOS switches in the IC, which cause almost no voltage loss at low currents.

Pin 2 of this IC also drives a diode multiplier which, even though it does suffer from diode voltage drops, still generates about +8.8V. With a minimum of components this device generates a pair of supply rails with an overall potential of more than 13.5V.

This voltage is supplied to IC2, a 4047B used as an oscillator whose frequency is set to about 400Hz by resistor R1 and capacitor C6. The oscillator employs an internal divide-by-two stage so that the outputs at pin 10 and pin 11 are complementary 200Hz square waves with perfect 1:1 duty cycles, ideal for multiplier driving.

With just the five diodes D3 to D7 shown, the circuit generates 60 volts with no output load. The output may be treated as having a fairly linear source resistance of about 200k since a drop of about 10V results for each 50uA of current drawn. In other words it will sustain 50V across a 1M resistor, or about 40V across one of 390k.

At no load the quiescent current was measured at just 650uA rising to about 1.9mA at 100uA load. The capacitors used were all resin-dipped ceramic types, though other non-electrolytic types with suitable voltage ratings could be used. The diodes are inexpensive 1N4148 signal types.

Increased output power could be obtained by using a higher frequency for IC2 and larger capacitors for C2 and C3, though this would increase the standby current. More (or less) stages in the multiplier chain could be used to obtain different output voltages if needed.

Finally, the value of capacitor C11 could be increased to allow the build-up of a larger charge, but this would obviously cause a time lag at switch-on or discharge. The circuit reaches full output voltage in well under a second.

PIC Based Air Quality Monitor

noreply@blogger.com (Administrator) — Fri, 27 Apr 2012 04:41:00 +0000

Idealy if you have a combustion heater in your home, there should be some means of monitoring the air quality. This is where the Air Quality Monitor comes in, it measures both carbon dioxide and carbon monoxide levels, and displays the results on LED bargraphs. If the concentration of either of these gases rises above a preset level, a loud alarm will sound, which means that you should turn off the heater and open the room up to fresh air.

Each bargraph comprises eight LEDs that light invidually to show eight distinct levels. In between values are displaed by lighting two adjacent LEDs. This gives a total of 15 levels that can be displayed. The four lower LEDs are green, followed by two orange and then two red LEDs. An automatic dimming circuit ensures that the LED displays are not too bright at night.

In addition, the alarm sounds if any of the top three LEDs light in either display. There are three alarm levels:

Main alarm sounds if the top LED lights. This consists of a 64ms-long 4kHz tone that repeats every 0.5s.
Less urgent alarm sounds if the second top LED is lit (top LED off). This alarm gives a 32ms-long 4kHz ‘chip’ every four seconds (4s).
Warning alarm sounds if the third top LED is alight. This alarm mode gives a brief 16ms 4kHz ‘chirp’ every 16 seconds (16s).

An internal fan at one end draws air through the box so that the internally mounted CO and CO2 sensors are presented with a continuous sample of the air that’s being monitored. Power for the unit comes from a 12V DC 500mA plugpack.

Circuit details

Take a look now at the Air Quality Monitor circuit of Fig.1. It’s based on two gas sensors and a PIC microcontroller (IC1). The microcontroller monitors the sensor signals and drives two multiplexed LED bargraph displays.

We’ll start by looking at the CO2 sensor. This consists of a heater coil and a solid electrolyte cell comprising a lithium (Li) cathode and a potassium (Ka) anode. When these electrodes react with carbon dioxide, a potential difference is produced between them that varies with the CO2 concentration.

The sensor is built into a metal housing and is exposed to air (and to CO2) via a stainless steel mesh. Its output in normal air (ie, with a normal CO2 concentration) is typically 325mV. This voltage falls with increased CO2 concentrations beyond 400ppm (parts per million) or 0.04%.

The CO2 concentration in normal air is 0.0314%, but this can increase to 5% in air that’s directly exhaled from the lungs. At this latter level, the sensor’s output will be well below 250mV (compared to 325mV in standard air).

The CO2 sensor’s output appears across its ‘A’ and ‘B’ terminals, and has a very high impedance, so any loading will drastically reduce this output. As a result, the manufacturer recommends monitoring the voltage using a circuit that has a 100Gohm to 10Tohm input impedance and an input current not exceeding 1pA. To comply with these requirements we have used an LMC6482 CMOS op amp (IC2a) to buffer the sensor signal. Its input impedance is 10Tohm while the input current is typically just 0.02pA.

CO2 threshold

IC2a is wired as a non-inverting stage with a gain of about 11, as set by the 10k and 1k feedback resistors. As a result, a 315mV output from the sensor (ie, in normal air) should result in a 3.47V output from the op amp.

In practice, we found that the output from the particular CO2 sensor we used was greater than 315mV in normal air, causing IC2a’s output to go above 5V. Consequently, trimpot VR2 has been added so that IC2a’s output can be level shifted, to correctly set the output to 3.47V in normal air. In practice, VR2 is adjusted so that the lowest LED in the CO2 bargraph is just off in normal air.

Note that the output from the sensor is valid only after it has been heated sufficiently. This is achieved by connecting a 6V supply across the internal heater element.

CO sensor

The output from the CO sensor is mounted at the AN3 input of IC1. However this sensor operates differently from CO2 sensor, in that it varies its resistance with CO concentration.

The specifications state that this sensor must initially be heated using a 5v supply connected acress its heater element for 60s. the heater current is then reduced by placing just 1.4v across the element for 90s period. The CO concentration is then measered. In practice, this means that measurements are repeated at 150s intervals.

Q5 is controlled by IC1’s RB1 output (pin 7) and turns on when its gate is pulled high. In operation, RB1 switches Q5 on for 60s to provide the heating current. RB1 then goes low for 90s, which switches Q5 off, so that the measurement can be made. During this 90s period (ie, with Q5 off), the CO sensor’s heater is effectively in series with the 2.2ohm and 100ohm resistors connected across Q5. As a result, the current through the sensor drops to 42.34mA, which means that the voltage across the heater is now 1.397V (ie, 33 x 0.4234). That is close enough for practical purposes to the 1.4v value specified.

During set-up, VR5 is set so that AN3 is at 0.5V when the sensor is in normal air. However, this signal voltage can rise to around 3V when the CO concentration is over 300ppm.

The maximum bargraph level for CO is adjusted using trimpot VR4. It effectively forms a voltage divider across the 5V supply, and its output is applied to the AN4 (pin 3) input of IC1. This voltage, along with the sensor voltage on AN3, is then used by IC1 to calculate the bargraph display level.

Display dimming

Automatic display dimming is achieved using a light-dependent resistor (LDR). As shown, the LDR is connected in series with trimpot VR6 across the 5V supply to form a voltage divider. The output of this voltage divider is connected to the AN1 input (pin 18) of IC1.

In bright light, the LDR’s resistance is 50kohm or less, and so the voltage applied to IC1’s AN1 input is pulled close to the 5V supply. This signals IC1 to drive the LEDs at full brightness.

Conversely, at lower ambient light levels, the LDR’s resistance increases and the voltage at AN1 decreases. As a result, IC1 now drives the LEDs with a reduced duty cycle. This is achieved by using a longer dead time, ie, the time between when one transistor switches off and the next one switches on. This effectively reduces the length of time that the LEDs are lit and hence reduces their brightness.

Piezo alarm

The alarm feature is provided by using the pin 6 PWM (pulse-width modulation) output of IC1 to drive a piezo transducer. Its frequency of peration is set to 4kHz (50% duty cycle) and there are three alarm modes, as described earlier.

Note that the alarm is only activated when one of the top three LEDs in either bargraph is lit.

Initial adjustments

Before using the unit, it’s necessary to adjust the full-scale sensitivity and threshold level of each ‘bargraph’ display. The initial procedure is as follows:

1) Adjust trimpot VR3 (CO2 level) to give 3V at test point TP3. This sets the CO2 full-scale sensitivity to about 10,000ppm or 1%.

2) Adjust trimpot VR4 (CO level) to give 3V at TP4. This sets the full-scale CO sensitivity to about 300ppm.

3) Adjust VR2 so that the bottom LED of the CO2 bargraph just extinguishes (ie, no LEDs lit). This should be with test point TP2 at just under 3V.

4) Wait 60s after applying power, then blow on the CO2 sensor to expose it to extra CO2 gas. Check that the CO2 bargraph now shows a fullscale reading (ie, top LED lit). If the piezo transducer is connected, check that the alarm sounds with any of the top three LEDs lit.

CO sensor adjustments

Making the adjustments for the CO sensor is a slow process, since it requires a ‘burn-in’ period of 48 hours. The unit must, therefore, be left on for 48 hours before making the final adjustment. Additionally, as stated in the circuit description, the sensor is heated for 60s and then allowed to respond to the gas over a 90s period before each measurement is made. This means that it will take 2.5 minutes to get the result after each adjustment.

Initially, however, you can bypass the 48-hour burn-in period and make the initial adjustments straight away.the final ‘touch up’ adjustment can then be made after the burn-in period.

The first step is to adjust trimpot VR5 so that TP5 is at 0.5V right towards the end of the 90s measurement period, ie, when the sensor is in fresh air. However, this requires some means of monitoring the heating and measurement cycles.

In practice, you can either use a second multimeter to monitor the drain (D) of Q5, or use a diagnostic tool that’s built into the Air Quality Monitor that shows the heating/measurement cycles. The procedure for each method is as follows:

METHOD 1: if you have a second multimeter, connect it between Q5’s tab (ie, its drain) and GND. Q5’s tab will be close to 0V during the heating cycle and at 4.3V during the measurement cycle. Adjust VR5 to set TP5 to 0.5V near the end of the 90s measurement cycle.

METHOD 2: if using the inbuilt diagnostic tool, start by adjusting VR4 fully clockwise, so that TP4 is at 0V. This will now cause the CO bargraph display to show the heating and measurement cycle.

During the 60s heating cycle, the top two red LEDs will be lit. Then, during the 90s measurement cycle, the red LEDs switch off and the four green and two orange LEDs initially light. These LEDs then extinguish one at a time, starting with the topmost orange LED and continuing at 15s intervals until the bottom green LED goes out at the end of the 90s period.

The unit then reverts to the heating mode again, with the top two LEDs lit. Note that if VR4 is not set all the way down to 0V, only the top LED will light. Additionally, the alarm will sound if VR4 is set below 2V, so the piezo transducer should be unplugged during this procedure.

Assuming VR4 is set for 0V at TP4, it’s just a matter of adjusting VR5 so that TP5 is at 0.5V when the lowest one or two green LEDs are alight. Once that’s done, be sure to readjust VR4 so that TP4 is at 3V.

Installation

The Air Quality Monitor should be mounted near to the combustion heater and preferably on a wall, so that the display can be easily seen.

The box has mounting holes that are accessed with the lid off, so it’s easy to fix in aposition.

Note that it’s normal for CO2 levels to rise while the heater is on. However, the ventilation should be increased if the indicated level rises past the low region on the bargraph.

The carbon monoxide (CO) level in the room should be kept to an absolute minimum and this can be achieved by ensuring that the heater is operating correctly. With wood heaters, this means allowing the temperature to rise sufficiently after the fire has been started, to ensure clean combustion, before reducing the air intake to slow the combustion process.

Finally, never use treated or manufactured timber such as treated pine,MDF,chip board, hardboard or similar in wood fires. These product can produce nonxious fumes during combustion.

Source:EPE

Download:code files

12V 10A High current Power Supply with battery backup

noreply@blogger.com (Anonymous) — Mon, 23 Apr 2012 11:00:00 +0000

The full circuit diagram of the 12V 10A Power Supply is shown in Fig.1.Power of about 18V to 25V is applied to screw terminals pins 1 (+V) and 2 (–V).Although the power supply was originally designed to run packet radio transceivers, the unit is not just confined to this type of radio. In fact, any radio can be used with this power supply. It can also be used as a main source of power; ie the battery, or used as a standby source of power in the event of a power failure. There would be a float charge for the battery when mains voltage is applied, and the battery can be relied on to supply current to equipment when the mains supply fails.

The input current to the circuit is limited by a 5A fuse (FS1) for protection purposes. Relay RLA is a small ‘homemade’ reed type that is set to close the contacts when the current drawn through this relay coil is in the order of about 1.5A. When this occurs, RLA’s contacts close, current is drawn through resistor R1, LED1, R2 and onto R3. The voltage across resistor R3 is sufficient to turn on FET TR2, which supplies power to a 12V cooling fan.

Capacitor C1 in the circuit forms a timing function, where a delay of about one second is experienced at switch on, and about five seconds at switch off.

Zener diode D2 limits the gate/source voltage of TR2 to 2.7V. Field effect transistor TR2 is used to switch a cooling fan on in order to cool the circuit down during high and prolonged current demand. A 1.3W 6.2V Zener diode (D1) is selected for an 18V input, or 12V for a 25V input. The Zener diode can be removed and a shorting link replace it if a 24V fan is used. An old redundant computer fan was used in the prototype to assist in the units cooling. When the reed relay contacts of RLA1 are closed, LED1 is illuminated to indicate that a current higher than 1.5A is being drawn from the mains derived power supply, and cooling is being effected.

Voltage regulation
Transistors TR1, TR3, TR5 and IC1 form a voltage regulator that is commonly seen in power supplies. TR1 is a Darlington NPN transistor with a high current gain that acts with TR3. These two transistors act with resistor R4 and the parallel resistors R5 and R6 to form a current limit device.

The preset potentiometer VR1 is used to set the current limit value, when there is 0.7V across the base/emitter junction of TR3. In the prototype the current limit was set to 3A.
During a current limit condition (usually a fault condition in a conventional series-regulated power supply), the circuit can become somewhat unstable. To increase the stability of the circuit, a 100µF electrolytic capacitor (C3) is included to smooth the supply to the base (B) of transistor TR1. In addition a 100nF capacitor (C4) is added in parallel with C3.

Transistor TR5 and IC1 act together as a high current voltage regulator. Including R7 and TR5 increases the current handling of IC1, which, on its own is limited to 1.0A. Choosing a value of 1ohm for R6 ensures that TR3 bypasses current around IC1 which is set at approximately 450mA, less than half the maximum current handling of this device, IC1.

Charging circuit
Diode D6 provides isolation between the battery and the charging circuit. Should the mains supply fail for any reason, this diode inhibits the battery from discharging through the charging circuit. Transistor TR4 supplies current to the relay coil RLB, which is then energised and hence the relay contacts are closed. The battery can now be charged and utilised in the power supply.

In the base circuit of TR4 is a 10V Zener diode (D5), this simplearrangement ensures the transistor is turned on when mains power is applied and the units output voltage is above 10.7V. Hence, relay RLB is energised when the mains supply is connected or the battery is showing a healthy state of charge. This occurs when the output from the regulator
(IC1) is above 10.7V. This also ensures the voltage at the battery is sufficient to power; in the event of a mains failure. Once the battery voltage drops below about 10.7V, relay RLB1 contacts will open; disconnecting the battery.

Charging of gel-cell batteries
The technique for charging gel-cell batteries uses a constant-voltage, current-limited system. The constant voltage is generated and set to a specific value, and for a 12V battery it’s 13.2V (2.275V per cell) – (Yuasa gel-cell, NP12-12 series – 12V, 12AH).

The maximum charging current should be set to one quarter of the batteries AH (amp/hour) capacity. For a 12AH battery, this would equate to 3A. The maximum charging current could be set lower, however the battery will take a little longer to reach its full capacity after a deep discharge.

Should a deeply discharged battery be attached to this power supply, the battery could take a very large current at switch on, and the output of the power supply could drop severely. The charging current will limit at the value determined by preset VR1, and the battery should slowly rise to a value around 12V within minutes.

Reed relay
Reed relay RLA’s coil is fabricated with approx. 18½ turns of 1.5mm enamelled copper wire wound on a 3.5mm drill; the drill is used as a mandrel, or coil former. The coil leads are formed to fit the PCB – see Fig.2. The coil is an air wound type, 18½ turns closely wound with an internal diameter of 3.5mm. The glass-encapsulated ‘reed’ insert (contacts) is set centrally within the coil, and should have an actuating current of approximately 1.5A. In the prototype, a rubber sleeve was positioned around the reed to help space it and keep it centrally mounted within the assembly.

Against convention
Details of the interwiring between the circuit board and off-board components is shown in Fig.4. Note resistor R13 and LED are located off the PCB and provide a power on indication.
Going against convention, where power supplied to a circuit is generally supplied via a socket. In this project 3-pin XLR connectors have been used. The output to the battery has a chassismounted plug, and the battery a cable socket.

This is deliberate, as it was considered that should anything accidently short across a socket, it would be preferable if it shorted across the lowest current source, ie, the power supply. The higher source of power, a gel-cell battery could deliver well over 100A if the terminals are shorted across, and explode as a consequence. The power supply current,
even under a short circuit condition, is set by VR1 to 3A.

The connecting wire and connectors on the input supply, and from the output of the circuit, must be adequately rated for the current drawn. The prototype used 5A wire on the input and 25A connecting wire on the output side.

Relay
The choice of the output relay RLB really limits the current-handling capacity of the power supply, since its contacts will be passing the full load current. The footprint of the relay on
the PCB provides a choice of contacts available from DPDT to SPDT, with contact ratings up to 15A in SPDT arrangement, which would be suitable.

In this project, a DPDT relay was used and was found to be adequate in passing the 10A which the unit was designed to supply. It must be stressed that the PCB copper tracks carrying this level of current must be built up with solder to reduce the track resistance, and hence the track current carrying capacity.

Setting up
For the initial setting up, include the 47ohm ‘test’ resistor. If not, a current limited power supply supplying 18V to 25V could always be used, with a current limit initially set to 100mA.

Ensure the preset pots VR1 and VR2 are both set fully anti-clockwise. This will ensure that the current limit is set to the minimum value, and the output voltage is set to maximum. Setting the output voltage to maximum will ensure that contacts RLB1 are closed.

No equipment or battery should at this point be connected to the circuit. Wait until it has been fully tested and the output voltage set to the nominal output voltage of 13.2V to 13.6V.
As a temporary measure, connect a pushbutton switch between pins 3 and 4 on the circuit board, and hold the pushbutton down as you adjust the output voltage until the relay contacts of RLB close. The output voltage of the circuit is adjusted with the aid of VR2 and is set to 13.2V. The output voltage is measured across terminals 3 and 6 on the circuit board.

Connect up the circuit supplying current to the power supply unit, slowly increasing the input voltage set to 18V (using a bench power supply and a 47ohm series test resistor), and constantly monitor the current at this point. The current drawn should be in the order of 123mA with the relay RLB energised. At this point LED2 should be illuminated and LED1 extinguished. The output voltage is monitored with a DVM and set to 13.2V for a 6-cell gelcell battery (float charging the battery at 2.275V per cell) as VR2 is adjusted).

The charging current is set by the manufacturer, and can, as a general rule of thumb, be set to a maximum current of one quarter of the maximum amp/hour (AH) capacity of the battery. For example, a 12AH battery can be charged at a maximum current of 3A. This is the current limit value set by the unit, using trimpot VR1. Setting preset VR1 to the 11 o’clock position will set the current limit (maximum charging current) to 3A. Y o’clock – 2.5 amps, and w o’clock – 2.0 amps.

Limiting current
To set the limiting current accurately, a discharged battery could be connected in circuit, in which case the 47ohm test resistor (if used) would have to be removed from the input side of the power supply. Monitor the current drawn by the circuit using an ammeter and set the current by adjusting VR1 to match the battery used. Important: Do not set the charging current too high, as this will cause premature failure of the battery. If a discharged battery is attached to the power supply, the voltage at the battery may be low and hence the charging circuit voltage. The battery voltage will depend upon the state of charge, low enough to want to draw the power supplies limiting current. In this case, an ammeter on the input side can be used to read this current.

Adjusting VR1 will adjust the current limit of the circuit and hence the maximum charging current for the battery. As mentioned earlier, the maximum value for this current will actually be dependant on the capacity of the battery. A larger capacity battery can, of course, be used, but from a deep discharge will take longer to reach a full charge.

In use
The power supply has been very successful in use with the homebase station radios, and no appreciable problems with 12AH and 24AH batteries. It has to be remembered that the power supply has to be left on after use to top up the battery; leaving it on overnight should not be a problem.

Source: EPE

Universal Remote Control Receiver

noreply@blogger.com (Anonymous) — Tue, 13 Mar 2012 05:57:00 +0000

The circuit diagram shown in Fig.1 uses a PNA4602M IR detector to receive an IR signal from the remote control. It’s a neat device, which contains an IR receiver, amplifier and demodulator together in a single 3-pin package, and it’s normally used in circuits where you want to decode an IR remote control signal.

The way the circuit (Fig.1) works is as follows. When there is no IR coded signal present, the output pin of IC1 remains high. This high signal is fed to the trigger input of the 555 timer (IC2), which being configured as a monostable timer, prevents the timer operating.

Whenever you briefly press any key on the old remote control its IR signal is received by IC1 and output pin 1 produces a train of fast moving high and low pulses, which mimic the IR signal code sent by the remote control. We are not interested in this code, but as soon as the signal switches low it triggers the monostable timer IC2 and its output pin 3 goes high for a short period of time, set by resistor R2 and capacitor C2.

The 555 timer (IC2) basically acts as a debounce circuit, and its output is fed into the clock pin (CLK) of the 4013B dual Dtype flip-flop IC3. Only one half of the chip is used in this application, and it is configured as a bi-stable flip-flop.

An initial clock signal into IC3 causes its output pin 2 to go high, and it remains high until another IR signal is received. To switch the output off you simply press any button on the old remote control again and the above operation is repeated, but this time IC3’s output is switched off and remains off until another signal is received. The output basically acts as a toggle switch.

The output pin in this circuit drives an LED to show the output status. However, the output could be easily interfaced to drive a transistor and relay to switch a higher load. The circuit could then be used to remotely switch equipment in your garage, shed or office.

Don’t forget that you need to use the circuit in an area where IR remote controls are not normally present, because the circuit will operate with virtually any IR remote control. Capacitor C1, positioned across the output of the IR receiver (IC1), stops spurious triggering from artificial lighting, which can occur if this isn’t in place.

Schematic:

Components Required:

Components	values	quantity
R1-R2	100k ohm	2
R3	270 ohm	1
R4	390 ohm	1
C1-C2	4.7uF/16v	2
LED1-LED2	Red LEDs	2
IC1	PNA4602M	1
IC2	555	1
IC3	4013	1

LDR Based Water Pump Controller

noreply@blogger.com (Anonymous) — Sun, 04 Mar 2012 05:50:00 +0000

Here is a simple solution for automatic pumping of water to the overhead tank. Unlike other water-level indicators, it does not use probes to detect the water level and hence there is no probe corrosion problem. It has no direct contact with water, so the chance of accidental leakage of electricity to the water tank is also eliminated.

Two important advantages of the circuit are that the water level never goes below a particular level and no modification in the water tank is required. Fig. 1 shows the circuit of the water pump controller. The circuit uses an LDR-white LEDs assembly to sense the water level. It forms a triggering switch to energise the relay for controlling the pump. The LDR-LEDs assembly (shown in Fig. 2) is fixed on the inner side of the cap of the water tank without making contact with water. The light reflected from the water tank is used to control the resistance of LDR1.

When the water level is high enough, light from the white LEDs (LED1 through LED3) reflects to fall on LDR1. This reduces the resistance of LDR1, increasing the voltage at the non inverting input (pin 3) of IC1. IC1 is used in the circuit as a voltage comparator. Resistors R4 and R5 form a potential divider to fix half of supply voltage to the inverting input of IC1. Normally, when the water tank is full, LDR1 gets more of reflected light because the distance between the water level and the face of LDR1 is minimal.

When white light falls on LDR1, the voltage at the non-inverting input (pin 3) of IC1 increases and its output goes high. This high output makes pnp transistor T1 non conducting and the relay remains de-energised. LED1 also remains ‘off.’ Since the water pump power supply is connected to the normally- open (N/O) contacts of relay RL1, pumping is stopped.

When water level falls, the amount of light reflected to LDR1 decreases and its resistance increases. This reduces the voltage at pin 3 of IC1 and its output goes low. This low output from IC1 makes transistor T1 conduct. Relay RL1 energises to close the N/O contacts and the motor starts pumping water. LED1 glows to indicate the curpumping of water.

Assemble the circuit on a general purpose PCB and enclose in a suitable cabinet. Solder the white LEDs-LDR1 assembly on a separate PCB and use a separate power supply for it. Mount LEDs behind the LDR. Otherwise, light from the LEDs will affect the working of the circuit. Connect LDR1 to the main circuit board at ‘A’ and ‘B’ points.

Fix the LEDs-LDR1 assembly on the inner side of the water-tank cap as shown in Fig. 3. Orient the LEDs and the LDR such that when the water tank is full, the light emitted from the LEDs and reflected from the water surface falls directly on LDR1. The distance between the upper level of water and the LEDs-LDR setup should be minimal, ensuring that water doesn’t touch LDR1. Otherwise, the circuit will not function properly. By using more white LEDs, this distance can be increased. Cover the LDR with a black tube to increase its sensitivity.

You can fix the main unit at a convenient place and connect it to the LEDs-LDR assembly through wire. Select the relay according to the horsepower (HP) of the water pump. After arranging the setup (with maximum water in the tank), adjust VR1 until LED1 stops glowing. In this state, the relay should de-energise. When the water level decreases, the relay automatically energises to connect mains to the motor and it starts pumping water.

Schematic:

Components Required:

Components	Values	quantity
R1-R3	100 ohm	3
R4-R5	12k ohm	2
R6	220 ohm	1
VR1	10k ohm	1
C1	220uF/25v	1
C2	470uF/25v	1
D1	1N4148	1
D2	1N4007	1
T1	BC558	1
IC1	uA741	1
RLA	See text	1
S1	On/off switch	1
Batt	9v battery	1
LED1-LED3	White LEDS	3
LED4	Red LED	1
LDR	---	1

Portable Lamp Flasher

noreply@blogger.com (Anonymous) — Thu, 23 Feb 2012 10:27:00 +0000

Here is a portable, high-power incandescent electric lamp flasher. It is basically a dual flasher (alternating blinker) that can handle two separate 230V AC loads (bulbs L1 and L2).

The circuit is fully transistorized and battery-powered. The free-running oscillator circuit is realized using two low-power, low-noise transistors T1 and T2. One of the two transistors is always conducting, while the other is blocking. Due to regular charging and discharging of capacitors C1 and C2, the two transistors alternate between conduction and non-conduction

states.

The collector of transistor T1 is connected to the base of driver transistor T4 through current-limiting resistor R5. Similarly, the collector of transistor T2 is connected to the base of driver transistor T3 through limiting resistor R6. These transistors are used to trigger Triac1 and

Triac2 (each BT136) through optotriacs IC1 and IC2, respectively, and switch on the power supply to external loads L1 and L2. IC1 and IC2 operate alternatively at a low frequency

determined by the values of capacitors C1 and C2.

The oscillator circuit built around transistors T1 and T2 generates low frequencies. When transistor T3 conducts, IC1 is enabled to fire Triac1 and bulb L1 glows. Similarly, when transistor T4 conducts, IC2 is enabled to fire Triac2 and bulb L2 glows.

Connect the power supply line (L) of mains to bulbs L1 and L2, and neutral (N) to T1 terminals of Triac1 and Triac2. You can also connect neutral (N) line of the external 230V mains supply to both loads (bulbs L1 and L2) as a common line and then route supply line (L) to respective loads (bulbs L1 and L2).

The circuit works off only 3 volts. Since current consumption is fairly low, two AA-type cells are sufficient to power the circuit. Assemble the circuit on a general-purpose PCB and enclose in a suitable plastic cabinet with integrated AA-size pen-light cell holder. Drill holes for mounting the ‘on’/‘off’ switch and power switching terminals. Also connect two bulb holders for bulbs L1 and L2.

Pinout:

Schematic:

Portable Lamp Flasher

Components Required:

Components	values	quantity
R1,R3	47k ohm	2
R2, R4	470 ohm	2
R5-R6	1k ohm	2
R7-R8	47 ohm	2
R9-R10	5.6k ohm	2
C1-C2	47uF/10v	2
C3	100uF/10v	1
T1-T4	BC547	4
IC1-IC2	MOC3021	2
L1-L2	60W bulb	2
Triac	BT136	2
S1	On/off switch	1
Mics	3v battery	1

Long-Range IR Transmitter

noreply@blogger.com (Anonymous) — Sun, 19 Feb 2012 10:21:00 +0000

Most of the IR remotes work reliably within a range of 5 metres. The circuit complexity increases if you design the IR transmitter for reliable operation over a longer range, say, 10 metres. To double the range from 5 metres to 10 metres, you need to increase the transmitted power four times.

If you wish to realise a highly directional IR beam (very narrow beam), you can suitably use an IR laser pointer as the IR signal source. The laser pointer is readily available in the market. However, with a very narrow beam from the laser pointer, you have to take extra care, lest a small jerk to the gadget may change the beam orientation and cause loss of contact.

Here is a simple circuit that will give you a pretty long range. It uses three infrared transmitting LEDs (IR1 through IR3) in series to increase the radiated power. Further, to increase the directivity and so also the power density, you may assemble the IR LEDs inside the reflector of a torch.

For increasing the circuit efficiency, a MOSFET (BS170) has been used, which acts as a switch and thus reduces the power loss that would result if a transistor were used. To avoid any dip during its ‘on’/‘off’ operations, a 100μF reservoir capacitor C2 is used across the battery supply. Its advantage will be more obvious when the IR transmitter is powered by ordinary batteries. Capacitor C2 supplies extra charge during ‘switching on’ operations.

As the MOSFET exhibits large capacitance across gate-source terminals, a special drive

arrangement has been made using npn-pnp Darlington pair of BC547 and BC557 (as emitter followers), to avoid distortion of the gate drive input. Data (CMOS-compatible) to be transmitted is used for modulating the 38 kHz frequency generated by CD4047 (IC1). However, in the circuit shown here, tactile switch S1 has been used for modulating and transmitting the IR signal.

Pinout:

Schematic:

Long-Range IR Transmitter

Components Required:

Components	values	quantity
IR1-IR3	IR LEDs	3
C1	390pF	1
C2	100uF/25v	1
R1-R2	10k ohm	2
R3	2.2k ohm	1
R4	10 ohm, 0.5W	1
VR1	1k ohm	1
T1-T2	BC547	2
IC1	CD4047	1
T3	BS170	1
S1	Push to on switch	1
S2	On/off switch	1
Mics	9v battery	1

Three Phase Appliance Protector

noreply@blogger.com (Anonymous) — Thu, 16 Feb 2012 10:16:00 +0000

Many of our costly appliances require three-phase AC supply for operation. Failure of any of the phases makes the appliance prone to erratic functioning and may even lead to failure. Hence it is of paramount importance to monitor the availability of the three-phase supply and switch off the appliance in the event of failure of one or two phases. The power to the appliance should resume with the availability of all phases of the supply with certain time delay in order to avoid surges and momentary fluctuations.

The complete circuit of a threephase appliance protector is described here. It requires three-phase supply, three 12V relays and a timer IC NE555 along with 230V coil contactor having four poles.

Relays RL1 and RL2 act as a sensing devices for phases Y and B, respectively. These relays are connected such that each acts as an enabling device for the subsequent relay. Therefore the combination of the relays forms a logical AND gate connected serially.

The availability of phase R energises relay RL1 and its normally opened (N/O) contacts close to connect phase Y to the input of transformer X2. The availability of phase Y energises relay RL2 and its N/O contacts close to connect phase B to the input of transformer X3, thus applying a triggering input to timer IC NE555 (IC1).

Therefore the delay timer built around NE555 triggers only when all the phases (R, Y and B) are available. It provides a delay of approximately four seconds, which energises relay RL3 and its N/O contact closes to connect the line to the energising coil of four-pole contactor relay RL4. Contactor RL4 closes to ensure the availability of the three-phase supply to the appliance.

The rating of contactor RL4 can be selected according to the full-load current rating of the appliances. Here the contact current rating of the four-pole contactor is up to 32A. The availability of phases R, Y and B is monitored by appropriate LEDs connected across the secondary windings of transformers X1, X2 and X3, respectively. Hence this circuit does not require a separate indicator lamp for monitoring the availability of the three phases. When

phase R is available, LED1 glows. When phase Y is available, LED2 glows. When phase B is available, LED3 glows.

The main advantage of this protector circuit is that it protects three-phase appliances from failure of any of phases by disconnecting the power supply through the contactor and automatically restores the three-phase supply to the appliance (with reasonable time delay) when all the phases are available.

Caution. To avoid the risk of electric shock, ensure that AC mains is disconnected during assembly of the circuit and double check everything before connecting your circuit to the

mains.

Schematic:

Three Phase Appliance Protector

Components Required:

Components	values	quantity
D1-D6	1N4007	6
C1-C3	100uF/35v	3
C4	4.7uF/35v	1
C5	10nF	1
R1-R3	3.3k ohm	3
R4	4.7k ohm	1
VR1	1M ohm	1
T1	SL100	1
IC1	NE555	1
X1-X3	230v to 12v 300mA	3
RL1-RL3	12v relay	3
RL4	230v coil rating, 4 contact, 32A	1
Mics	LEDs	3